RU2783402C1 - Method for processing radar signals for detecting targets and measuring their motion parameters in the selection zone and a radar sensor implementing it - Google Patents

Abstract

FIELD: radar technology.

SUBSTANCE: invention relates to the field of short-range radar with pulse modulation of radiation, intra-pulse reception and signal processing. The signal processing method and radar sensor (RS) can be used in systems for target detection and measurement of their motion parameters in the selection zone. A radar sensor for detecting targets and measuring their motion parameters in the selection zone, implementing the claimed method, contains an antenna, a Doppler receiving and transmitting module with quadrature outputs I(t) and Q(t) of converted signals, a signal processing unit and a synchronization and pulse generation unit, and the first and second analog are additionally introduced into it-digital converters, the signal inputs of which are connected to the quadrature outputs I(t) and Q(t) of the Doppler receiving and transmitting module, and the outputs to the signal inputs of the signal processing unit.

EFFECT: obtaining a range resolution from an RS with intra-pulse reception and processing of radio signals and providing the possibility of separate detection of multiple location objects in the selection zone, determining the distance to them and the speed of movement, as well as vibration parameters and the law of movement of the location object when processing each probing radio pulse.

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Description

The invention relates to the field of short-range radar with pulsed radiation modulation, intra-pulse reception and signal processing. The signal processing method and the radar sensor (RLD) that implements it can be used in systems for detecting targets and measuring their movement parameters in the selection zone.

The principle of operation of classical long-range radars with pulse modulation of radiation is based on the separation in time of the processes of formation of probing and reception of reflected radio signals. The reflected radio signals are received in the time interval between the emission of probing radio pulses. The distance to the target is determined by the delay time of the reflected radio pulse, and the speed of movement is determined by the frequency of the Doppler signal (see pp. 175-215, [1]).

The disadvantage of classical radars is the presence of a "dead zone", determined by the total time of formation of a probing radio pulse and the recovery time of the receiver sensitivity [2].

Known radar with pulse modulation of radiation [3-10], which has no dead zone near the antenna. These radars use the method of receiving a reflected radio signal [11] during the emission of a probing radio pulse, when the delay time τ of the reflected radio signal is less than the duration t _and probing radio pulses (τ<t _and ). In this case, the process of extracting information about the reflected signal is performed by the phase difference of the emitted and received radio signals due to their mutual coherence. The relative movement of the target and the RLD causes corresponding changes in the phase difference of these radio signals. These phase changes are converted into video pulses of duration t _and -τ at the output of the receiver mixer. Selection and "stretching" in time of these video pulses for the period of their repetition T _p by temporary gating by the sample-hold circuit and subsequent filtering provide the formation of a Doppler signal, which is in demand for measuring the speed of the target and solving the problem of its detection.

The RLD selection zone is determined by the normalized amplitude of the Doppler signal relative to the receiver's own noise level. The near boundary of the range selection zone R _bl of these RLDs is equal to zero, and the far one is limited by the duration of the probing radio pulse: R _d =ct _and /2, where c is the speed of radio waves in free space. From the sides, the selection zone is limited by the width of the antenna radiation pattern in terms of "azimuth" and "elevation". Due to the noted properties, RRDs with intra-pulse signal reception and processing have increased resistance to passive interference outside the target selection zone and, therefore, are widely used in transport security and facility protection systems, as well as technological process control systems [12-17].

It should be noted that RLDs with intrapulse signal reception have both homodyne and autodyne construction [18] of transceivers. With homodyne construction, there is no separate local oscillator in the receiver, which simplifies the design of the radar. However, the use of an autodyne transceiver provides an additional simplification of the radar. It lacks a mixer and a decoupling device between transmitter and receiver, such as a circulator. An autodyne generator (autodyne) combines both functions simultaneously and thus has the simplest design, small dimensions and weight, and low cost of the transceiver [19]. These advantages are often decisive in favor of the use of autodyne RLDs in many applications [3-7; fifteen; 16].

Known radar, with increased resistance to passive interference, located in the selection zone in close proximity to the antenna. These interferences are caused, for example, by sleet, hail and rain, as well as by insects and birds. To protect against these interferences in RLD [15; 16] alternate transmission of probing radio pulses at two different frequencies is used. The resulting Doppler signals are proposed to be processed taking into account their phase difference. At the same time, signals from interference in the near range of the radar are in phase and can be easily eliminated by subtraction, and signals from distant targets differ in phase, which is used to determine the distance to the target.

The principle of interference suppression in the near zone of the radar, proposed in [20-25], is based on the successive transmission of two probing radio pulses, one of which is constant in width, and the other is subjected to pseudo-random modulation of duration. The video pulses generated at the output of the mixer are further amplified and smoothed by a low-pass filter, as described above. At the same time, the property of the radar suppression of radio signals from interference located outside the far boundary of the selection zone is preserved.

The expansion of the functionality of radio pulse radar ensures the use in the transceiver of not conventional (simple), but quadrature mixers [8; 26-28]. This achieves the formation at the output of the receiver of two Doppler signals I(t) and Q(t), shifted in phase relative to each other by an angle of ±π/2. A change in the direction of movement of the target causes a change in the sign of the phase difference of the signals to the opposite sign. This phenomenon is used to identify location objects in the direction of movement.

The closest analogue (prototype) in terms of technical essence, principle of operation and achieved positive effect is a radar designed to detect moving targets in a controlled selection area, which is claimed according to US patent 6492933 B1, publ. 12/10/2002, IPC ⁷ G01S 13/08, G01S 13/53. SSB pulse Doppler sensor and active reflector system / McEWan [28].

RLD contains (see Fig. 1 description of the prototype) series-connected reference clock generator, modulating pulse shaper, microwave generator, quadrature mixer, antenna, as well as an analog signal processing unit connected to the outputs of the quadrature mixer.

The method of generating and processing signals in RLD prototype in accordance with the description of the principle of its operation is as follows. The target moving in the selection zone is irradiated with probing radio pulses of a periodic sequence, during the emission of probing radio pulses, radio pulses reflected from the moving target are received, divided into two quadrature channels, mixed with a part of the probing radio pulses, the parts of these radio pulses overlapping in time are converted into the low-frequency region in the form of quadrature video pulses I(t) and Q(t), then the resulting video pulses are amplified in amplitude and smoothed by filtering, while Doppler signals are obtained, the amplitude of which determines the presence (detection) of a target in the selection zone, the frequency determines the speed of movement, and the the sign of the phase difference is the direction of its movement.

An analysis of the prior art has shown that the known technical solutions for RLDs with intra-pulse signal reception and processing have common significant drawbacks associated with the features of the operation of RLDs and signal generation. The essence of these features is as follows.

RLD with intra-pulse reception and signal processing provides determination of the range to the target by the moment of appearance of the Doppler signal when the target enters the selection zone [29]. In this case, it is possible to measure the current range to the target and the speed when it moves inside the selection zone in the radial direction relative to the radar. The determination of the current range to the target within the selection zone is found through counting the number n of periods of the Doppler signal relative to the far spatial boundary R _d of the selection zone. The current distance R _c to the target is found by the formula R _c =R _d -nλ/2, where λ is the radiation wavelength. This refers to the target, which can be represented as an equivalent point reflector, i.e. the size of the target is much smaller than the size of the selection zone.

However, the method described here has obvious disadvantages. For example, when changing the relative direction of movement of the target within the selection zone, an uncertainty of its position arises. A similar situation arises when the target enters the selection zone from the side, where the boundaries of this zone are determined by the antenna pattern. In such cases, it is not possible to determine the distance to the target, although it is in the selection zone. The situation with determining the current position of the target becomes even more uncertain in the conditions of multiple targets in the selection zone and in the case of location of spatially distributed objects (see pp. 194-199, [30]).

Thus, the essence of the problem of the prototype lies in the fact that the known methods of signal processing do not provide the possibility of reliable detection and determination of the motion parameters of individual objects of the location in the presence of multiple targets in the selection zone. The lack of range resolution of known radars is their significant limitation, especially in today's conditions of increased complexity of radar scenarios.

и

где k=0, 1, 2, … - порядковый номер зондирующего радиоимпульса; m=0, 1, 2, …М, затем в обоих каналах из текущих отсчетов

и

начиная с m=1, вычитают предыдущие значения отсчетов

и

:The solution to this problem is achieved by the fact that a method for processing radar signals for detecting targets and measuring their motion parameters in the selection zone is proposed, consisting in the fact that the controlled area of space with the targets located in it is periodically irradiated with probing radio pulses, during the emission of probing radio pulses, radio pulses are received, reflected from the targets, separate them into two quadrature channels, mix them with probing radio pulses, convert the parts of these radio pulses overlapping in time into the low-frequency region in the form of two quadrature video pulses I(t) and Q(t), then the converted quadrature video pulses obtained in these channels video pulses are sampled in time and stored at a set of time points t _km quadrature values

and

where k=0, 1, 2, ... - serial number of the probing radio pulse; m=0, 1, 2, …M, then in both channels from the current readings

and

starting from m=1, subtract the previous readings

and

:

and by the obtained values of the differences x _km and y _km calculate the amplitude A _km and the phase Ф _km , the signal for each sample, starting from m=1, according to the following formulas:

до элементов разрешения по дальности, в которых обнаружены цели, по формуле:further, for the obtained data of amplitude readings A _km , the values of the signal-to-noise ratio q _km =A _km /σ _w are calculated, where σ _w is the root-mean-square value of the intrinsic noise level at the receiver output, those values of sequence numbers are selected from the set of values q _km of the signal-to-noise ratio m, amplitudes A _km and phases Ф _km at which the signal-to-noise level exceeds the threshold: q _{km ≥q} thr , where q _thr is the threshold signal-to-noise ratio, distances are calculated

to the range resolution elements in which targets are detected, according to the formula:

сигнала, последовательно при изменении номера k зондирующего радиоимпульса выполняют дифференцирование, при этом получают значения мгновенной частоты

доплеровского сигнала n-й цели:where n is the serial number of the detected target; m is the number of the count to the n-th target, for which the excess of the threshold level q _then was detected; t _and - the duration of the radio pulse; c is the propagation velocity of microwave radiation; M is the number of readings during the time t _and , and in relation to the samples of the received readings of phase

signal, successively when changing the number k of the probing radio pulse, differentiation is performed, while obtaining the values of the instantaneous frequency

Doppler signal of the n-th target:

и

- производные по времени от x_km и y_km в каналах I(t), Q(t) соответственно, значения которых определяют методом конечных разностей, для этого берут соседние значения переменных, например, при k и k±1, по полученным результатам расчета доплеровской частоты

вычисляют относительные скорости движения целей по следующей формуле:where

and

- time derivatives of x _km and y _km in the channels I(t), Q(t), respectively, the values of which are determined by the finite difference method, for this, the neighboring values of the variables are taken, for example, at k and k±1, according to the results of the calculation doppler frequency

calculate the relative velocities of the targets according to the following formula:

n-ой цели последовательно при изменении номера зондирующего радиоимпульса k сглаживают, например применением операции «скользящее среднее» или алгоритма фильтрации Калмана, далее численным методом, например, конечных разностей для каждой n-ой цели решают дифференциальное уравнение

при этом предыдущее решение этого уравнения, начиная с нулевого, берут в качестве постоянной интегрирования последующего решения и, тем самым, получают для каждой n-ой цели закономерность изменения полной фазы Ф⁽ⁿ⁾(t) во времени, далее определяют искомую функцию закона движения n-ой цели по формуле:where c is the propagation velocity of microwave radiation; ω ₀ \u003d 2πƒ ₀ - circular frequency of microwave radiation, the results of calculating the speed

The n-th target is successively smoothed when the number of the probing radio pulse k is changed, for example, by using the “sliding average” operation or the Kalman filtering algorithm, then by a numerical method, for example, finite differences for each n-th target, the differential equation is solved

at the same time, the previous solution of this equation, starting from zero, is taken as a constant of integration of the subsequent solution and, thus, for each n-th goal, the regularity of the change in the total phase Ф ⁽ⁿ⁾ (t) in time is obtained, then the desired function of the law of motion is determined n-th goal according to the formula:

where λ is the wavelength of microwave radiation, the results of calculating the function of the law of motion R ⁽ⁿ⁾ (t) of the n-th target are successively smoothed when the number of the probing radio pulse k is changed, for example, by using the "moving average" operation or the Kalman filtering algorithm.

The proposed method for processing radar signals for detecting targets and measuring their motion parameters in the selection zone is implemented in a radar containing an antenna, a Doppler transceiver module with quadrature outputs I(t) and Q(t) of the converted signals, a synchronization and pulse shaping unit, and a processing unit signals, and the antenna is connected to the high-frequency port of the said Doppler transceiver module, to the control input of which the first output of the synchronization and pulse shaping unit is connected, the first and second analog-to-digital converters are additionally introduced into it, the signal inputs of which are connected to quadrature outputs I(t) and Q(t) of the Doppler transceiver module, and the outputs - to the signal inputs of the signal processing unit, while the second output of the synchronization and pulse shaping unit is connected to the clock input of the signal processing unit, and the third output is connected to the clock inputs of the first and second analog-to-digital converters founders, while

the synchronization and pulse shaping unit contains a reference clock generator connected to the inputs of the frequency multiplier and the counter-frequency divider, while the output of the frequency divider counter is the first output of the synchronization and pulse shaping unit, and the output of the frequency multiplier is its second output, while the first and the second output is also connected to the inputs of the pulse selector, the output of which is the third output of the synchronization and pulse shaping unit;

Doppler transceiver module with quadrature outputs I(t) and Q(t) of the converted signals, made according to the homodyne scheme and contains a transmitter, the first and second mixers, the first and second power dividers, a circulator and a phase shifter with quadrature outputs of high-frequency signals, while the transmitter is connected its high-frequency output to the input of the first power divider, the first output of which is connected to the first input of the circulator, and the second - to the input of the phase shifter, while the input of the second power divider is connected to the third input of the circulator, and the second input of the circulator is a high-frequency port of the Doppler transceiver module, while the outputs of the second power divider are respectively connected to the signal inputs of the first and second mixers, the heterodyne inputs of which are connected to the quadrature outputs of the phase shifter, while the transmitter control input and the quadrature outputs I(t) and Q(t) of the first and second mixers are connected to the same inputs and outputs additional Lehr transceiver module;

Doppler transceiver module with quadrature outputs I(t) and Q(t) of the converted signals, made according to the autodyne scheme and contains a microwave generator connected to the first port of the quadrature mixer "per pass", made on the basis of a transmission line with the first and second placed in it mixers located at a phase distance equal to an odd number of one eighth of a wavelength, while the signal outputs of the first and second mixers and the second port of the quadrature mixer "per pass" are connected, respectively, to the outputs I(t), Q(t) and the high-frequency port of the Doppler transceiver module.

The technical result of the proposed method and the device that implements it is to obtain a new property for radar with intra-pulse reception and processing of radio signals - range resolution, which made it possible to separately detect a plurality of location objects in the selection zone and determine the parameters of their movement, and this possibility is realized when processing as at least two probing radio pulses. At the same time, the properties of the prototype device were preserved - protection from the effects of passive interference outside the selection zone and the ability to measure the speed of objects within the selection zone.

As a result of the search for alternative solutions in this and related fields of application of radar with intra-pulse reception and signal processing for detecting and measuring the motion parameters of many objects among patent sources and literature, it has been established that the known devices do not have the above features. Therefore, the proposed technical solution is novel, since the authors are not aware of devices of a similar purpose containing features that appear in the proposed invention as distinguishing features.

An analysis of the results of a patent search showed that the proposed solution does not follow explicitly from the prior art. From the prior art defined above, the influence of the essential features of the proposed technical solution on the emergence of new properties of the object and on the achievement of the above technical result has not been revealed. Therefore, the claimed technical solution meets the condition of patentability "inventive step".

The invention is aimed at expanding the functionality and improving the technical characteristics of the radar used, as noted above, in security systems in transport, security of objects, control of process parameters, etc. It follows that the claimed invention meets the criterion of "industrial applicability".

The essence of the invention is illustrated by drawings.

In FIG. 1 shows a block diagram of the RLD, which contains an antenna (A), a Doppler transceiver module (DPTM) with quadrature outputs I(t) and Q(t) of the converted signals, a synchronization and pulse shaping unit (BSFI), the first and second analog-to-digital converters (ADC-1 and ADC-2), signal processing unit (BOS), output data bus (SHVD), data exchange bus (SHOD) with a personal computer (PC). The BOS is based on a programmable logic integrated circuit (FPGA) and a digital signal processor (DSP), while the FPGA contains a read-only memory (ROM), an arithmetic logic unit (ALU) and a random access memory (RAM), and the DSP contains a read-only memory device (DSP ROM), DSP computing core and random access memory (DSP RAM), as well as the first and second transceivers of the serial port buses PP-1 and PP-2.

In FIG. Figure 2 shows the homodyne (a) and autodyne (b) versions of the DPSM with quadrature outputs I(t) and Q(t) of the converted signals, where A is the antenna; AG - oscillator; DM-1, DM-2 - the first and second power dividers; CM-I and CM-Q - I(t) and Q(t) quadrature channel mixers, respectively; FV - quadrature phase shifter; C - circulator; d _in - phase distance between the mixers.

In FIG. 3 shows one of the options for the implementation of the BSFI, containing a reference clock generator (OTG), a frequency multiplier (FC), a frequency divider counter (FFS) and a pulse selector (SI).

In FIG. Figure 4 shows the time diagrams of the processes of generating and processing signals in the RLD for detecting targets and measuring their motion parameters in the selection zone: (a) - output voltage u _yn (t) of the BSFI for controlling the DPDM transmitter; (b) - voltage of the probing u _probe (t) radio signal; (c) - voltage u _neg.1 (t) of the radio signal received (reflected) from the first target; (d) - voltage u _neg.2 (t) of the radio signal received (reflected) from the second target; (e) - output voltage u _cm (t) at the outputs of the quadrature mixer; (e) - voltage of clock pulses u _ti (t) at the third output of the BSFI; (g) - voltage u _in-z (t) signals after the operation of sampling-storage (storage); (h) - voltage u _vych (t) of the signals after the operation of subtraction from the current values of the previous samples.

RLD (see Fig. 1) contains an antenna (A) connected to a Doppler transceiver module (DPTM) with quadrature outputs I(t) and Q(t) of the converted signals, a synchronization and pulse shaping unit (BSFI) connected by its first output to the control input of the DPPM, the first and second analog-to-digital converters (ADC-1 and ADC-2) and the signal processing unit (BOS). In this case, the signal outputs I(t) and Q(t) of the DPMM are respectively connected to the signal inputs of the first ADC-1 and the second ADC-2 analog-to-digital converters, and the outputs of the latter are connected to the signal inputs of the signal processing unit (BOS), while the second the output of the BSFI is connected to the clock input of the BOS, and the third one is connected to the clock inputs of ADC-1 and ADC-2. In addition, the BOS has an output data bus (SHD) for transmitting the results of signal processing to the end user (not shown in Fig. 1) and a data exchange bus (SOD) with a personal computer (PC).

Antenna A can have various versions, for example, in the form of a slot or strip phased array, horn, dielectric, lens antenna or antenna of the “leaky wave” type (see pages 21, 114, 142, 151, 200 in [31]) .

DPSM with outputs I(t) and Q(t) quadrature signals also has alternative technical solutions. It can be made according to a homodyne or autodyne scheme (see descriptions of analogues: Fig. 4 [8], Fig. 5a, 5b [25]; Fig. 3 [26], Fig. 1a [27], patent [32]); Fig. 5 in [33], Fig. 1 in [34]). DPPM may include a controller-programmed frequency synthesizer. By the way, ready-made DPPM modules are produced by the industry (see, for example, the description of the Doppler module: “24.125 GHz Ranging Sensor Head, Dual Channel, Short Range”, Model SSD-24303-20M-DW on the site https://www.eravant.com /).

In the homodyne version, the DPMM contains separate transmitter and receiver units connected to the antenna A through a decoupling device, for example, a circulator (see Fig. 1a [27]), while the receiver, made on the basis of a quadrature mixer, provides the formation of two signals I at its outputs (t) and Q(t) shifted in phase by 90° (see expressions (3.6) and (3.7), p. 106, [35]).

An example of performing DPPM according to the homodyne scheme, shown in Fig. 2,a, contains an oscillator AG, connected by its high-frequency output to the input of the first power divider DM-1, the first output of which is connected to the first input of the circulator C, and the second - to the input of the phase shifter FV, while the input of the second power divider is connected to the third input of the circulator C DM-2, and the second input of the circulator C is connected to antenna A. At the same time, the outputs of the power divider DM-2 are respectively connected to the signal inputs of the mixers CM-I and CM-Q, the heterodyne inputs of which are connected to the outputs of the PV phase shifter, while the transmitter control input and quadrature outputs I(t) and Q(t) of the mixers CM-I and CM-Q are connected to the same-name inputs and outputs of the DPMM. The composition of the DPPM may include additional elements that do not change the essence of the invention. For example, a low-noise amplifier can be installed in front of the second power divider DM-2, and a power amplifier can be installed between the first power divider DM-1 and the circulator C. Converted signal amplifiers can be used at the outputs of the CM-I and CM-Q mixers.

In the autodyne version of the DPSM, as noted above, the functions of the transmitter and receiver are performed by a single element - the self-oscillator. To obtain quadrature signals in an autodyne DPMM, a variant with two detector diodes in the segment of the transmission line between the oscillator and the antenna is known (see Fig. 9b, [36]). Another option is based on the use of the so-called "two-diode autodyne" (see Fig. 10, [36]), which is a system of two mutually synchronized partial auto-oscillators with strong coupling [37]. In this case, phase-shifted signals I(t) and Q(t) are recorded in the power circuit of these partial self-oscillators (see Fig. 14, 15, 22, articles [36]). To reduce the degree of signal distortion due to autodyne frequency changes, it is desirable to stabilize autodyne oscillators by means of external high-quality resonators [18].

An example of the implementation of the DPPM according to the autodyne scheme is shown in Fig. 2b. The DPMM contains an oscillator of the AG, connected by its high-frequency output to the input of the first CM-I and the second CM-Q "per pass" mixers connected in series, having a phase distance d _in between them, a multiple of an odd number 1/8 of the wavelength in the transmission line, and the output the second CM-Q mixer is designed to connect antenna A, and the transmitter control input and quadrature outputs I(t) and Q(t) of the CM-I and CM-Q mixers are connected to the same-named DPDM inputs and outputs. The structure of DPPM made according to the autodyne scheme may include additional elements that do not change the essence of the invention. For example, the outputs of the CM-I and CM-Q mixers can use converted signal amplifiers.

BSFI can be made on the basis of "hard" logic, FPGA or using specialized microcircuits and have a different functional construction. One of the options for the implementation of the BSFI (see Fig. 3) contains a reference clock generator (OTG) connected to the inputs of a frequency multiplier (FC) and a frequency divider counter (FFS), as well as a pulse selector (SI), and the FFS output is connected to to the first input of the SI and is the first output of the BSFI, and the output of the UCH is connected to the second input of the SI and is the second output of the BSFI, while the output of the SI is the third output of the BSFI. This block diagram of the FSFI is implemented on a Si5368 microcircuit, containing a reference clock generator and two independent programmable frequency multipliers / frequency dividers with a low jitter (jitter) phase of the output oscillations in the frequency range from 2 kHz to 1.4 GHz (see the Silicon Laboratories website: http://www.silabs.com). In this case, the logical element "AND" is used as the SI.

As ADC-1 and ADC-2 in the proposed RLD, it is preferable to use high-speed ADC microcircuits [38; 39]. For example, the AD9689 chip from Analog Devices is a dual 14-bit ADC with a JESD204B interface, with a speed of 2.6 GB / s (see website: https://www.analog.com/en/products/ad9689.html# product overview). This ADC is capable of direct sampling of analog signals with -3 dB bandwidth up to 9 GHz. Similar ADCs of the DAC38RF82 and DAC38RF89 types are manufactured by Texas Instruments.

BOS (see Fig. 1) can be implemented based on high-speed FPGA chips and/or digital signal processors (DSPs). Modern FPGAs, for example, from Agere Systems, provide transfer rates up to 10 Gb / s; Cypress Semiconductor - up to 12 Gb / s; bipolar FPGAs of the SH100K series from Infinion operate at a clock frequency of up to 5 GHz (see pages 132-133 [38]), and FPGAs of the UltraScale family have a speed of up to 32.75 Gb / s (see pages 149-189 [39] ). The speed of the DSP is somewhat lower than that of FPGA-based devices. However, new technical solutions based on radio photonic technology make it possible to extend the DSP frequency range up to 5 GHz [40]. From the results of the performed analysis of the technical condition of digital signal processing means, it follows that today the preferred embodiment of BOS 9 is to use the advantages of FPGAs in terms of speed in primary signal processing and the advantages of DSPs in terms of the ability to perform complex operations with signals during their final processing.

BOS (see Fig. 1) is based on FPGA and DSP. In this case, the FPGA contains: a read-only memory (ROM) that stores the program for the primary signal processing; arithmetic logic unit (ALU) that performs arithmetic operations (addition, subtraction, multiplication, etc.); random access memory (RAM) that stores current data while performing operations. The DSP contains: read only memory (ROM) storing the DSP signal processing program; high-speed computing core that performs the functions of digital signal processing (spectral analysis, digital signal filtering and data generation, indications); random access memory (RAM) that performs the functions of storing current values and results of processing DSP signals; transceivers PP-1 and PP-2 of serial port buses for transmission and exchange of information via ShVD and ShOD.

RLD, made in accordance with the proposed method for detecting and measuring the motion parameters of location objects in the selection area, works as follows.

After applying the supply voltage to the RLD in the BOS (see Fig. 1), in accordance with the algorithm of its operation, the computing core of the DSP includes the command "Initialization" [41], which is used to configure the peripheral devices of the DSP, allocate internal memory, set the values of internal variables, copying the executable code from the low-performance ROM to the high-performance RAM of the DSP and sending the command to the ALU "Fetch from the ADC and store the results in memory", according to which the FPGA switches to the ready mode for receiving digitized signals from ADC-1 and ADC-2, followed by the formation of a data array in FPGA RAM memory.

In the BSFI (see Fig. 3), after the supply voltage is applied, the OTG reference clock generator starts. The output signal of this generator is fed to the PA frequency multiplier, at the output of which clock pulses are generated with a repetition period T _ti , as well as to the counter frequency divider of the FFS, at the output of which pulses of duration t _and with a repetition period T _p are formed to control the DPSM transmitter (see Fig. Fig 4a). These pulses, which also arrive at the first input of the SI pulse selector (see Fig. 3), form bursts of pulses from 0 to M at its output from the sequence of pulses from the MF output (see diagram (e) for the case of M = 10 in Fig. 4 ) filling the time interval t _and , which are further fed to the clock inputs of ADC-1 and ADC-2. In this case, the number of clock pulses during the time t determines _the range resolution of the RLD: Δr=t _and c/2M, where M is the number of readings during the time t _and .

The trigger pulses of the DPPM transmitter from the first output of the BSFI u _im (t), (see Fig. 4,a) are fed to the control input of the DPPM, providing conditions for excitation of microwave oscillations in it. The probing radio pulses u _probe (t) generated by the DPPM transmitter (see Fig. 4b), having passed through the first power divider DM-1 and the circulator C, are radiated through the antenna A into the controlled space. The expression for the probing signal u _probe (t) with a rectangular envelope U _probe (k,t) has the form [11]:

where

A ₀ - amplitude of the probing signal;

- единичная функция зондирующего сигнала;

- unit function of the probing signal;

ω ₀ =2πƒ ₀ - circular frequency of radiation;

t _and and T _p =2π/Ω _p - duration and repetition period of pulses;

ϕ _k - random initial phase of the k-th radio pulse;

k is an integer, the serial number of the probing radio pulse.

Part of the transmitter signal power from the second output of the first power divider DM-1, passing through the quadrature phase shifter FV, is divided equally and fed to the CM-I and CM-Q mixers as heterodyne signals with a relative phase shift of π/2. These signals will be called "direct" (index "prm"), in general they are written as

where

A _prm - amplitude of direct signals;

- единичная функция прямого сигнала.

- unit function of the direct signal.

If there are N point objects of location in the radiation field of antenna A, the radio signals reflected from them are received by antenna A, pass through the circulator C, are equally divided by the second power divider DM-2 and arrive at the signal inputs of the first and second mixers CM-I and CM-Q, respectively ( see Fig. 2a). We write the expression for these radio signals in the form:

where

A ₀ - amplitude of the probing radio signal;

- безразмерный коэффициент ослабления амплитуды излучаемого сигнала по пути распространения до цели и обратно, приведенный к входу-выходу антенны А;

- dimensionless coefficient of attenuation of the amplitude of the emitted signal along the propagation path to the target and back, reduced to the input-output of antenna A;

τ _n =2R _n /s; - delay time of the reflected radiation from the n-th target;

R _n - current distance to the n-th target;

c is the speed of propagation of radio waves;

ϕ _k - random initial phase of the k-th radio pulse;

k is an integer;

ϕ _n - constant phase shift associated with the reflective properties of the n-th target;

R _probe - the average power of the probing radio signal;

P _min - minimum detectable (threshold) signal;

G _A - antenna gain A;

λ=s/ƒ ₀ - radiation wavelength;

σ _n - effective scattering area of the n-th target;

c is the propagation speed of electromagnetic radiation;

- единичная функция отраженного сигнала.

is the unit function of the reflected signal.

In FIG. 4, c and d show the timing diagrams of the signals for the case of the presence in the radiation field of antenna A of two point objects of the location (N=2), located respectively at distances R ₁ and R ₂ , when the delay times are τ ₁ =2R ₁ /s and τ ₂ \u003d 2R ₂ / s.

As a result of the non-linear interaction of direct and reflected vibrations in the CM-I and CM-Q mixers, the signals are converted to the low-frequency region. At the same time, at the outputs I(t) and Q(t) of the mixers, the converted signals u _pr (t) are formed in the form of video pulses. The expressions obtained for the kth probing radio pulse have the form:

where

And _with =k _CR And ₀ A _PM /2 - amplitude factor of the output signals of the mixers CM-I and CM-Q;

- единичная функция преобразованного сигнала на выходах смесителей CM-I и CM-Q;

- unit function of the converted signal at the outputs of mixers CM-I and CM-Q;

R _n (t) - function of the movement of the n-th target in the area of selection RLD;

- синфазная и ортогональная составляющие собственных шумов смесителей и шумов СВЧ генератора, преобразованных на выходы смесителей CM-I и CM-Q;

- in-phase and orthogonal components of the intrinsic noise of the mixers and the noise of the microwave generator, converted to the outputs of the mixers CM-I and CM-Q;

A _them =k _d A ₀ A _PRM /2 - the amplitude of the pulse received at the outputs of the mixers CM-I and CM-Q as a result of detecting a direct signal;

k _CR and k _d - respectively, the coefficients of conversion and amplitude detection mixers CM-I and CM-Q voltage (assuming that these mixers have identical parameters);

λ=2πс/ω ₀ - wavelength of microwave radiation.

Note that the initial phase _ϕk in (5), (6) is absent, since, under the condition τ<t _{, both} direct and reflected radiation within the _kth radio pulse are coherent and the phase ϕk in the converted signal is subtracted.

The first terms in (5) and (6), representing the result of the reflected signal conversion in the CM-I and CM-Q mixers, contain information about the distance to the location objects and their speed. At the same time, for real-life speeds of movement of location objects, the condition is true that during the time t _and the action of a probing radio pulse, the distance between the antenna A and the location objects will practically not change. Then, according to (5) and (6), the received video pulses at the outputs I(t) and Q(t) of the quadrature mixer remain practically constant during the action of these radio pulses. Therefore, they look in the form of step functions of time, while the “height” of the steps is proportional to the level of the reflected signal, and their sign (up or down) depends on the current phase difference of the DPDM emitted by the transmitter and the oscillations reflected from the location object (see oscillograms in Fig. 17 article [18], the timing diagrams of these signals for the case of two point targets without taking into account the noise in Fig. 4e of this description). With the relative movement of the location object, instantaneous changes in the height of the steps occur with the Doppler frequency [18]. Thus, signals received from location objects that are located at shorter distances than the next location object cause changes in the height of the video pulses I(t) and Q(t) at the output of the quadrature mixer from the moment of arrival of the reflected radiation τ ₁ to the end of the radio pulse t _and . In this case, the formation of changes due to the action of the reflected radiation from the next location object occurs from the time τ ₂ also until the end of the radio pulse t _and . It follows that in the device under consideration, additive summation of radio pulses of signals occurs.

и

в (5) и (6) отображают результат преобразования собственных шумов смесителей CM-I и CM-Q и шумов СВЧ генератора. Присутствие этих шумов выражается в шумовой модуляции высоты видеоимпульсов I(t) и Q(t) на выходах смесителей CM-I и CM-Q. При этом необходимо отметить, что шумовые составляющие

и

на выходах смесителей CM-I и CM-Q представляют собой независимые стационарные нормальные процессы с нулевым средним значением. Взаимная корреляция этих составляющих отсутствует.Second terms

and

in (5) and (6) represent the result of the conversion of the intrinsic noise of the mixers CM-I and CM-Q and the noise of the microwave generator. The presence of these noises is expressed in the noise modulation of the height of video pulses I(t) and Q(t) at the outputs of mixers CM-I and CM-Q. It should be noted that the noise components

and

at the outlets of the CM-I and CM-Q mixers are independent stationary normal processes with zero mean. There is no mutual correlation of these components.

The third terms in (5) and (6) are due to the detection of the direct signal of the microwave generator acting on the inputs of the mixers CM-I and CM-Q. Therefore, the video pulses formed at the outputs of the CM-I and CM-Q mixers, obtained as a result of receiving the reflected radio signals, are located on some “parasitic” pedestal.

In the implementation of the RRD with autodyne DPDM (see Fig. 2, b) with the arrival of the trigger pulses of the transmitter, the excitation of microwave oscillations in the self-oscillator of the AG occurs. The probing radio pulses u _probe (t) formed in the AG (see Fig. 4,b), passing through the CM-I and CM-Q feed-through mixers, spaced in the transmission line by an odd number of 1/8 of the wavelength, are radiated through the antenna A into the controlled space. The expression for the probing signal u _probe (t) with a rectangular envelope U _probe (k,t) is generally written in (1).

If there are N point objects of location in the radiation field of antenna A, the radio signals reflected from them are received by antenna A and sent by the transmission line through the feed-through mixers CM-Q and CM-I to the self-oscillator AG (see Fig. 2b). In this line, direct (probing) and reflected radio signals cause wave interference. The phase relationships of these waves are such that in the CM-I and CM-Q mixers, the signals are converted to the low-frequency region and the formation of quadrature signals I(t) and Q(t), respectively, is observed at their outputs [33]. Expressions describing the process of formation of these signals have the form (5) and (6).

From the outputs of the mixers CM-I and CM-Q (see Fig. 1), the video pulses are then fed, respectively, to the signal inputs of the ADC-1 and ADC-2, where the operation of sampling signals (5) and (6) in time is performed first. During the action of clock pulses u _ti (t) (see diagram (e) in Fig. 4) in ADC-1 and ADC-2, the instantaneous values of signals (5) and (6) are sampled and stored in the form of pulses, the amplitude of which equal to the instantaneous values of these signals (see diagram (g) in Fig. 4). The levels of these pulses are further converted in ADC-1 and ADC-2 into digital values, which are sent to the FPGA RAM in the form of a parallel code as an array of data obtained for the received signal from the kth probing radio pulse:

where

- цифровые отсчеты мгновенных значений принятого сигнала от k-го зондирующего радиоимпульса, полученные для m-го тактового импульса (здесь m=0, 1, 2, …М).

- digital readings of the instantaneous values of the received signal from the k-th probing radio pulse obtained for the m-th clock pulse (here m=0, 1, 2, ... M).

и

соответственно, полученных в моменты времени t=mT_ти, поступают далее в АЛУ ПЛИС, где в соответствие с программой, заложенной в ПЗУ ПЛИС, выполняется операция вычитания, при которой из текущих значений отсчетов, начиная с m=1, вычитаются значения предыдущих отсчетов:Sequences of instantaneous values digitized for each k-th video pulse

and

respectively, received at time points t=mT _ti , are further sent to the FPGA ALU, where, in accordance with the program embedded in the FPGA ROM, a subtraction operation is performed, in which the values of the previous samples are subtracted from the current values of the samples, starting from m=1:

The data array x _km and y _km obtained as a result of subtracting (9) and (10) enters the FPGA RAM:

where

x _km , y _km - digital readings of the instantaneous values of the differences from (9) and (10) for the k-th probing radio pulse obtained for the m-th clock pulse (here m=1, 2, ...M).

и

на выходах смесителей CM-I и CM-Q в результате дискретизации и оцифровки мгновенных значений отсчетов вследствие эргодичности процессов в среднем по реализациям k и отсчетам m сохраняют свое среднеквадратическое значение уровня шума σ_ш. Значение уровня шума σ_ш может быть рассчитано или измерено экспериментально и учтено в программе работы БОС.As a result of the subtraction operation in the obtained values x _km , and y _km , the useful signal is separated from the parasitic pedestal, due to the detection of the direct signal of the microwave generator by the CM-I and CM-Q mixers, as well as the separation of the additive superposition of the signals of the previous elements (m-1) resolving targets for subsequent elements (m). In the diagram (h) of Fig. 4, using the example of the presence of two targets in the selection zone, the formation of difference signals that are located in different cells of the FPGA RAM is shown. In this case, the noise components

and

at the outputs of mixers CM-I and CM-Q, as a result of discretization and digitization of instantaneous values of samples, due to the ergodicity of processes, on average over realizations k and samples m, they retain their root-mean-square value of the noise level _σsh . The value of the noise level _σsh can be calculated or measured experimentally and taken into account in the BOS work program.

After that, in the ALU of the FPGA, the values of the relative amplitude A _km and the phase Ф _km of the signal are calculated and entered into the memory of the DSP RAM, calculated by the formulas:

и

обнаруженных целей поступает в ОЗУ ПЛИС:Further, with the obtained values of the amplitude A _km for each sample of the k-th probing radio pulse in the FPGA ALU, they are normalized with respect to the root-mean-square value of the noise level σ _w : q _km =A _km /σ _w , which is equivalent to determining the signal-to-noise ratio in each resolution element by range. According to the program embedded in the ROM of the DSP, the computational core of the DSP performs enumeration and analysis of the values q _km . During these operations, the presence (detection) of targets in the RLD selection zone is determined by the excess of the pre-selected threshold value of the signal-to-noise ratio q _then . The data array obtained as a result of the analysis of the signal-to-noise values q _km

and

detected targets enters the FPGA RAM:

where

- цифровые отсчеты значений относительной амплитуды и фазы из (13) для k-го зондирующего радиоимпульса и m-го тактового импульса (здесь m=1, 2, …М), полученные для n-ой обнаруженной цели в зоне селекции (здесь n=1, 2, …N - порядковый номер цели, считая от РЛД, причем N≤M).

- digital readings of the values of the relative amplitude and phase from (13) for the k-th probing radio pulse and the m-th clock pulse (here m=1, 2, ... M), obtained for the n-th detected target in the selection zone (here n= 1, 2, …N - serial number of the target, counting from the radar, and N≤M).

до n-ой цели в зоне селекции по формуле:For the obtained values of m, at which location objects are detected, the computational core of the DSP BOS calculates the current distance

to the n-th target in the selection zone according to the formula:

where m is the sequence number of the sample corresponding to the n-th goal in the selection zone; t _and - the duration of the radio pulse; c is the propagation velocity of microwave radiation; M is the number of readings during the time t _and .

согласно (16) для каждой реализации k через шинный приемопередатчик ПП-1 ЦСП передаются по шине ШВД конечному потребителю:Calculation results

according to (16) for each implementation k through the bus transceiver PP-1, the DSPs are transmitted over the bus SHVD to the end user:

и фаз

сигналов n-й цели последовательно применяются операции «сшивания» и «сглаживания». Первая операция обеспечивает непрерывность вращения вектора

(без скачков) на комплексной плоскости при изменении k в процессе перемещения целей (здесь j - мнимая единица) [42]. Для этого сначала разрешается неопределенность значений фаз

полученных вычислением арктангенса в (13), ограниченного диапазоном однозначного определения фазовых углов ±π/2. Цель этой операции достигается учетом в значениях

перехода фазы из одной полуплоскости в другую на краях границ однозначности. Для этого, продифференцировав полную фазу

получим выражение для мгновенной частоты

доплеровского сигнала n-й цели:In the signal processing unit (BOS) by the DSP computing core to the set of amplitude values stored in the DSP RAM

and phases

signals of the n-th target are sequentially applied the operations of "stitching" and "smoothing". The first operation ensures the continuity of the rotation of the vector

(without jumps) on the complex plane when changing k in the process of moving targets (here j is an imaginary unit) [42]. To do this, the uncertainty of the phase values is first resolved

obtained by calculating the arc tangent in (13), limited by the range of unambiguous determination of the phase angles ±π/2. The purpose of this operation is achieved by taking into account in the values

phase transition from one half-plane to another at the edges of the uniqueness boundaries. To do this, differentiating the total phase

we obtain an expression for the instantaneous frequency

Doppler signal of the n-th target:

и

- производные по времени от х_km и y_km в каналах I(t), Q(t), соответственно, значения которых определяются методом конечных разностей, беря соседние значения переменных, например, при k и k±1.where

and

- time derivatives of x _km and y _km in channels I(t), Q(t), respectively, whose values are determined by the finite difference method, taking neighboring values of variables, for example, at k and k±1.

вычислительным ядром ЦСП выполняется вычисление относительной скорости движения n-ой цели по следующей формуле:According to the results of calculation of the Doppler frequency obtained according to (18)

the computational core of the DSP calculates the relative speed of the n-th target according to the following formula:

where c is the propagation speed of electromagnetic radiation;

ω ₀ =2πƒ ₀ - circular frequency of radiation.

n-ой цели согласно (19) вычислительным ядром ЦСП последовательно при изменении k применяется операция «сглаживания», которая выступает в качестве фильтра нижних частот «зашумленных» данных. Эта операция выполняется вычислительным ядром ЦСП БОС, например, применением операции «скользящее среднее» или алгоритма фильтрации Калмана [43]. «Сглаженные» данные текущей скорости

нового массива через шинный приемопередатчик ПП-1 ЦСП передаются по шине ШВД конечному потребителю.To the results of calculating the speed

For the nth target, according to (19), the DSP computational core successively applies the “smoothing” operation as k changes, which acts as a low-pass filter for “noisy” data. This operation is performed by the computational core of the DSP BFB, for example, by applying the “moving average” operation or the Kalman filtering algorithm [43]. "Smoothed" current speed data

of the new array through the bus transceiver PP-1 DSP are transmitted over the bus SHVD to the end user.

от времени определяется вычислительным ядром ЦСП через решение дифференциального уравнения

численными методами, например, как отмечалось выше, методом конечных разностей, беря соседние значения переменных при k и k±1. При этом предыдущее решение дифференциального уравнения для каждой n-ой цели является постоянной интегрирования последующего решения. Последовательное решение этого уравнения приводит к «сшиванию» отдельных фаз

и получению закономерности изменения полной фазы Ф⁽ⁿ⁾(t) во времени, что позволяет определить искомую функцию, которая описывает закон движения n-ой цели:Total phase dependence

from time is determined by the computational core of the DSP through the solution of the differential equation

numerical methods, for example, as noted above, by the method of finite differences, taking neighboring values of variables at k and k±1. In this case, the previous solution of the differential equation for each n-th goal is the constant of integration of the subsequent solution. Sequential solution of this equation leads to "crosslinking" of individual phases

and obtaining the pattern of change in the total phase Ф ⁽ⁿ⁾ (t) in time, which allows you to determine the desired function that describes the law of motion of the n-th target:

The results of calculating the function of the law of motion R ⁽ⁿ⁾ (t) of the n-th target to reduce the influence of noise and discretization in the process of implementing a sequence of probing radio pulses by the computational core of the DSP BOS are smoothed, for example, by using the “moving average” operation or the Kalman filtering algorithm.

о перемещении n-ой цели после сглаживания через шинный приемопередатчик ПП-1 ЦСП с заданным темпом передаются по шине ШВД конечному потребителю. Последующий анализ полученной функции

позволяет определить величину перемещения цели, амплитуду, частоту и характер вибраций поверхности объекта локации, а также его иные параметры и характеристики.Current data

about the movement of the n-th target after smoothing through the bus transceiver PP-1 DSP with a given rate are transmitted over the bus SVD to the end user. Subsequent analysis of the resulting function

allows you to determine the amount of target movement, the amplitude, frequency and nature of vibrations of the surface of the location object, as well as its other parameters and characteristics.

The proposed method was implemented in the form of an operating layout of the 8-mm range radar, made according to an autodyne scheme based on a microwave generator based on a Gunn diode and a quadrature mixer on diodes with a Schottky barrier. With the duration of the probing radio _pulses t _and =100 ns, the distance to the border of the selection zone was 15 m. In this case, the range resolution of the RLD was _Δr =0.15 m. The RLD potential in the Doppler frequency band of 0.05…20 kHz was found to be equal in the range of 95…100 dB. Simulation tests carried out in an open area with corner reflectors confirmed the possibility of detecting and determining the motion parameters of several targets located in the selection zone. The results of laboratory studies of the RLD sample confirmed the possibility of restoring the complex law of target movement according to the proposed method for processing radar signals.

Thus, the proposed method for processing radar signals and a device that implements it while maintaining the functionality of the prototype (protection from the effects of passive interference and the ability to measure the speed of the target in the selection zone) provides the ability to detect multiple targets, determine the distance to them and the speed of movement, as well as parameters vibrations and complex laws of movement of the location object.

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Claims (21)

1. A method for processing radar signals for detecting targets and measuring their motion parameters in the selection zone, consisting in the fact that the controlled area of space with the targets located in it is periodically irradiated with probing radio pulses, during the emission of probing radio pulses, radio pulses reflected from the targets are received, and they are separated into two quadrature channels, they are mixed with probing radio pulses, the parts of these radio pulses overlapping in time are converted into the low-frequency region in the form of two quadrature video pulses I(t) and Q(t), characterized in that the converted quadrature video pulses obtained in these channels are sampled by time and memorize at a set of time points t _km quadrature values

and

where k=0, 1, 2, ... - serial number of the probing radio pulse; m=0, 1, 2, …M, then in both channels from the current readings

and

starting from m=1, subtract the previous readings

and

:

and according to the obtained values of the differences x _km and y _km , the amplitude A _km and the phase Ф _km of the signal are calculated for each sample, starting from m=1, according to the following formulas:

further, for the obtained data of amplitude readings A _km , the values of the signal-to-noise ratio q _km =A _km /σ _w are calculated, where σ _w is the root-mean-square value of the intrinsic noise level at the receiver output, those values of sequence numbers are selected from the set of values q _km of the signal-to-noise ratio m, amplitudes A _km and phases Ф _km at which the signal-to-noise level exceeds the threshold: q _{km ≥q} thr , where q _thr is the threshold signal-to-noise ratio, distances are calculated

to the range resolution elements in which targets are detected, according to the formula:

where n is the serial number of the detected target; m is the number of the count to the n-th goal, for which the excess of the threshold level q _then was detected; t _and - the duration of the radio pulse; c is the propagation velocity of microwave radiation; M is the number of readings during the time t _and , and in relation to the samples of the received readings of phase

signal, successively when changing the number k of the probing radio pulse, differentiation is performed, while obtaining the values of the instantaneous frequency

Doppler signal of the n-th target:

where

and

- time derivatives of x _km and y _km in the channels I(t), Q(t), respectively, the values of which are determined by the finite difference method, for this, the neighboring values of the variables are taken, for example, at k and k±1, according to the results of the calculation doppler frequency

calculate the relative velocities of the targets according to the following formula:

where c is the propagation velocity of microwave radiation; ω ₀ \u003d 2πƒ ₀ - circular frequency of microwave radiation, the results of calculating the speed

The n-th target is successively smoothed when changing the number of the probing radio pulse k, then by the numerical method, for example, finite differences for each n-th target, the differential equation is solved

at the same time, the previous solution of this equation, starting from zero, is taken as a constant of integration of the subsequent solution and thus, for each n-th goal, the regularity of the change in the total phase Ф ⁽ⁿ⁾ (t) in time is obtained, then the desired function of the law of motion n- th goal according to the formula:

where λ is the wavelength of microwave radiation, the results of calculating the function of the law of motion R ⁽ⁿ⁾ (t) of the n-th target are successively smoothed when the number of the probing radio pulse k is changed. 2. The method according to p. 1, characterized in that the results of calculating the speed

and the function of the law of motion R ⁽ⁿ⁾ (t) of the n-th target is successively smoothed by the change in the number of the probing radio pulse k by using the "moving average" operation. 3. The method according to claim 1, characterized in that the results of calculating the speed

and the function of the law of motion R ⁽ⁿ⁾ (t) of the n-th target is successively smoothed with a change in the number of the probing radio pulse k by applying the Kalman filtering algorithm. 4. Radar sensor for detecting targets and measuring their movement parameters in the selection area, implementing the method according to paragraphs. 1, 2 and 3, contains an antenna, a Doppler transceiver module with quadrature outputs I(t) and Q(t) of the converted signals, a signal processing unit and a synchronization and pulse shaping unit, the antenna being connected to the high-frequency port of said Doppler transceiver module, to the control input of which the first output of the synchronization and pulse shaping unit is connected, characterized in that it additionally contains the first and second analog-to-digital converters, the signal inputs of which are connected to the quadrature outputs I(t) and Q(t) of the Doppler receiver of the transmitting module, and the outputs - to the signal inputs of the signal processing unit, while the second output of the synchronization and pulse shaping unit is connected to the clock input of the signal processing unit, and the third output is connected to the clock inputs of the first and second analog-to-digital converters. 5. A radar sensor for detecting targets and measuring their movement parameters in the selection zone according to clause 4, characterized in that the synchronization and pulse shaping unit contains a reference clock generator connected to the inputs of a frequency multiplier and a frequency divider counter, while the output of the counter is The frequency divider is the first output of the synchronization and pulse shaping unit, and the output of the frequency multiplier is its second output, while the first and second outputs are also connected to the inputs of the pulse selector, the output of which is the third output of the synchronization and pulse shaping unit. 6. A radar sensor for detecting targets and measuring their movement parameters in the selection zone according to clause 4, characterized in that the Doppler transceiver module with quadrature outputs I(t) and Q(t) of the converted signals is made according to the homodyne scheme and contains transmitter, first and second mixers, first and second power dividers, circulator and phase shifter with quadrature outputs of high-frequency signals, while the transmitter is connected by its high-frequency output to the input of the first power divider, the first output of which is connected to the first input of the circulator, and the second to the input of the phase shifter , while the input of the second power divider is connected to the third input of the circulator, and the second input of the circulator is a high-frequency port of the Doppler transceiver module, while the outputs of the second power divider are respectively connected to the signal inputs of the first and second mixers, the heterodyne inputs of which are connected to the quadrature outputs of the phase shifter , while this m transmitter control input and quadrature outputs I(t) and Q(t) of the first and second mixers are connected to the same inputs and outputs of the Doppler transceiver module. 7. A radar sensor for detecting targets and measuring their movement parameters in the selection zone according to clause 4, characterized in that the Doppler transceiver module with quadrature outputs I(t) and Q(t) of the converted signals is made according to an autodyne scheme and contains A microwave generator connected to the first port of a “per pass” quadrature mixer made on the basis of a transmission line with the first and second mixers placed in it, located at a phase distance equal to an odd number of one-eighth of a wavelength, while the signal outputs of the first and second mixers and the second port of the quadrature mixer "per pass" is connected respectively to the outputs I(t), Q(t) and the high-frequency port of the Doppler transceiver module. 8. Radar sensor for detecting targets and measuring their movement parameters in the selection zone in accordance with paragraphs. 4 and 6, characterized in that the microwave generator is made with frequency stabilization using a programmable frequency synthesizer. 9. Radar sensor for detecting targets and measuring their movement parameters in the selection area in accordance with paragraphs. 4 and 7, characterized in that the microwave generator is made with frequency stabilization by an external high-Q resonator.

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