RU2592264C2 - Electrically driven module - Google Patents

Abstract

FIELD: electronic equipment.

SUBSTANCE: invention can be used for fans, pumps, etc. Electrically driven module (1) includes electric motor (2) with permanent magnets, inverter (3) feeding electric motor (2), step (4) of direct current supply to inverter, controller (8) containing a modulator (5), designed for actuation of inverter, controlled by first digital signal (Vs_act), which is amplitude of phase voltage applied to electric motor, and by second digital signal (freq_act) which electric current frequency of phase voltage. Drive module (1) includes analogue/digital stage (6) for calculating optimum value of angle (δ_opt) advance in phase voltage applied to electric motor relative to an opposite electromotive force, as linear function of peak value of phase current, and analogue/digital stage (12) for measurement of angle (ϕ_act) between voltage applied to motor and phase current. Controller (8) is programmed to determine with sampling at frequency of electric current angle (γ_act) between phase current and an opposite electromotive force as difference between said optimum value of angle (δ_opt) in phase and advance angle (ϕ_act), measured between voltage applied to motor and phase current.

EFFECT: improved operational characteristics.

7 cl, 8 dwg

Description

FIELD OF THE INVENTION

This invention relates to an electric drive module and, in particular, to a sinusoidal (in jargon - AC (alternating current)) electric drive module.

State of the art

Below, an example of a non-limiting nature is described an electric drive module containing a brushless three-phase electric motor with permanent magnets generating a counter-electromotive force (CEMF) with a sinusoidal shape for actuating electromagnetic valves and pumps.

The use of a solenoid valve and an electric pump requires minimal acoustic noise and a reduction in both energy consumption and costs.

These needs led to the development of brushless motors with sinusoidal CEMF (brushless AC motors) driven by inverters configured to apply sinusoidal currents in the windings, which led to the discontinuation of brushless DC motors driven in six stages in PWM (pulse width modulation).

The sinusoidal shape of the CEMF, and with it the relative phase current, provide minimal (almost zero) fluctuation in the active torque and, as a result, small mechanical vibrations, and, therefore, low acoustic radiation.

It is also possible to minimize current absorption to generate a certain torque on the shaft and thus maximize the electromechanical conversion efficiency by optimally driving brushless AC motors, which are usually driven by current-controlled inverters with an applied voltage.

Such a drive requires that the proximity switches are switched in such a way as to guarantee at every moment that the polar axis of the magnetic field of the rotor remains at 90 electrical degrees relative to the polar axis of the magnetic field generated by currents circulating in the stator windings, regardless of the torque moment and speed of rotation.

Expensive sensors, such as absolute encoders, position sensors or Hall effect sensors, are usually used to obtain continuous information about the rotor's angular position.

The output signals generated by the sensors are then conveniently processed to control the proximity switches of the inverter so as to maintain an angular shift of 90 electrical degrees between the magnetic fields of the rotor and stator.

The presence of position sensors makes operation relatively expensive and, as a result, various strategies have been developed for the drive, in which the sensors are not used (in the jargon - "sensorless") and which are specifically designed to reduce operating costs.

Among these strategies, there are strategies based on the orientation of the stator and rotor fields (in the jargon - FOC), which guarantee the aforementioned orthogonal field correlation when using technically complex and expensive integrated circuits (ICs) with high computing capabilities (in the jargon - DSP) that perform the calculation the rotor angular position in real time solely on the basis of electrical quantities (voltage at the motor terminals and currents circulating in the windings), the presence of which is provided corresponding of a given scheme.

If the dynamic characteristics of the driven machine are not too intense, and this is the case for electric fans and electric pumps, then it is possible to apply the optimal criterion obtained directly from the aforementioned basic principle described below (the polar axis of the magnetic field of the rotor is maintained at every moment at 90 electrical degrees relative to the polar axis of the magnetic field generated by currents circulating in the stator windings): the drive operates in such a way that CEMF and phase current are maintained in phase; Naturally, the aforementioned criterion is fulfilled at each point of the working field (torque, rotation speed, DC voltage).

“Sensorless” drives that use strategies for a drive based on the aforementioned criteria are based on reading electrical quantities (for example, voltage at the motor terminals, currents circulating in the motor windings) for the purpose;

- detecting the transition through zero CEMF and current;

- estimates of the relative phase between CEMF and current

- and the use of suitable methods for driving the proximity switches of the inverter, which supports the phase of the two above-mentioned values.

The first drawback of these strategies is the fact that, in order to detect the transition through zero of CEMF, that is, to read the sign of CEMF at the moment when the current transition in the windings becomes zero, this state must be maintained for a sufficiently long period of time so that provide the ability to read CEMF, which differs from the necessary sinusoidal current curve to obtain low acoustic noise.

A solution to this problem was proposed in patent EP 2195916 in the name of the same applicant. However, in the disclosed solution, an additional cost is introduced due to the use of a network of “analog” hardware for the phase impedance of the motor.

A second drawback of the aforementioned control strategies is the need to read the phase current. There are generally two approaches to this reading, and both are costly in accordance with the prior art.

The first approach uses at least one IC device containing a Hall effect sensor that is sensitive to the magnetic field generated by the phase current (solution with galvanic isolation). In the second approach, at least one IC device is used, containing an amplifier for processing the voltage in the “external earth” shunt through which the phase current flows (solution without galvanic isolation).

In the first case, the IC device should be located near one of the conductors through which the phase current passes, and should have a very low sensitivity to "spurious" magnetic fields.

In the second case, the common-mode input voltage, which the amplifier must receive without damaging itself, must be at least equal to the inverter supply voltage (Vbus).

Disclosure of invention

In such a context, the main technical objective of the present invention is to provide an electric drive module in which the aforementioned disadvantages are eliminated both in terms of performance and cost.

One object of the present invention is to provide a silent electric drive module with low power consumption.

Another objective of this invention is to create an electric drive module based on a simple control architecture and cost-competitive.

The technical task is essentially accomplished by means of an electric drive module containing the technical features described in independent claim 1.

Brief Description of the Drawings

Other features and advantages of the present invention are apparent from the following non-limiting description of a preferred non-limiting embodiment of an electric drive module shown in the accompanying drawings, in which:

- figure 1 shows a functional diagram of an electric drive module according to the invention;

- figure 2 shows an equivalent phase diagram of a brushless AC motor;

- figa shows a vector diagram relative to the circuit shown in figure 2;

- Fig.3b shows a second vector diagram relative to the circuit shown in Fig.2;

- figure 4 shows a vector diagram relative to the optimal functioning of the circuit shown in figure 2;

- figure 5 shows a partial functional diagram of an analog submodule forming part of the drive module shown in figure 1;

- figure 6 shows a diagram of the main signals in the analog submodule shown in figure 5;

- figure 7 shows a diagram of the signals related to the function of "catch the rotation of the rotor" of the drive module shown in figure 1.

The implementation of the invention

When referring to the accompanying drawings, in particular when referring to FIG. 1, reference numeral 1 denotes an electric drive module according to the invention.

It should be noted that the functional diagram shown in Fig. 1 does not depict components that, from the point of view of operation, can be considered as the existing prior art, that is, drives with MOSFETs and an interface for user control for analog or digital input signals from the outside, that is, for control devices that set the speed or frequency for the electric drive module 1.

The electric drive module 1 contains a permanent magnet electric motor 2, a three-phase bridge or an inverter 3 for supplying electric current to the Vbus voltage of the electric motor 2, a direct current stage 4 for supplying electric power to the inverter 3, and a controller 8 for driving the inverter 3.

The controller 8 is a very simple and inexpensive receiving / processing device and is equipped with a memory for storing data.

The drive module 1 also contains, as described in more detail below, a low-cost analog module 11 for measuring the peak phase current value Is, known as "pk_detect", and an analog module 18, known as "zc_E-I_detect", for detecting a zero-current transition the first phase, for example, the phase indicated by U, of the electric motor 2, both analog modules communicating with the controller 8.

In the shown embodiment, the inverter 3 has three branches U, V, W, each of which contains a pair of MOSFETs, respectively Q_high_U, Q_low_U, Q_high_V, Q_low_V and Q_high_W, Q_low_W, connected in accordance with the prior art.

As shown in the drawing, stage 4 is provided with a filter stage, for example, comprising an equalizing capacitor (Cbus) and an inductance (Lbus).

For example, without any limitation on the scope of legal protection of the invention, reference is further made to a brushless permanent magnet motor equipped with a bipolar isotropic rotor.

The three-phase stator winding contains three windings of the same type and with an equal number of turns, and these windings provide a phase shift of 120 ° and a star connection with an inaccessible neutral point of the star or a triangle connection.

Figure 2 shows, for example, a model of the phase diagram of the electric motor, and figure 3 shows a vector diagram of electrical quantities.

Each of the three windings is characterized by its resistance Rs, synchronous inductive resistance Ls and CEMF, which has a sinusoidal curve and arises due to the rotation of the rotor with permanent magnets; Is is a phase current that is also sinusoidal and passes through each of the three windings.

The vector Vs represents the applied voltage, and the vector Es represents the CEMF induced in each of the three stator windings.

For convenience, the moduli of the vectors Es, Vs, Is are identified as peak values of the quantities to which they relate.

CEMF Es is a function of the temperature T _{mag of the} magnets, since it is proportional to the residual induction Br of the used permanent magnets, which, in turn, depends on the temperature of the magnets, and if α _mag is the coefficient of variation of the temperature of the permanent magnets with residual induction, then the following relation applies:

_{E S (T mag) = E} S0 · [ 1 _{+ α mag · (T mag -T} 0mag)],

where E _S0 is CEMF at a reference temperature T _{0mag of} magnets.

The longitudinal axis d is oriented in the direction of flow of the rotor Φr, and the transverse axis q forms an angle of 90 ° with the longitudinal axis d.

Based on the law of electromagnetic induction (e = dϕ / dt), the induced CEMF Es in the stator winding is always directed along the transverse axis q, that is, it is 90 ° ahead in phase with respect to the rotor flux Φr.

The voltage Vs applied by the drive device to the stator windings is represented, as indicated above, by the vector Vs.

The electromagnetic power of the electric motor is calculated by the formula 3Eslscos (γ), where γ is the angle between Es and Is (γ is positive if Is is ahead of Es).

The efficiency of electric motor 2 is maximum when, under equal conditions, the power Is is minimal, cos (γ) = 1, and γ is zero. This condition means that CEMF Es and current Is are in phase, as shown in FIG. 4.

The electric drive module 1 is configured to control a device with low dynamic characteristics, and, in the shown example, without any limitation of the scope of legal protection of the invention, the device is formed by a fan 7.

The drive module contains a shunt Rsh with low inductance, connected, as shown in figure 1, with three branches of the inverter 3, and the shunt is made so that currents circulating on the inverter flow through it, as described in more detail below.

The controller 8 comprises a modulator 5 controlled by a first digital signal "Vs_act" representing the amplitude of the phase voltages applied to the electric motor, and by means of a second digital signal "freq_act" representing the frequency of the electric current of the phase voltages applied to the electric motor 2.

For example, a three-phase bridge 3 generates, by means of a sinusoidal triangular pulse-width modulation, essentially of a known type, three voltages with a variable frequency, phase shifted by 120 electrical degrees.

The modulator 5 generates, in a completely normal / standard way (for example, by a sinusoidal-triangular method), the drive signals of six proximity switches equipped with MOSFETs, an inverter 3 from the aforementioned first digital signal "Vs_act" and the second digital signal "freq_act".

In particular, the first digital signal "Vs_act" represents the amplitude of the phase voltages in a three-phase symmetrical circuit applied to the motor 2, and the second digital signal "freq_act" represents the frequency of the electric current of the voltages applied to the motor, and these signals are obtained, as described in more detail below , by timely correction of the reference value of the frequency "freq_set" of the electric current installed inside or outside the controller 8, and directly related to the rotation speed "N_set" using the well-known ratio freq_set = N_set * p / 120, where p is the number of poles of the motor 2.

In turn, "N_set" is calculated directly from the input signal from the outside, setting the required speed entered using the aforementioned interface for a user who is not shown.

Modulator 5 calculates the number of switching on of six MOSFETs Q_high_U, Q_low_U, Q_high_V, Q_low_V and Q_high_W, Q_low_W.

The amplitude of the fundamental harmonic of the motor supply voltages represented by "Vs_act" can be programmed independently of the drive frequency represented by "freq_act".

Prior to a detailed description of the control circuits for "Vs_act" and "freq_act", a description is given of the coefficients used, obtained both by measuring some physical quantities and by processing, based on the values of the characteristic electrical parameters of the motor 2, conveniently stored in the memory of the controller 8, configured to implementation of control circuits.

Regarding the vector diagram shown in FIG. 4, it should be noted that it is possible to obtain an approximate expression of the optimum lead angle δ _{opt of the} voltage advance Vs applied with respect to CEMF Es.

Based on the trigonometric data from the vector diagram shown in FIG. 4, the exact expression can be written as follows:

Wherein:

"electrical" efficiency under normal loading conditions, defined as Es / (Es + Rs · Is) exceeds 0.9,

an electric motor 2 is used to drive a load characterized by a resistive torque proportional to the current, and this moment is a nonlinear function of the speed of rotation.

, при этом электрический импульс задан формулой ω_el=ω_mecc·р/2, где ω_mecc - механический импульс, и p - количество полюсов электродвигателя 2.Since the component Rs · Is under various loading conditions does not exceed 10% of Es, we can write Is≅k ′ · ω _el , and therefore Rs · Is≅k · ω _el , where $$k\text{'}\text{'}=\frac{k}{Rs}$$

in this case, the electric pulse is given by the formula ω _el = ω _mecc · p / 2, where ω _mecc is the mechanical pulse, and p is the number of poles of the electric motor 2.

The correctness of this linear approximation for ω _{el is} widely tested in technical practice for high-efficiency electric motors made with the possibility of driving fans and pumps, as indicated in the preferred case in the example.

As is known, CEMF Es also depends, in addition to the temperature of permanent magnets, linearly on a mechanical impulse, which under steady state conditions exceeds the electric impulse by a factor of 2 / p.

This allows the entire denominator of the exact expression tgδ _opt to be also considered proportional to ω _el , which makes it possible to simplify the exact expression by introducing a convenient correction coefficient K _corr (more than 1, usually less than 1, 2, which is established experimentally), as shown below:

where K _E represents the constant CEMF, measured in V / (rpm), and p represents the number of poles.

It should be noted that the dependence on ω _el disappears in a simplified expression.

If the influence of the temperature of permanent magnets on CEMF Es is taken into account by the coefficient α _{mag of} variation of the temperature of permanent magnets with residual induction, then it is necessary to introduce a linear dependence of the constant K _E CEMF on the temperature of the magnets according to the expression:

K _E (T _mag ) = K _E0 · [1 + α _mag · (T _mag -T _0mag )]

where K _E0 is the CEMF constant at the reference temperature T _{0mag of} permanent magnets, and, as already mentioned, α _mag is the coefficient of variation of the temperature of the permanent magnets with residual induction (α _mag = -0.1% / K for magnets from Nd-Fe-B )

This component is especially important in the presence of wide variability of the ambient temperature (for example, from -40 ° C to 120 ° C for electric fans forming part of the radiator cooling systems for engines operating under thermal loading).

Since in practice the value of the angle δ _opt , as a rule, is less than 30 electrical degrees, it is possible to approximate the tangent of the angle by the angle itself and, thus, write:

In other words, if the voltage drop Rsls on the active resistance is negligible with respect to Es, and the tangent of the lead angle δ _opt can be approximated by the angle itself, then we can say that in practice the lead angle δ _opt depends linearly on the phase current Is and the constant temperature T _mag magnets as described in the above simplified ratio.

The controller 8 comprises a stage 9, known as "gain_lpk", for calculating the lead angle δ _{opt of the} voltage Vs applied to the motor 2 relative to CEMF Es, as a linear function of the peak value of the phase current Is.

According to a first embodiment of the invention, the drive module 1 comprises a temperature sensor 10 located near the permanent magnets and in communication with the gain_lpk stage 9 to provide the information stage 9 with respect to the temperature of the permanent magnets.

Stage 9 is configured to calculate the lead angle δ _opt as a linear function of the peak value of the phase current Is according to the following equation:

where, as indicated, “Ls” is the synchronous inductance of the motor expressed in Henry, “p” is the number of motor poles, “K _E0 ” is the CEMF constant at the reference temperature “T _0mag ” of permanent magnets, “α _mag ” is the coefficient of variation permanent magnet temperature with residual induction, "T _mag " is the permanent magnet temperature measured by the temperature sensor 10.

In other embodiments of the invention, δ _opt above formula for calculation may be used for α _mag = 0, thus eliminating the dependence of expression on the temperature.

In order to ensure that the aforementioned signal is proportional to the current Is at the input of stage 9 "gain_lpk", the drive module 1 contains the aforementioned module 11 of an inexpensive type and a fully analog type, designed to measure the peak value of the phase current Is, this module being known as "pk_detect".

The analog module 11 receives, outside the controller 8, a voltage signal present on the Rsh shunt as an input signal, and returns, as an output signal, an analog signal whose level is directly proportional to the amplitude of the current passing through the Rsh shunt. The analog module 11, shown schematically, is described in EP 2195916, to which reference is made in its entirety in this document for completeness.

Stage 9 "gain_lpk" is located, as mentioned above, in the controller 8 and performs the following operations:

- analog-to-digital conversion of an analog signal proportional to the current Is at the output from the pk_detect stage with a time interval between two consecutive samples, preferably less than 1/100 of the minimum electric period;

- search for the maximum among the values read during each electrical period;

- the use of the maximum value found for the calculation of δ _opt , according to the formula defined above.

The pk_detect analog module 11 and the gain_lpk stage 9 form the first analog-to-digital stage 6 to calculate the optimum value δ _{opt of} the voltage advance angle Vs applied with respect to CEMF Es.

As shown in FIG. 3b, the angle between the applied voltage Vs and the phase current Is is known as ϕ and is positive when Vs is ahead of Is.

By indicating a positive δ when the voltage Vs is ahead of the CEMF Es and a positive γ when Is is ahead of the CEMF Es, the basic relationship connecting the three angles that can be obtained by considering fig.3b, can be written, that is:

γ = δ-ϕ.

The optimal control strategy implemented in the controller 8 provides independent control of the applied voltage Vs and the frequency of the electric current applied to the electric motor 2.

The optimal functioning of the drive module 1 is achieved, as described in detail below, by calculating only the angle "γ" selected at the frequency of the electric current.

More specifically, optimal efficiency is achieved at γ = 0 and with minimal current absorption.

In a first embodiment of the invention, the supply voltage is controlled independently of the stability control of the electric motor, that is, independently of the frequency.

The voltage control is performed so that the supply voltage changes so that "γ" is 0, as described in detail below.

In the calculation γ performed by the controller 8, the aforementioned relation between δ and ϕ is used, where δ is replaced by the optimal value of the aforementioned angle δ _opt calculated by the controller 8, and ϕ is measured in this way:

γ _act = δ _opt -ϕ _act .

In practice, in this solution, “γ” is not directly measured, since this would lead to expensive solutions, but it is obtained by “indirect” calculation of δ _opt and direct measurement of ϕ _act .

Thus, the drive module 1 contains a second analog / digital stage 12 for measuring the angle ϕ _act between the voltage Vs applied to the electric motor and the phase current Is.

To achieve an advantage, the controller 8 is programmed to determine, at a frequency of the electric current, the angle γ _act between the phase current Is and the anti-electromotive force Es as the difference between the optimal value of the phase advance angle δ _opt and the angle ϕ _act between the voltage Vs applied to the electric motor and phase current Is measured by measuring stage 12.

In a more detailed description, the controller 8 comprises a first subtractor 13 communicating with the first analog / digital stage 6 for receiving, as an input signal, the optimum value of the lead angle δ _opt , and with stage 12 for measuring the angle ϕ _act to calculate the difference between δ _opt and ϕ _act and obtaining γ _act .

The first task of the electric drive module 1 is to maintain the optimal efficiency of the electromechanical conversion.

As shown in FIG. 1, the controller 8 comprises an integrator 14 having an integration constant Ki made to integrate the angle “γ _act ” between the phase current Is and the anti-electromotive force Es and determine the first digital signal “Vs_act”.

In other words, the optimal control of the applied voltage is based on the calculated angle γ _act .

Operation of integration, performance integrator 14, a difference "δ _opt -φ _act" is such that the difference "δ _opt -φ _act" becomes equal to zero in steady state conditions.

In practice, the integration operation ensures that, under steady state conditions, "δ _opt = ϕ _act " and, as a result, "Vs_act" is set to a value consistent with the vector diagram shown in FIG. 4, with respect to the optimal functioning of the circuit shown in FIG. 2, that is, the minimum phase current to obtain a certain torque.

Preferably, the integration operation is performed in digital format by the controller 8.

In practice, control is performed by measuring peak phase 1 (from which δ _opt is derived) and measuring the angle ϕ _act ”.

The combination of "δ _opt " and "ϕ _act " gives "γ _act ", which after integration provides the supply voltage.

The second task is to stabilize the drive module 1.

The controller 8 is programmed to calculate the aforementioned second digital signal “freq_act” as the difference between the value of the reference frequency “freq_set” and the correction coefficient Δfreq proportional to the angle “γ _act ” between the phase current Is and the anti-electromotive force Es.

The freq_set frequency is set outside the controller 8 via the aforementioned control interface or inside the controller 8, as described in detail below.

The controller 8 includes a first calculation module 15, designed to calculate the frequency "freq_set".

The controller 8 comprises a second calculation module 16, which receives, as an input signal, the angle “γ _act ” between the phase current Is and the anti-electromotive force Es to apply the proportionality constant Kp to obtain a correction coefficient “Δfreq” proportional to the angle “γ _act ”.

The controller 8 contains a second subtracting node 17, communicating with the first calculation module 15 for receiving, as an input signal, the value "freq_set", and with the second calculation module 16 for receiving, as an input signal, a correction coefficient "Δfreq", and calculating the second digital signal "freq_act" as the difference between the value of "freq_set" and the value of the correction coefficient "Δfreq".

It is known from the literature (see, for example, A Sensoriess, Stable V / f Control Method for Permanent-Magnet Synchronous Motor Drives "- IEEE Transactions on Industry Applications, vol. 39, no. 3, May / June 2003) that sinusoidal drives modules based on the application of voltage and frequency to the drives of a permanent magnet synchronous electric motor demonstrate "system" instability of rotation speed, which can be shown by analyzing equations describing the dynamic characteristics of the electric motor - voltage equations and torque equations.

Instability manifests itself in fluctuations of the angle "δ" at a well-defined frequency, depending on the inertia of the load, K _{E of the} motor and its synchronous inductance Ls. At the same time, it was demonstrated that by modulating the frequency applied in proportion to the disturbances in the input power of the engine, the fluctuations in the angle "δ" are suppressed.

According to this invention, the oscillations of the angle "δ" are canceled, that is, the stabilizing effect is achieved by introducing a proportional correction "Δfreq" in the "γ _act " frequency "freq_set" according to the constant Kp of proportionality.

The correction is made through the second subtracting node 17 (see figure 1).

Since "γ _act " is obtained by simple sampling at an electric current frequency, unlike the prior art, a cheap 8-bit microcontroller can be used as a controller.

Since in the conditions of steady state, γ _act tends to zero, establishing optimal control of the applied voltage Vs as described above, the drive module 1 ensures that the speed set using the above external interface does not change to the correction factor Δfreq under stationary conditions (steady state).

Considering in more detail the analog / digital stage 12 for measuring the angle ϕ _act between the voltage Vs applied to the electric motor and the phase current Is, it should be noted that this stage contains an analog module 18, known as "zc_E-I_detect", for detecting zero crossing the current of the first phase of the motor 2, for example, the phase indicated by U, and a digital module 19, known as "fi_calc", inside the controller 8.

The analog module 18 generates, as an output signal, a third digital signal "zc_E-I_phaseU", the transition of which, performed as a "high-low", determines, in the first working configuration, described in more detail below, the transition through zero current in the first phase U.

As shown in particular in FIGS. 1 and 5, the drive module 1 comprises, inside the controller 8, preferably for the functions described in detail below, an enable module 20 of the analog module 18.

The resolution module 20 generates a high or low resolution signal “zce_run_on_fly".

There is the aforementioned first operational configuration of the analog module 18 when the enable signal "zce_run_on_fly" is high, as well as a second operational configuration of the analog module 18 when the enable signal "zce_run_on_fly" is low.

According to the shown embodiment, when the enable signal “zce_run_on_fly” is kept at “high”, in the first operating configuration, the analog module 18 is given permission to read the transition through zero phase current Is.

When the enable signal zce_run_on_fly is kept low, in the second operating configuration, the analog module 18 is given permission to read the transition through zero of the anti-electromotive force.

The first and second operating states are described in more detail below.

The controller 8 contains the first deviator software sw1 for controlling the enabling module 20.

The first deviator software sw1 is controlled by the fifth digital bridge_enabled signal generated by the controller 8.

The low logic level “zce_run_on_fly” is activated only if the controller 8 has verified that the modulator 5 does not control any of the six MOSFETs Q_high_U, Q_low_U, Q_high_V, Q_low_V and Q_high_W, Q_low_W of the three-phase bridge 3, and vice versa, when the resolution signal r_ne_on z is high.

As shown in figure 5, the analog module 18 contains a first input in communication with the drain electrode "d" of the low-voltage MOSFET Q_low_U, which is part of the supply branch of the inverter 3 of the first phase U of the motor 2, the second input in communication with the source electrode "s" a low voltage MOSFET Q_low_U, a third input communicating with a gate electrode g of a low voltage MOSFET Q_low_U, and a fourth enable input communicating with a resolution module 20 for receiving a zce_run_on_fly resolution signal.

The main input signals of the analog module "zc_E-I_detect" are:

- drain-source voltage of one of the low-voltage field-effect MOSFETs (Q_low_U in figure 1);

- gate voltage of the MOSFET Q_low_U ,;

- enable signal, known as "zce_run_on_fly".

As mentioned above, the analog module 18 is designed to detect when the first phase U of the motor 2 passes through zero current in accordance with the voltage drop between the first and second inputs when the resolution signal is high.

As shown in FIG. 5, the analog module 18 contains extremely cheap comparators.

In a preferred embodiment, shown by way of example, the analog module 18 comprises a first comparator "COMP_1", a second comparator "COMP_2" and a third comparator "COMP_3".

As mentioned above, the analog module 18 generates, as an output signal, a digital signal "zc_E-I_phaseU", the transition of which is performed as a "high-low", made to determine the transition (zc) through zero current in one of the phases of the motor ( in the functional diagram shown in FIG. 1, this is the current in phase U) in the first operating configuration.

The first and second comparators "СОМР_1" and "СОМР_2" have their own outputs, made as a "free collector" and short-circuited, and they provide an output in the logical AND format (AND). The first comparator "СОМР_1" reads the voltage drop in the MOSFET Q_low_U both during its operation, when the voltage drop reading is proportional to the current, and when it is modulated, when the voltage drop reading loses any correlation with the current circulating in phase U.

The total output of the first and second comparators "COMR_1", "COMR_2" is indicated by "ZC".

The second comparator "COMP_2", connected in the logical "AND" to "COMP_1", and driven by the gate signal of the MOSFET, eliminates, at the output of ZC, the indication of the undesired part of the voltage drop on the MOSFET Q_low_U, leaving for the third comparator "COMR_3" is only a function of a low-pass filter and a hysteresis comparator.

Thus, the third comparator "СОМР_3" determines the transition through zero current with a high signal-to-noise ratio.

A diagram of the signals (voltages and currents) obtained from simulations of circuits that describe the operation of the circuit is shown in Fig.6.

Curve "A" describes the zero-crossing curve, that is, the signal "zc_E-I_phaseU", in particular the zero-crossing in the current phase.

Curve "B" describes the curve of the aforementioned signal ZC at the common output "COMP_1" and "COMP_2".

Curve "C" describes the current flowing through the MOSFET Q_low_U, which is positive when the MOSFET is operating.

Curve "D" describes the voltage Vds on the MOSFET Q_low_U.

In practice, the analog module 18 performs synchronized reading on the gate electrode of the MOSFET Q_low_U.

Using the drain-source voltage on a low-voltage MOSFET Q_low_U provides, in the first operating configuration, a sign of the phase current U.

The read performed is a differential read of the "virtual ground", since it is performed on the Q_low_U low-voltage MOSFET.

It should be noted that, therefore, to determine the current transitions through zero, there is no need for Hall effect sensors or in-phase amplifiers.

As shown in FIG. 5, in the exemplary embodiment, the analog module 18 "zc_E-I_detect" contains, in series with the non-inverting input "COMP_1" connected to the drain Q_low_U, a resistor Rd and a protective circuit 21 formed by two diodes, connected in antiparallel.

The inverting input "COMP_1" has a resistor Rs and a protective circuit 22 in series connection, also formed by two diodes connected in antiparallel.

Module 18 contains a resistor Rg in series with a non-inverting input of the second comparator "COMP_2" and a load boost resistor.

At the common output "COMP_1" and "COMP_2", a load-boosting resistor R6 is also provided.

The comparator "COMR_3" is set, according to the prior art, not described in this document, for the formation, as mentioned above, of a hysteresis comparator.

The aforementioned digital module 19 comprises a timer 26 and communicates with the output of the analog module 18 to obtain, as an input signal, a third digital signal "zc_E-I_phaseU".

The digital module 19 communicates with the aforementioned modulator 5 for receiving, as an input signal, the fourth digital signal "zc_V_phaseU", the transition of which, performed as a "high-low", determines the transition through zero of the voltage applied to the first phase U of the motor 2.

The digital module 19 measures, by means of a timer 26, the period "T _ϕ " of the time between the transition through zero of the voltage applied to the first phase and the transition through zero of the current in the first phase, from the signal "zc_E-I_phaseU", and obtains the angle ϕ _act between the voltage Vs applied to the motor 2, and the phase current Is from the measurement of the period "T _ϕ " time based on the second digital signal, according to the formula: ϕ _act = 2 · π · freq_act · T _ϕ .

If in the previous formula “freq_act” is expressed in Hz and “T _ϕ ” in seconds, then “ϕ” is expressed in “electric” radians.

In other words, the digital module 19 located in the controller 8 receives, as an input signal, a digital signal "zc_E-I_phaseU" and measures, by means of a timer 26, the period "T _ϕ " of the time between the transition through zero of the phase voltage applied to phase U, and transition through zero current in phase U (transition type "high-low" signal "zc_E-I_phaseU"). In this case, the moment of transition through zero of the voltage applied to the phase U is obtained by traditional methods inside the modulator 5: the angle ϕ _act is derived from the measurement of the aforementioned time period based on the frequency of the electric current indicated by freq_act according to the above formula.

There are operating conditions under which the aforementioned fan 7 rotates in a direction corresponding to that during normal operation, even if the motor 2 driving it is not energized. Under these conditions, the so-called “catch rotor rotation” function may be requested.

To avoid the occurrence of extremely harmful transient extracurrents (for example, in the case of forced restart of the motor from zero speed), the drive module 1 has a system for reading CEMF.

To ensure this reading, the drive module 1 contains an additional resistor, known as "Rzc_fcem_run_on_fly", with a value equal to Rd, in the phase with a delay of 120 electrical degrees (phase W in the circuit shown in figure 1), relative to which and on which reading the current transition through zero (phase U in the case of this example).

If the "catch rotor rotation" function is requested, then the input signal "zce_run_on_fly" of the step "zc_E-I_detect" is controlled at the "low" logic level by the deviator software "sw1" controlled by the signal "bridge_enabled" inside the controller 8, forming the aforementioned second working configuration analog module 18.

It was shown, using the circuit analysis of a system with a three-phase bridge 3 having a high impedance (motor 2 is not energized), that the high-low transition of the output signal "zc_E-I_phaseU" of the step "zc_E-I_detect" coincides with the transition through zero (negative to positive) voltage between the two terminals U and W (positive when VU exceeds VW).

Between the aforementioned zero crossing and the zero crossing CEMF of phase U there are 30 electrical degrees used for the above calculations of the optimum voltage applied during operation when voltage is applied to the motor.

In other words, the CEMF reading system is based on the associated reading of CEMF between phases U and W.

Modeling of the circuit, confirming the above, is shown in Fig.7.

On the left side, regarding the situation in which the bridge 3 is turned on, you can see the curve "A" describing the zero-crossing curve, that is, the signal "zc_E-I_phaseU", in particular the zero-phase current crossing curve, and the "C" curve, describing the current passing through phase U, positive when exiting the motor.

On the right side, in which bridge 3 is turned off or no voltage is applied to it (curve "C" at zero), curve "A" represents the signal curve "zc_E-I_phaseU", representing CEMF transitions through zero.

The signal "zc_E-I_phaseU", which can be used to determine the transition of CEMF through zero for soft "catching the rotation of the rotor", is the same signal used to determine the transition through zero of the phase current when the "bridge_enabled" is at a "high" logic level.

The right side of the diagram shown in FIG. 7 also shows a “G” curve representing a CEMF curve connected between phases U and W, a “H” curve representing a drain-source voltage curve in a Q_low_U MOSFET with bridge 3 off and a curve "L" representing the CEMF curve Es of phase U.

In the case of a fan 7, rotating in the direction corresponding to the direction during normal operation, and supplying the motor 2 during its rotation, when the voltage is not applied, that is, when it performs the above function to “catch the rotation of the rotor” with obtaining smooth acceleration, it is necessary:

- attach the initial voltage Vs. equal to CEMF created by the motor, which is not energized;

- attach the initial frequency corresponding to the engine speed at the time of voltage supply.

The above calculation module 15 comprises a module 24, known as "freq_calc" and located in the controller 8, to determine the value of "freq_set" used in the case of using the "catch the rotation of the rotor" function.

Module 24 receives, as an input signal, the signal "zc_E-I_phaseU" from analog module 18.

The freq_calc module contains a timer 25 for measuring the Tfly time between two successive high-low transitions of the zc_E-I_phaseU signal.

The freq_calc module calculates the frequency F_fly = 1 / Tfly corresponding to the aforementioned time for applying the calculated value denoted by freq_fly as the initial value of freq_set for the case of using the catch rotor rotation function.

As shown, the drive module 1 comprises a second deviator software sw2 controlled by a bridge_enabled signal to set the freq_fly value to the freq_set frequency when the bridge 3 is turned off. When bridge 3 is turned on, "freq_set" corresponds to the value set by the aforementioned external interface - "freq_set_run".

In the presence of the aforementioned subtracting node 17, in order to avoid unwanted transitions to "freq_act", the drive module 1 contains the third deviator software "sw3" (Fig. 1), controlled by the signal "bridge_enabled" for contextually resetting the input of the second calculation module 16.

In practice, the zc_E-I_detect circuit measures the frequency of the electric current corresponding to the speed of the motor to which no voltage is applied, and then provides the freq_set synchronization to prevent the motor from restarting from zero speed.

Having the signal "freq_fly" and remembering that the memory of controller 8 stores the value of the constant K _E0 CEMF, the temperature coefficient α _mag and the reference temperature T _{0mag of} permanent magnets, and also that the controller 8 receives the current temperature T _{mag of} magnets from the sensor 10 , the controller 8 calculates the value "Vs_act" applied initially.

Having determined and "activated" the initial values "Vs_act" and "freq_act" applied to the engine, controller 8, resetting the bridge_enabled signal to high, is able to return the enable signal zce_run_on_fly to the logic level high and perform the above operations optimal and stable start-up (operation of integrating γ _act to achieve optimal voltage and restoring proportional action with γ _act to correct "freq_set" for stability purposes). The presence of the aforementioned resistor "Rzc_fcem_run_on_fly" does not affect the operation of the system in any way when "zce_run_on_fly" is at the logic level "high" and does not lead to any excessive additional current absorption of the drive device, when the drive device does not need to start the control action "freq_set_run "

Claims (7)

1. An electric drive module comprising:
- an electric motor (2) with permanent magnets,
- inverter (3) supplying an electric motor (2),
- DC stage (4) supplying the inverter,
- a modulator (5) driving the inverter,
- a controller (8) controlling the modulator (5) by means of a first digital signal (Vs_act) representing the amplitude of the phase voltages (Vs) applied to the electric motor (2), and by means of a second digital signal (freq_act) representing the frequency of the electric current in the phase voltages applied to the electric motor (2) based on a reference frequency (freq_set) installed outside or inside the controller (8), the drive module comprising:
- the first analog / digital stage (6) for calculating the optimum value of the angle (δ _opt ) of the lead in phase voltage (Vs) applied to the electric motor (2) relative to the anti-electromotive force (Es),
- a second analog / digital stage (12) for measuring the angle (ϕ _act ) between the voltage (Vs) applied to the electric motor and the phase current (Is),
characterized in that it comprises a temperature sensor (10) located near the permanent magnets and in communication with the first analog / digital stage (6) to provide the first analog / digital stage (6) with information regarding the temperature of the permanent magnets, the first analog / digital stage (6) is configured to calculate the optimum phase advance angle (δ _opt ) as a linear function of the peak value of the phase current (Is) according to the following equation:

where "Ls" is the synchronous inductance of the electric motor, expressed in Henry, "p" is the number of poles of the electric motor, "K _corr " is a correction factor with a value greater than 1 and less than 1.2, "K _E0 " is the counter electromotive force, which is a constant at a reference temperature “T _0mag ” of permanent magnets, “α _mag ” is the coefficient of variation of the temperature of the permanent magnets with residual induction, “T _mag ” is the temperature of the permanent magnets measured by the temperature sensor (10), and troller (8) is programmed for
- determining, with a sample at the frequency of the electric current, the angle γ _act between the phase current (Is) and the anti-electromotive force (Es) as the difference between the aforementioned optimal value of the phase advance angle (δopt) and the angle (ϕ _act ) measured between the voltage (Vs ) applied to the electric motor (2) and phase current (Is). 2. The electric drive module according to claim 1, wherein the controller (8) comprises an integrator (14) for
- integrating the angle (γ _act ) between the phase current (Is) and the anti-electromotive force (Es) to determine the first digital signal (Vs_act). 3. The electric drive module according to claim 1 or 2, in which the controller (8) is programmed to calculate the correction (Δfreq) of the frequency proportional to the angle (γ _act ) between the phase current (Is) and the anti-electromotive force (Es), and to calculate the second digital signal (freq_act) as the difference between the value of the reference frequency (freq_set) and the correction (Δfreq) of the frequency. 4. The drive module according to claim 1 or 2, in which the second analog / digital stage (12) comprises an analog module (18) generating, as an output, a third digital signal (zc_E-I_phaseU) and having a first input in communication with the electrode drain of the low-voltage MOSFET (Q_low_U) of the supply branch of the inverter (3) of the first phase (U) of the electric motor (2), the second input communicating with the source electrode of the low-voltage MOSFET (Q_low_U), the third input communicating with the gate electrode of the low-voltage field MOSFET (Q_low_U), fourth resolution yuschy input,
moreover, the drive module contains a resolution module (20) of the analog module (18), generating a high or low resolution signal (zce_run_on_fly) and communicating with the fourth input for transmitting the resolution signal (zce_run_on_fly) to the analog module (18), while the analog module (18) is driven by a third input and is configured to detect a transition through zero current (Is) in the first phase (U) of the motor (2) based on a voltage drop between the first and second inputs, the high-low transition of the third digitally th signal (zc_E-I_phaseU) determines the transition through zero, at a high resolution signal (zce_run_on_fly), of the current in the first phase (U) when the inverter (3) is turned on. 5. The drive module according to claim 1 or 2, in which the second analog / digital stage (12) contains an analog module (18), generating, as an output, a third digital signal (zc_E-I_phaseU) and having a first input in communication with the electrode drain of the low-voltage MOSFET (Q_low_U) of the supply branch of the inverter (3) of the first phase (U) of the electric motor (2), the second input communicating with the source electrode of the low-voltage MOSFET (Q_low_U), the third input communicating with the gate electrode of the low-voltage field MOSFET (Q_low_U), fourth resolution yuschy input,
wherein the drive module contains:
- a resistor (Rzc_fcem_run_on_fly) in the second phase (W) with a delay of 120 electrical degrees relative to the first phase (U),
- enable module (20) of the analog module (18) generating a high or low resolution signal (zce_run_on_fly) and communicating with the fourth input for transmitting the enable signal (zce_run_on_fly) to the analog module (18), while the analog module (18) contains:
- a resistor connected in series with the first input with a value equal to the value of the resistor (Rzc_fcem_run_on_fly) in the second phase (W), driven by the third input and configured to detect the transition through zero of the anti-electromotive force connected between the first phase (U) and the second phase (W), with a low resolution signal (zce_run_on_fly) and the inverter (3) turned off. 6. The electric drive module according to claim 5, in which the controller (8) contains:
- the first calculation module (15), communicating with the analog module (18) for receiving, as an input, the third digital signal (zc_E-I_phaseU) and calculating the frequency (freq_fly) of the electric current corresponding to the time between two successive high-low transitions "the third digital signal (zc_E-I_phaseU),
- controller (8), programmed to give the value of the reference frequency (freq_set) equal to the frequency (freq_fly) of the electric current corresponding to the time between two successive high-low transitions of the third digital signal (zc_E-I_phaseU), and when the inverter ( 3) off, setting the reference frequency (freq_set) occurs inside the controller (8). 7. The drive module according to claim 6, in which the controller (8) is configured to generate a fifth digital signal (bridge_enabled) that takes a high logic level, if the modulator (5) controls at least one branch of the inverter (3), and a low logic level if all branches of the inverter (3) are disconnected,
moreover, the resolution module (20) is controlled by the fifth digital signal (bridge_enabled) to give a high value to the resolution signal (zce_run_on_fly) when the fifth digital signal (bridge_enabled) has a high logic level, or to give a low value to the resolution signal (zce_run_on_fly) when the fifth digital signal (bridge_enabled) has a low logic level,
wherein the calculation module (15) is controlled by the fifth digital signal (bridge_enabled) to give the reference frequency (freq_set) a value equal to the frequency (freq_fly) of the electric current corresponding to the time between two successive high-low transitions of the third digital signal (zc_E- I_phaseU) when the fifth digital signal (bridge_enabled) has a low logic level, or giving the reference frequency (freq_set) a predetermined frequency value (freq_set_run), when the fifth digital signal (bridge_enabled) has a high logic level,
moreover, the setting of the reference frequency (freq_set) occurs inside the controller (8) when the fifth digital signal (bridge_enabled) has a low logic level, and outside the controller (8) when the fifth digital signal (bridge_enabled) has a high logical level.

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