RU2522907C2 - Pulse-doppler radio altimeter - Google Patents

Abstract

FIELD: instrumentation.

SUBSTANCE: invention can be used in pulse Doppler radio altimeters (RA). It consists in emitting probing pulse packet towards underlying terrain. Mote here that the number of radio pulses (RP) and pulse time are selected to ensure maximum number of RP for a priori delay (AD) set by exchange controller (EC). Besides, it serves to rule out ambiguity of altitude measurement and getting of emitted signal in uncertainty zone wherein reflected signal is sought. Packet of radio pulses reflected from underlying terrain is received to convert video pulses into the train of digital binary signals (DBS) at sampling rate. Theses are memorised in synchronism with RP and, at termination of radiation, the address of memory cell is defined that corresponds to AD of reflected signal relative to radiation packet start. Narrow-band Doppler filtration of DBS read from memory cells in the range of memory cell search is performed. Filtration total result is accumulated over all digital binary signals of received packet for every estimated delay. Decision is made on signal availability by excess over preset threshold of accumulation to define the reflected signal delay relative to RP packet start. Data on measured altitude is output to RA via EC.

EFFECT: enhanced performances, higher radiation hiding.

8 dwg

Description

The present invention relates to the field of radar, in particular on-board flight altitude meters, and can be used in pulse-Doppler radio altimeters for flight control systems of aircraft.

Known [1] is a pulse-Doppler radio altimeter comprising a transmitter with radiation means, a receiver with means of receiving reflected signals, a computational circuit that measures height by emitting pulse sounding signals, receiving, narrow-band processing of reflected signals by a Doppler filter, measuring the delay time of their propagation to the reflecting surface and back. In order to maintain operability in a wide range of aircraft altitudes and speeds, the pulse repetition period and, accordingly, the radiation session time of the radio altimeter have to be increased, and the passband of the Doppler filter must be expanded, which reduces the signal-to-noise ratio at the output of the Doppler filter, which can be compensated by either increasing the radiated power, or by increasing the time of the radiation session, as a result, the secrecy of the radiation worsens.

Known [2] is a pulsed radar in which the repetition rate of the probe pulses depends on the measured height, it has to be lowered with increasing height in order to maintain operability, which leads to an increase in the time of the radiation session, as a result, the secrecy of radiation is worsened.

Known [3, 4] are pulse-Doppler radar stations operating with a high repetition rate of probe pulses, the pulse emission period is less than the maximum possible delay of the reflected signals from the target, and the radiation session has a short time when measuring range. However, in these stations there are problems associated with the ambiguity of range reports to the target, which are eliminated, for example, by the two-frequency method of eliminating the ambiguity of range reports (for example, [3], p. 375, Fig. 13.20), which allows for unambiguous measurement of the range and stably works only with targets having a small effective scattering area (for example, when selecting moving targets, over a calm water surface). Pulse-Doppler radar stations, as a rule, work with a narrow radiation pattern of the antenna, which has a tracking system of angular scanning. To account for the influence of the evolution of the aircraft, radio altimeters in most cases work with antennas with a wide radiation pattern [5].

As shown in [5] on page 146, a rougher reflective surface, ceteris paribus, corresponds to a wider envelope of the echo pulse and a wider spectrum of envelope fluctuations.

Above an extended earth's surface (such as a forest, arable land, an agitated water surface), the two-frequency method of eliminating ambiguity does not take into account the effect of the expansion of the envelope of reflected pulses on the station's operation, which can lead to a complete loss of its operability due to the overlap in time of two adjacent range signals. To maintain the operability of the station, the pulse repetition rate must be reduced, which increases the time of the radiation session when measuring the height and worsens the secrecy of the radiation.

The closest in technical essence is a pulse-phase meter of the thickness of the layers of dissimilar liquids, as well as their relative changes with increased accuracy [6], containing a synchronizer, a computing device, an analog-to-digital converter, a phase shifter, a pulse modulator, a controlled attenuator, a video amplifier, BOSU, gain control unit, attenuation control unit, voltage controlled current source, exchange controller, antenna system connected in series, circulator, low-noise UV H, phase detector, directional coupler, discretely controlled microwave generator, the output of which is connected to the input of the directional coupler, the second output of which is connected to the second input of the phase detector, the first input of which is connected to the output of the low-noise amplifier, the input of which is connected to the output of the circulator, input / output which is connected to the antenna system, and the first output of the directional coupler is connected to the first input of the pulse modulator, the second input of which is connected to the second output of the synchronizer, the output of the pulses the other modulator is connected to the second input of the phase shifter, the first input of which is connected to the first output of the synchronizer, and the output to the first input of the controlled attenuator, the output of which is connected to the input of the circulator, and the second input to the output of the current source, voltage controlled, the input of which is connected to the output attenuation control unit, all the first inputs of which are connected via data bus to all first inputs of the gain control unit, all sixth inputs of the BOZU, all first inputs / outputs of the exchange controller, all third whose inputs / outputs are the inputs / outputs of the meter, as well as all twelve inputs of the computing device, the second, third, fourth, fifth outputs of which are connected respectively to the second inputs of the attenuation control unit, gain control unit, exchange controller, BOSE, and the sixth, seventh , thirteenth outputs, respectively, with the third, fourth, seventh inputs of the BOZU, eighth, ninth outputs, respectively, with the second and third inputs of the synchronizer, tenth, eleventh outputs, respectively, with the second and the first inputs of a discretely controlled microwave generator, the first input is with the fourth output of the synchronizer, the first input of which is connected to the first output of the BOZU, the first input of which is connected to the third output of the synchronizer and the second input of the ADC, all outputs of which are connected to all fifth inputs of the BOZU, and the first ADC input - with the output of the video amplifier, the first input of which is connected to the output of the phase detector, the second input - with the output of the gain control unit.

A pulse-phase meter emits in the direction of the underlying surface and receives short packets of radio pulses. In this case, the radio pulses in the receiver are converted into bipolar video pulses, the envelope of which fluctuates in amplitude with a frequency determined by the Doppler frequency shift of the signals reflected from the underlying surface (Fig. 1).

The disadvantage of a pulse-phase meter is that with an increase in the range of measured heights, it is necessary to increase the repetition period, respectively, the duration of the packets, or the power of the probe radio pulses, which impairs the secrecy of radiation.

The aim of the present invention is to expand the functionality of the meter, increasing the secrecy of radiation and the maximum measured height without increasing the radiated power.

This goal is achieved by the fact that in a pulse-phase meter containing a synchronizer, a computing device, an analog-to-digital converter, a phase shifter, a pulse modulator, a controlled attenuator, a video amplifier, a BOZU, a gain control unit, an attenuation control unit, a voltage-controlled current source, a controller exchange, transceiver antenna, circulator, low-noise UHF, phase detector, directional coupler, discretely controlled microwave generator, the output of which is connected to the input of the directional coupler, the second output of which is connected to the second input of the phase detector, the first input of which is connected to the output of a low-noise amplifier, the input of which is connected to the output of the circulator, and the first output of the directional coupler is connected to the first input of the pulse modulator, the second input of which is connected to the second output of the synchronizer, the output the pulse modulator is connected to the second input of the phase shifter, the first input of which is connected to the first output of the synchronizer, and the output to the first input of the controlled attenuator, the output is It is connected to the input of the circulator, and the second input to the output of a voltage-controlled current source, the input of which is connected to the output of the attenuation control unit, all the first inputs of which are connected via the data bus to all the first inputs of the gain control unit, all the sixth inputs / outputs of the BOZU, all the first inputs / outputs of the exchange controller, all the third inputs / outputs of which are the inputs / outputs of the meter, as well as all twelve inputs / outputs of the computing device, the second, third, fourth, fifth output which are connected respectively to the second inputs of the attenuation control unit, gain control unit, exchange controller, BOSU, and the sixth, seventh, thirteenth outputs, respectively, with the third, fourth, seventh inputs of the BOSU, eighth, ninth outputs, respectively, with the second and third inputs of the synchronizer , tenth, eleventh outputs - respectively, with the second and first inputs of a discretely controlled microwave generator, the first input - with the fourth output of the synchronizer, the first input of which is connected to the output of the BOZU, the first input to which is connected to the third output of the synchronizer and the second input of the ADC, all the outputs of which are connected to all fifth inputs of the BOZU, and the first input of the ADC to the output of the video amplifier, the first input of which is connected to the output of the phase detector, the second input to the output of the gain control unit, input / the output of the transceiver antenna is connected to the input / output of the circulator and the divider with a variable division ratio (DPD) of the synchronizer is controlled by a computer, all twelve inputs / outputs of the computing device are connected inens with the fourth inputs of the synchronizer in such a way that a packet of probing radio pulses is emitted in the direction of the underlying surface (Fig. 2), and, during the a priori delay Ta set by the exchange controller, N pulses are emitted, the repetition period Tni of which is set equal to the sum of some constant for the data N and Ta, the values of Tn and the variable Tm, and the value of Tn is determined depending on N, Ta, the value Th of the uncertainty zone of the delay of the reflected signal, the duration of the generated direct (induced) signal and the range zone Tm of period change (wobble), and the values of N and Tn are determined from the solution of the system of equations:

$$\{\begin{array}{l}N*Tn=Ta+Th\\ (N-\mathrm{one})*(Tn+Tm)=Ta-To,\end{array}$$

receive a packet of radio pulses reflected from the underlying surface, convert the video pulses into a sequence of digital binary signals with a sampling period Tk, remember this sequence synchronously with the beginning of the packet of emitted radio pulses, and at the end of the radiation, determine the memory cell address Adr (k, Te) corresponding to the delay of the k-th burst video pulse - Those whose value is in the range from the a priori delay value Ta to the value Ta + Th and is determined from the equation:

$$AdR(k,Tc)=({\displaystyle \sum _{i=0}^{k}Tui+Tc})/Tk,$$

narrow-band Doppler filtering of digital binary signals is performed, read sequentially from memory cells with the address Adr (k, Tc), the accumulation of the total filtering result for all video pulses of the packet at each value of the estimated delay, the decision on the presence of a signal when the predetermined accumulation threshold is exceeded in advance and the delay is determined .

At the same time, the radiation session time is reduced due to the fact that the accumulation of reflected signals, which is already consistent with the uncertainty zone Th, is the same as in the prototype, which is achieved much faster, since during the a priori delay (corresponding to the flight altitude at which the measurement ) several pulses are emitted with a period defined as described above, which can be significantly less than the delay of the reflected signal, and since the signal is processed on an unrealistic time scale without radiation, then itelno increased stealth emitted radio pulses. In addition, the use of a digital filter tuning (central frequency and bandwidth) when processing a sequence of digital binary memory signals (depending on the flight speed) allows not to expand its bandwidth at high flight speeds, rebuilding the central frequency, and significantly narrow it at low aircraft flight speeds. This adaptation of the filter bandwidth in speed allows at high flight speeds to keep the upper limit of the measured altitude range unchanged.

Comparative analysis with the prototype shows that the inventive pulse-Doppler radio altimeter is distinguished by its connections with other units. Thus, the claimed device meets the criteria of the invention - "novelty." The proposed implementation of the pulse-Doppler radio altimeter is unknown, leading to an increase in its useful properties - expanding functionality, increasing the secrecy of radiation and the maximum measured height without increasing the radiated power.

This allows us to conclude that the technical solution meets the criterion of "significant differences".

Figure 1 shows the timing diagram of the packs of emitted radio pulses and received video signals, figure 2 - the formation of the start pulses of the pulse modulator depending on the a priori delay Ta (for example), figure 3 is a functional diagram of a pulse-Doppler radio altimeter, Fig. 4 is a functional diagram of a synchronizer; FIG. 5 shows a block diagram of an algorithm for operating a digital filter.

The proposed pulse-Doppler radio altimeter (Fig. 3) contains a discretely controlled microwave generator 1, a directional coupler 2, a pulse modulator 3, a phase shifter 4, a controlled attenuator 5, a circulator 6, a transceiver antenna 7, a low-noise UHF 9, a phase detector 10, a video amplifier 11, ADC 12, BOSU 13, synchronizer 14, computing device 15, exchange controller 16, gain control unit 17, attenuation adjustment unit 18, voltage controlled current source 19.

Moreover, the output of the discretely controlled microwave generator 1 is connected to the input of the directional coupler 2, the first output of which is connected to the first input of the pulse modulator 3, the second input of which is connected to the second output of the synchronizer 14, the first output of which is connected to the first input of the phase shifter 4, the second input of which connected to the output of the pulse modulator 3, and the output to the first input of the controlled attenuator 5, the output of which is connected to the first input of the circulator 6, the second input / output to the input / output of the transceiver antenna 7, and the output of the circulator 6 is connected to the input of the low-noise UHF 9, the output of which is connected to the first input of the phase detector 10, the second input of which is connected to the second output of the directional coupler 2, and the output to the first input of the video amplifier 11, the second input of which is connected to the output of the unit 17 gain control, and the output with the first input of the ADC 12, all outputs of which are connected to all fifth inputs of the BOZU 13, the first input of which is connected to the second input of the ADC 12 and the third output of the synchronizer 14, the fourth output of which is connected to the first input computing device 15, the second, third, fourth, fifth outputs of which are connected respectively to the second inputs of the attenuation adjustment block 18, the gain control block 17, the exchange controller 16, the BOS 13, the sixth, seventh, thirteenth outputs, respectively, with the third, fourth, seventh inputs BOSU 13, the tenth, eleventh outputs, respectively, with the second and first inputs of a discretely controlled microwave generator 1, the eighth, ninth outputs, respectively, with the second and third inputs of the synchronizer 14, the first input of which is connected to output BOSU 13, all sixth inputs / outputs of which are connected via data bus to all twelve inputs / outputs of computing device 12, all first inputs / outputs of exchange controller 16, all third inputs / outputs of which are inputs / outputs of a pulse-Doppler radio altimeter, and - with all the first inputs of the gain control unit 17, all fourth inputs of the synchronizer 14, all the first inputs of the attenuation adjustment unit 18, the output of which is connected to the input of the voltage controlled current source 19, the output for which it is connected to the second input of the controlled attenuator 5, the input / output of the transceiver antenna 7 is connected to the input / output of the circulator 6.

Pulse-Doppler radio altimeter (PB) works as follows.

After supplying power to the RV, the computing device 15 carries out the signal 40 the initial setting of the trigger 28 of the radiation flag of the synchronizer 14 (figure 4), the signals 65 and 64 records the zero value of the gain and attenuation in blocks 17 and 18 of the regulation of gain and attenuation (Nus = 0, Nrel = 0), writes the signals zero through code 55 via the data bus 55 to the counter 44 of the RAM address BOSU 13 (this sets the logical signal level 39 — the end of the radiation and accumulation modes), polls the exchange controller 16 with external systems, which P revodit caliper delay measurement signal reflected by the underlying surface mode, sets the signals 63 and 67 of the microwave carrier frequency - the generator 1 to the middle working range.

After that, the computing device 15 starts the subroutine setting the parameters of the transceiver module (MRP) and the start of radiation and accumulation. The algorithm of the subroutine is shown in figure 5. The subroutine sets the carrier frequency Fnes on a discretely controlled microwave generator 1, writes the gain and attenuation values to the gain and attenuation adjustment blocks 17 and 18, writes the code zero to counter 44 of the RAM address of the RAM, starts the timer for the time tpp.ppm - parameter setting time in PPM (discretely controlled microwave generator 1, directional coupler 2, pulse modulator 3, phase shifter 4, controlled attenuator 5, low-noise UHF 9, phase detector 10, video amplifier 11), after which the radiation mode is started Nia and storage, emission analysis flag 42.

Through the circulator 6, the transceiver antenna 7 (input / output 2) provides radiation of radio pulses in the direction of the underlying surface.

The radio pulses received from the underlying surface of the transceiver antenna 7 are fed through a circulator 6 to a low-noise UHF 9, a phase detector 10, are detected, the video signals are amplified in a video amplifier 11, converted into a digital form in ADC12 and recorded in BOSU 13.

At the end of the subroutine, the computing device 15 reads the data BOSU 13, and then carries out data processing, scanning the range of delays, determining the time delay of the digital signals from the underlying surface.

The radiation and timing of the ADC 12 and recording in the BOSE 13 begin at time t0 (Fig.6) [3]. Radio pulses are emitted at ticks with numbers 0, km + 1, 2km + 3, ..., nk (m + 1), ... (figa, 6b), where m is the vernier parameter, k is the time range extender for the signal, and n is radiation number, Tizl - period of clock pulses.

The conversion of the received signal (pigv) in the ADC 12 and recording in the BOSU 13 is carried out for each clock cycle with a period of Tozu (Fig.6g, 6d). If the condition for the coincidence of the fronts of the clock pulses of radiation and the timing of the ADC 12 and BOSU 13:

$$km*T\mathrm{and}sl=k(m-\mathrm{one})*T\mathrm{about}s\mathrm{at}(\mathrm{one})$$

Condition (1) can be written as:

$$F\mathrm{and}sl/m=F\mathrm{about}s\mathrm{at}/(m-\mathrm{one})(2)$$

where Fsl = 1 / Tsl is the frequency of radiation clock pulses; Fozu = 1. / Tozu - the frequency of the clock pulses of the ADC 12 and BOSU 13.

If the second and subsequent radiations are emitted at moments n * (km + 1) * Tizl, then the next clock cycle of the ADC 12 and BOSU 13 will come at the moment n * (k (m-1) +1) * Toze with a delay dtn, then from the equation :

n * (km + 1) * Tizl + dtn = n * (k (m-1) +1) * Tozu

it can be shown that the fifth radiation starts earlier than n * (k (m-1) +1) of the ADC 12 and BOSU 13 clock pulses by:

dtn = n * Tozu / m.

From this we get that in order to restore the received signal with a step of Toz / m, Nisl = m is necessary, and the volume of the BOSE 13 Loz is determined from the equation:

m * (km + 1) * Tizl = Lozu * Tozu,

then the volume of BOSE 13 necessary to restore the received signal is equal to Lose = (km + 1) (m-1).

The above reasoning allows us to obtain that to restore the value of the received signal at the delay i * dt in digital form, we can derive the expression:

Ui = RAM {(i mod M) * m + [i / M]},

where: brackets {...} mean the contents of the RAM cell with the given number, the expression (i mod M) is the remainder of dividing i by M, and the bracket [...] is the integer part of the number, M = k (m-1) +1. 6e shows an example of a reconstructed signal for k = 1, m = 8.

When reflected from the underlying surface, the change in the attenuation of the reflected signals at the input of the low-noise UHF 9 can be of the order of 30 dB, which leads to amplitude fluctuations and a change in the steepness of the reflected signal front and, as a result, to an additional error in measuring the delay of the reflected signal and the flight altitude of the aircraft.

To maintain a stable steepness of the front of the reconstructed signal at the delay i * dt, the computing device 15 starts, after each radiation cycle and accumulation, the automatic gain control routine of the video amplifier 11 and the emitted radio pulse power through the gain control unit 17, the gain control unit 18, and the current source 19 of the UN, respectively ( Fig.7). The level of the reconstructed reflected signal is estimated. If the signal level exceeds the threshold, then the radiated power and the gain of the video amplifier 11 are reduced, if it does not exceed the threshold, then an increase.

After stabilization of the steepness of the front of the reflected signal (approximately 5-6 radiated packets of radio pulses), the computing device 15 provides the result of the measured flight altitude to the exchange controller 16.

If the measured height becomes higher than a certain height Htek (current height), when the energy potential of the RS decreases to a critical value at which the height measurement mode can give large errors, the RS programmatically using the computing device 15 turns on the Doppler filtering mode for measuring the height.

After turning on the RV at a height higher than Htec, the computing device 15 carries out with the signal 40 (Fig. 4) the initial setting of the trigger 28 of the radiation flag of the synchronizer 14, with the signal 83 the initial setting of the shift register 82 of the AP control unit 7, the signals 65 and 64 record the zero gain and attenuation value in blocks 17 and 18 control gain and attenuation (Nus = 0, Nrel = 0), writes signals 52 and 53 on the data bus 55 to the counter 44 of the RAM address BOSU 13 code zero value (thereby setting a low logic signal level 39-end of the radiation modeaccumulation), polls the controller 16 exchange with external systems (which sets the a priori value of the delay Ta), on the bus 55 sets the division coefficient DPKD 32 at the inputs dN0, dN1, dA [7] (respectively, the pulse repetition period of the start pulses of the pulse modulator 3) depending from the a priori delay Ta and puts the PB in the measurement mode of the delay of the signal reflected from the underlying surface, sets the carrier frequency of the microwave generator 1 to the middle of the operating range with signals 63 and 67.

After that, the computing device 15 starts the subroutine for setting the parameters of the transceiver module (MRP) and the start of radiation and accumulation, after which the radiation and accumulation modes are started, and the radiation flag is analyzed 42.

Packs of sounding radio pulses are emitted in the direction of the underlying surface, and, depending on the value of the a priori height recorded from the exchange controller 16 to the computing device 15, which determines the operating mode of the synchronizer 14, which controls the pulse modulator 3, which opens (during the delay of the reflected signal) to radiation N pulses with a period Tui determined by the sum of a certain constant for a given N, Ta value of Tn and a variable Tm, and the value of Tn is determined depending on N, Ta, the value of Th of the zone is undefined of the delay of the reflected signal (signal search range), the range of Tm of the period (wobble) change, the duration of the generated direct (induced) signal at the input of the analog-to-digital converter 12, a packet of radio pulses reflected from the underlying surface are received, video pulses from the output of the video amplifier 11 are converted to analog -digital converter 12 to a sequence of digital binary signals with a sampling period Tk, stored in the BOSU 13 synchronously with the beginning of the pack of emitted radio pulses, and at the end radiation, the computing device 15 determines the address of the memory cell Adr (k, Tc), performs narrow-band Doppler filtering of digital binary signals read sequentially from the memory cells of the BOZU 13, accumulates the total filtering result for all video pulses of the packet for each value of the estimated delay, in accordance with a given program makes a decision about the presence of a signal when the predetermined accumulation threshold is exceeded and determines the delay of the reflected signal.

To implement Doppler filtering (Fig. 8), a digital low-pass filter is used, the prototype of which is the well-known normalized Butterworth 3rd-order analog filter with a transfer function of the form [8]:

. $$H(s)=\frac{\mathrm{one}}{(s+\mathrm{one})({\mathrm{<figure-callout\; id="2"\; label="s"\; filenames="00000017.png,00000018.png"\; state="\{\{state\}\}">s</figure-callout>}}^2+s+\mathrm{one})}$$

.

To go to the digital filter, a bilinear transformation with the substitution of the form is used

. $$s=\frac{\mathrm{one}-z^{-\mathrm{one}}}{\mathrm{one}+z^{-\mathrm{one}}}$$

.

As a result, the expression for the transfer function takes the form

. $$H(z^{-\mathrm{one}})=\frac{\frac{\mathrm{one}}{6}{(\mathrm{one}+z^{-\mathrm{one}})}^3}{\mathrm{one}+\frac{\mathrm{one}}{3}{z}^{-2}}$$

.

If the signal is digitized with a period Δt, then the cutoff frequency of this filter is determined by the expression

. $${\mathrm{<figure-callout\; id="0"\; label="\omega "\; filenames="00000021.png"\; state="\{\{state\}\}">\omega </figure-callout>}}_0=\frac{\mathrm{one}}{\mathrm{four}}\cdot \frac{2\pi }{\Delta t}$$

.

To obtain a digital low-pass filter with a given cutoff frequency ω _s , the bandwidth is converted using substitution

, где $$\alpha =\frac{\sin \left(\frac{{\omega }_0-{\omega }_{с}}{2}\cdot \Delta t\right)}{\cos \left(\frac{{\omega }_0+{\omega }_{с}}{2}\cdot \Delta t\right)}$$

. $$z^{-\mathrm{one}}=\frac{z^{-\mathrm{one}}-\alpha }{\mathrm{one}-\alpha \cdot z^{-\mathrm{one}}}$$

where $$\alpha =\frac{\sin \left(\frac{{\omega }_0-{\omega }_{\mathrm{from}}}{2}\cdot \Delta t\right)}{\cos \left(\frac{{\mathrm{<figure-callout\; id="0"\; label="\omega "\; filenames="00000021.png"\; state="\{\{state\}\}">\omega </figure-callout>}}_0+{\omega }_{\mathrm{from}}}{2}\cdot \Delta t\right)}$$

.

After performing all the transformations, the transfer function of the Doppler filter is written as

$$H(z^{-\mathrm{one}})=\frac{{\mathrm{<figure-callout\; id="0"\; label="b"\; filenames="00000021.png"\; state="\{\{state\}\}">b</figure-callout>}}_0\cdot {(\mathrm{one}+z^{-\mathrm{one}})}^3}{\mathrm{one}+a_{\mathrm{one}}{z}^{-\mathrm{one}}+a_2{z}^{-2}+a_3{z}^{-3}},$$

where the filter coefficients are determined by the expressions

, $$a_1=-\frac{\alpha {({\alpha }^2+11)}^3}{{\alpha }^2+3}$$

, $$a_2=-\frac{11{\alpha }^2+1}{{\alpha }^2+3}$$

, $$a_3=-\frac{\alpha \cdot (3{\alpha }^2+1)}{{\alpha }^2+3}$$

. $${\mathrm{<figure-callout\; id="0"\; label="b"\; filenames="00000021.png"\; state="\{\{state\}\}">b</figure-callout>}}_0=\frac{{(\mathrm{one}-\alpha )}^3}{2({\mathrm{<figure-callout\; id="2"\; label="\alpha "\; filenames="00000017.png,00000018.png"\; state="\{\{state\}\}">\alpha </figure-callout>}}^2+3)}$$

, $$a_{\mathrm{one}}=-\frac{\alpha {({\mathrm{<figure-callout\; id="2"\; label="\alpha "\; filenames="00000017.png,00000018.png"\; state="\{\{state\}\}">\alpha </figure-callout>}}^2+\mathrm{eleven})}^3}{{\mathrm{<figure-callout\; id="2"\; label="\alpha "\; filenames="00000017.png,00000018.png"\; state="\{\{state\}\}">\alpha </figure-callout>}}^2+3}$$

, $$a_2=-\frac{\mathrm{eleven}{\mathrm{<figure-callout\; id="2"\; label="\alpha "\; filenames="00000017.png,00000018.png"\; state="\{\{state\}\}">\alpha </figure-callout>}}^2+\mathrm{one}}{{\mathrm{<figure-callout\; id="2"\; label="\alpha "\; filenames="00000017.png,00000018.png"\; state="\{\{state\}\}">\alpha </figure-callout>}}^2+3}$$

, $$a_3=-\frac{\alpha \cdot (3{\mathrm{<figure-callout\; id="2"\; label="\alpha "\; filenames="00000017.png,00000018.png"\; state="\{\{state\}\}">\alpha </figure-callout>}}^2+\mathrm{one})}{{\mathrm{<figure-callout\; id="2"\; label="\alpha "\; filenames="00000017.png,00000018.png"\; state="\{\{state\}\}">\alpha </figure-callout>}}^2+3}$$

.

The filtering algorithm is described by a difference equation

y _n = b ₀ x _n + 3b ₀ x _n-1 + 3b ₀ x _n-2 + b ₀ x _{n-3-a 1} y _n-1 -a ₂ y _{n-2-a 3} y _n-3 ,

where x _n are the signal values at the filter input,

y _n are the output values of the signal.

The computing device 15 compares the output values of the signals with a threshold, and if the accumulated amount is less than the threshold, it increases (without radiation) the search discret Ncm by one step and repeats in full the digital binary signal processing mode, thereby providing a signal search in the Th range if the accumulated the amount exceeds the threshold, it stops processing and gives the value of the found height Hp to the output of the exchange controller 16:

Hrv = Ncm * 2 * C * Tk-ΔNcm,

where: C is the speed of light,

ΔNcm - the absolute value of the bias estimates of the measured height [5].

If the signal reflected from the underlying surface is not detected, then the subroutine for setting the parameters of the transceiver module (MRP) and the start of radiation and accumulation are started, after which the radiation and accumulation modes are started, the reflected signals are stored in BOSU 13 simultaneously with the beginning of the packet of emitted radio pulses, and the end of the radiation, the computing device 15 determines the address of the memory cell Adr (k, Tc) by narrow-band Doppler filtering of digital binary signals read sequentially from the cell Bozu memory 13 stores the total result of filtering for all video pulse burst whenever the estimated value of delay in accordance with a predetermined program decides the presence of excess accumulation signal preassigned threshold and determining the delay of the reflected signal.

According to the proposed technical solution, a circuit diagram has been developed and an RW with a short radiation session time has been made, in which, when processing digital memory signals, depending on the a priori flight speed Va, the parameters of the digital filter are tuned (central frequency and bandwidth), which makes the processing more optimal and, as a result, increase the maximum measured height without increasing the radiated power and reduce the radiation session time in a wide range of aircraft speeds [9].

Literature

1. French patent No. 2.120.208.1972

2. French Patent No. 2,596.873.1988

3. Grigorina-Ryabova VV Radar devices. M., Soviet Radio, 1970

4. Skolnik M., Trofimova K.N. Radar devices and systems. M., Soviet Radio - 1979

5. Zhukovsky A.P., Onoprienko E.I., Chizhov V.I. Theoretical Foundations of Radio Altimetry. M., Soviet Radio - 1979

6. RF patent No. 2188399 dated 06/21/1999, class. ⁷ G01F 23/284. Publ. 06/21/1999

7. Pukhalsky GI, Novoseltseva T.Ya., Design of discrete devices on integrated circuits. M., Radio and Communications - 1990, 274 p.

8. Goldenberg L.M., Matyushkin B.D., Polek M.N. Digital signal processing. Textbook for higher education, M., Radio and communications - 1990, 2560.

9. Radio altimetry - 2010. Proceedings of the Third All-Russian Scientific and Technical Conference, Kamensk-Uralsky, October 19-21, 2010, 71 pp.

Claims (1)

 A pulse-Doppler radio altimeter containing a synchronizer, a computing device, an analog-to-digital converter (ADC), a phase shifter, a pulse modulator, a controlled attenuator, a video amplifier, a buffer random access memory (BOS), a gain control unit, an attenuation control unit, a voltage-controlled current source , exchange controller, transceiver antenna, circulator, low-noise high-frequency amplifier (UHF), phase detector, directional coupler, discretely controlled ultra-high frequency (microwave) generator, the output of which is connected to the input of the directional coupler, the second output of which is connected to the second input of the phase detector, the first input of which is connected to the output of the low-noise amplifier, the input of which is connected to the output of the circulator, and the first output of the directional coupler is connected to the first input of the pulse a modulator, the second input of which is connected to the second output of the synchronizer, the output of the pulse modulator is connected to the second input of the phase shifter, the first input of which is connected to the first output of synchronizer, and the output is connected to the first input of the controlled attenuator, the output of which is connected to the first input of the circulator, and the second input of the controlled attenuator is connected to the output of the voltage-controlled current source, the input of which is connected to the output of the attenuation control unit, all first inputs of which are connected via the data bus with all the first inputs of the gain control unit, all six inputs / outputs of the BOZU, all the first inputs / outputs of the exchange controller, all of the third inputs / outputs of which are inputs / outputs of a radio altimeter as well as all twelve inputs / outputs of the computing device, the second, third, fourth, fifth outputs of which are connected respectively to the second inputs of the attenuation control unit, gain control unit, exchange controller, BOSE, and the sixth, seventh, thirteenth outputs, respectively, with the third, the fourth, seventh inputs of the BOZU, the eighth, ninth outputs, respectively, with the second and third inputs of the synchronizer, the tenth, eleventh outputs, respectively, with the second and first inputs of a discretely controlled microwave generator RA, the first input - with the fourth output of the synchronizer, the first input of which is connected to the output of the BOZU, the first input of which is connected to the third output of the synchronizer and the second input of the ADC, all outputs of which are connected to all fifth inputs of the BOZU, and the first input of the ADC - with the output of the video amplifier, the first input of which is connected with the output of the phase detector, the second input is with the output of the gain control unit, the input / output of the transceiver antenna is connected to the input / output of the circulator, characterized in that the divider is controlled from a variable m division coefficient (DKPD) of the synchronizer using a computing device, while all twelve inputs / outputs of the computing device are connected to the fourth inputs of the synchronizer.

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