RU2490783C1 - Selective amplifier for precision analogue-to-digital interface - Google Patents

Abstract

FIELD: radio engineering, communication.

SUBSTANCE: instrumentation amplifier for a precision analogue-to-digital interface has an input precision converter of first and second sources of input voltage with first and second anti-phase outputs, an active adder, the inverting input of which is connected to the first output of the input precision converter, and the non-inverting input is connected to the second output of the input precision converter, the output of the device is connected to the output of the active adder, wherein the first and second sources of input voltage are connected to inputs of the input precision converter relative the common bus of the power supply, the active adder has first and second voltage-to-current converters and a first output current-to-voltage converter.

EFFECT: improved amplitude characteristics of output signals and obtaining maximum coefficient of attenuation of the input in-phase voltage.

3 cl, 13 dwg

Description

The present invention relates to the field of radio engineering and communication and can be used as a device for amplifying analog signals of sensors of various functional purposes (for example, in telemetry devices, measuring systems of medical equipment, automatic control and special measurement techniques).

The creation of analog and analog-to-digital interfaces of mixed systems on a chip (SoC), focused on interaction with sensitive elements (sensors) of the bridge type, always involves the use of instrumental amplifiers (DUTs) with both fixed and controlled parameters that perform the functions of suppressing common-mode voltage and gain differential voltage. These devices are the basis for both analog ports and a whole class of complex functional blocks (SF blocks) of SoC. A sufficiently large dynamic range of the measured values and relatively high conversion accuracy predetermined the use of precision operational amplifiers (op amps) in such interfaces. Most of the currently known solutions are associated with the use of the classical structure of constructing a DUT consisting of three op amps and a set of precision resistors [1-16].

The closest prototype of the claimed device is a tool amplifier, presented in patent US 2010/0259323 A1 of figure 1 by Paul L. Bugyik. It contains an input precision converter 1 of the first 2 and second 3 sources of input voltage with the first 4 and second 5 antiphase outputs, an active adder 6, the inverting input of which 7 is connected to the first 4 output of the input precision converter 1, and the non-inverting input 8 is connected to the second 5 output input precision Converter 1, the output of the device 9, associated with the output of the active adder 6, and the first 2 and second 3 sources of input voltages are fed to the inputs of the input precision Converter 1 rel respect to the total power supply bus 10.

Significant disadvantages of the known device, the architecture of which is also present in many other instrumentation amplifiers [1-16], are as follows:

1. The first flaw. Even when using strictly identical operational amplifiers in the structure of the input precision converter 1, the limiting value of the attenuation coefficient of the input common-mode signal (K _sn ) will be determined by the following relation:

$$K_{cn}=\frac{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{26}}{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{25}+{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{26}}\left(\mathrm{one}+\frac{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{2\mathrm{four}}}{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{23}}\right)-\frac{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{2\mathrm{four}}}{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{23}}=\frac{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{26}-{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{25}\frac{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{2\mathrm{four}}}{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{23}}}{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{25}-{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{26}}=\frac{u_{\mathrm{at}sx.\mathrm{from}}}{u_{\mathrm{at}x.\mathrm{from}}},(\mathrm{one})$$

where R ₂₃ ÷ R ₂₆ - resistors included in the structure of the active adder (6) of figure 1.

Therefore, a deep attenuation of the input common-mode signal in the DUT of Fig. 1 is possible only with strictly agreed resistors R ₂₃ ÷ R ₂₆ .

It can be shown that even when the conditions of strict identity of the resistances of the resistors (R ₂₃ = R ₂₄ = R ₂₅ = R ₂₆ = R) are met, the coefficient K _{sn is} not better than

where Θ _R is the error of the resistances of the resistors of the active adder (6) (Fig. 1).

Thus, from the above relations (1) and (2), it can be seen that the maximum realized attenuation coefficient of the input common-mode signal of the DUT of Fig. 1 is limited by the error of the resistors of the active adder Θ _R. As a result, even for precision technologies for which Θ _R = 0.1%, the attenuation coefficient of the input common-mode voltage of the DUT of Fig. 1 does not exceed 60 dB, which is clearly not enough to build even non-precision measuring and sensor systems. Therefore, in the production of such DUT circuits, an expensive, precision laser tuning of resistors R ₂₃ ÷ R _{26 is used} , aimed at achieving the required scatter of qualitative indicators of resistors (for example, Θ _R = 0.01%), at which the attenuation coefficient of the input common-mode signal does not exceed 75 dB .

2. The second drawback. The inefficiency of using the pass-through characteristics of the first (18) and second (19) operational amplifiers in the structure of the DUT of FIG. 1. This is because the main part of the amplitude characteristic of the first (18) and second (19) operational amplifiers contains a component of the input common-mode voltage:

, $$U_{19}=U_{сн}+\frac{R_{21}}{R_{22}}{U}_{д},(3\text{)}$$

$$U_{\mathrm{one}8}=U_{\mathrm{from}n}-\frac{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{20}}{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{22}}{U}_d$$

, $$U_{\mathrm{one}9}=U_{\mathrm{from}n}+\frac{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{2\mathrm{one}}}{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{22}}{U}_d,(3\text{)}$$

where U ₁₈ and U ₁₉ are the voltage at the output of the first (18) and second (19) operational amplifiers, respectively, U _sn are the common-mode voltages at the inputs of the instrument amplifier, U _d are the differential voltage at the inputs of the instrument amplifier, R ₂₀ ÷ R ₂₂ are the resistors feedback circuits included in the structure of the input precision Converter 1 of figure 1.

The main objectives of the invention are as follows.

1. To exclude the precision resistors R ₂₃ ÷ R ₂₆ from the structure of the active adder 6 and, therefore, the need for expensive precision laser tuning of these resistors. This will allow not only to increase the yield of suitable products during production, but also to obtain the highest possible attenuation coefficient of the input common-mode voltage in the structure of the proposed DUT.

2. To exclude the in-phase component in the output signals of the input precision Converter 1, which will further improve the efficiency of using their amplitude characteristics.

The problem is solved in that in the instrumental amplifier of figure 1, containing an input precision converter 1 of the first 2 and second 3 input voltage sources with the first 4 and second 5 antiphase outputs, an active adder 6, the inverting input of which 7 is connected to the first 4 output of the input precision the converter 1, and the non-inverting input 8 is connected to the second 5 output of the input precision converter 1, the output of the device 9, connected to the output of the active adder 6, the first 2 and second 3 input sources yarn is fed to the inputs of the input precision converter 1 relative to the common bus of power supplies 10, new elements and connections are provided - the active adder 6 contains the first 11 voltage-current converter, the inverting input of which is connected to the output of the device 9, and the non-inverting input is connected to the common bus power supplies 10, the second 12 voltage-current converter, the inverting input of which is connected to the inverting input 7 of the active adder 6, and the non-inverting input is connected to the non-inverting input 8 ac active adder 6, the first 13 current output of the first 11 voltage-current converter is connected to the first 14 current output of the second 12 voltage-current converter and is connected to the non-inverting current input of the first 15 current-voltage converter, the second 16 current output the first 11 voltage-current converters are connected to the second 17 current output of the second 12 voltage-current converters and connected to the inverting current input of the first 15 current-voltage converters, and the signal at the first 13 the output of the first 11 voltage-current converters is out of phase with the signal at the second 16 current output of the first 11 voltage-current converters, and the signal is at the first 14 current output of the second 12 voltage-current converters is out of phase with the signal at the second 17 current output of the second 12 the voltage-current converter and is in phase with the signals at the first 13 current output of the first 11 voltage-current converters.

The amplifier circuit of the prototype is shown in the drawing of figure 1.

The drawing of figure 2 presents a diagram of the inventive device in accordance with claim 1 and claim 2, and in the drawing of figure 3 - in accordance with claim 3 of the claims.

The drawing of figure 4 presents a simplified graphical representation of the proposed active adder 6.

The drawing of Fig. 5 shows a macro model of the inventive instrumental amplifier of Fig. 3 in a PSpice environment on models of components of the bipolar-field analog base matrix crystal ABMK_1_3, which was used to study its main characteristics

The drawing of Fig.6 and Fig.7 respectively shows the logarithmic amplitude and phase frequency characteristics of the differential voltage gain of the DUT of Fig.5 for various resistance values of the third (46) feedback resistor in the structure of the input precision transducer 1 of Fig.3, which determines the realized value this ratio.

On Fig shows the frequency dependence of the attenuation coefficient of the input common mode signal of the DUT of Fig.5 (Fig.3).

The drawing of Fig.9 shows graphs of the boundary stresses when applying a differential signal to the inputs of the instrumentation amplifier of Fig.5 (Fig.3).

The drawing of figure 10 shows the voltage values of zero drift at the output of the instrumentation amplifier of figure 5 (figure 3) when the temperature changes from 10 to 100 degrees Celsius.

The drawing of Fig.11 shows a histogram reflecting the possible values of the zero drift voltage of the instrumentation amplifier of Fig.5 (Fig.3) as a result of applying the Monte Carlo method (Gaussian distribution, the spread of resistor resistances of 1.5%).

The drawing of Fig. 12 shows graphs of the spread of the attenuation coefficient of the input common-mode voltage of the DUT of Fig. 5 (Fig. 3) using the Monte Carlo method (Gaussian distribution, spread of resistor values of 1.5%) in the frequency range.

The drawing of Fig.13 shows a histogram showing the possible values of the attenuation coefficient of the input common-mode signal.

The graphs of Fig. 11 - Fig. 13 show that the inventive device, in contrast to the DUT of Fig. 1, is characterized by a higher attenuation coefficient of the input common-mode signal, weakly dependent on the error of the resistive elements. This will avoid expensive laser precision tuning of resistors, and, therefore, increase the yield of products during production. In addition, the zero drift voltage of the DUT of FIG. 2, FIG. 3, determined by the zero bias voltage of the active adder 6, also has a low dependence on the error of the resistive elements.

The circuit of the inventive instrumental amplifier of figure 4 is characterized by high efficiency using the amplitude characteristics of the input precision Converter 1.

The instrument amplifier of Fig. 2 contains an input precision converter 1 of the first 2 and second 3 input voltage sources with the first 4 and second 5 antiphase outputs, an active adder 6, the inverting input of which 7 is connected to the first 4 output of the input precision converter 1, and the non-inverting input 8 is connected to the second 5 output of the input precision Converter 1, the output of the device 9, associated with the output of the active adder 6, and the first 2 and second 3 sources of input voltage are fed to the inputs of the input precision ion converter 1 relative to the common bus of power supplies 10. The active adder 6 contains the first 11 voltage-current converter, the inverting input of which is connected to the output of the device 9, and the non-inverting input is connected to the common bus of the power sources 10, the second 12 voltage-current converter ", The inverting input of which is connected to the inverting input 7 of the active adder 6, and the non-inverting input is connected to the non-inverting input 8 of the active adder 6, the first 13 current output of the first 11 voltage converter -current ”is connected to the first 14 current output of the second 12 voltage-current converters and connected to the non-inverting current input of the first 15 current-voltage converters, the second 16 current output of the first 11 voltage-to-current converters is connected to the second 17 current the output of the second 12 voltage-current converter and is connected to the inverting current input of the first 15 output of the current-voltage converter, and the signal at the first 13 current output of the first 11 voltage-current converter is out of phase with the signal at the second 16 current output of the first 11 voltage-current converter, and the signal at the first 14 current output of the second 12 voltage-current converter is out of phase with the signal at the second 17 current output of the second 12 voltage-current converter and in phase with the signals at the first 13 current the output of the first 11 voltage-current converter.

In the drawing of FIG. 2, in accordance with claim 2, the input precision converter 1 in a particular case contains the first 18 and second 19 operational amplifiers, the non-inverting input of the first 18 operational amplifiers is connected to the first 2 source of input voltage, the non-inverting input of the second 19 operating the amplifier is connected to the second 3 input voltage source, between the output of the first 18 operational amplifier, connected to the first 4 output of the input precision transducer 1 and the inverting input of the first 18 operational of the amplifier, the first 20 feedback resistor is turned on, between the output of the second 19 operational amplifier connected to the second 5 output of the input precision converter 1 and the inverting input of the second 19 operational amplifier, the second 21 feedback resistor is connected, and between the inverting inputs of the first 18 and second 19 operating amplifiers included a third 22 feedback resistor.

In the drawing of FIG. 3, in accordance with claim 3, the input precision transducer 1 comprises a third 28 voltage-current converter, the non-inverting input of which is connected to the first 2 input voltage source, and the fourth 29 non-inverting voltage-current converter the input of which is connected to the common bus of power supplies 10, the first 30 current output of the third 28 voltage-current converter is connected to the first 31 current output of the fourth 29 voltage-current converter and is connected to the inverting current in an ode of the second 32 current-voltage converter, the second 33 current output of the third 28 voltage-current converter is connected to the second 34 current output of the fourth 29 voltage-current converter and is connected to the non-inverting current input of the second 32 current-to-current converter voltage ”, and the signal at the first 30 current output of the third 28 voltage-current converter is out of phase with the signal at the second 33 current output of the third 28 voltage-current converter, and the signal at the first 31 current output of the fourth 29 the voltage-current converter is out of phase with the signal at the second 34 current output of the fourth 29 voltage-current converter and is in phase with the signals at the first 30 current output of the third 28 voltage-current converter, the output 35 of the second 32 of the current-voltage converter being connected with the first 4 output of the input precision converter 1, the fifth 36 voltage-current converter, the non-inverting input of which is connected to the second 3 input voltage source and connected to the inverting input of the third 28 voltage converter IE-current ", the sixth 37 voltage-current converter, the non-inverting input of which is connected to a common bus of power supplies 10, the first 38 current output of the sixth 37 voltage-current converter is connected to the first 39 current output of the fifth 36 voltage-current converter ”And is connected to the non-inverting current input of the third 40 current-voltage converter, the second 41 current output of the sixth 37 voltage-current converter is connected to the second 42 current output of the fifth 36 voltage-current converter and is connected to invert the current input of the third 35 current-voltage converter 1, and the signal at the first 38 current output of the sixth 37 voltage-current converter is out of phase with the signal at the second 41 current output of the sixth 37 voltage-current converter, and the signal at the first 39 the current output of the fifth 36 voltage-current converter is out of phase with the signal at the second 42 current output of the fifth 36 voltage-current converter and is in phase with the signals at the first 38 current output of the sixth 37 voltage-current converter, and the output is 43 third 4 0 of the output current-voltage converter is connected to the second 5 output of the input precision converter 1, the inverting input of the fifth 36 voltage-current converter is connected to the inverting input of the third 28 voltage-current converter, output 35 of the second 32 current-to-output converter voltage ”is connected to the non-inverting input of the fourth 29 voltage-current converter through the first 44 additional feedback resistor, output 43 of the third 40 of the current-voltage converter is connected to the inverting input a sixth inverter 37 house "voltage-current" through the second 45 additional feedback resistor, and between the inverting input 37 of the sixth and fourth transducers 29, "voltage-current" is included a third resistor 46 additional feedback.

Consider the operation of the DUT figure 2.

Signals containing in-phase and differential components are fed to input 1 (Bx.1) and input 2 (Bx.2) of the input precision transducer (1) and fed to the non-inverting inputs of the first (18) and second (19) operational amplifiers, in accordance with the selected ratio of the first (20) and third (22) feedback resistors and the second (21) and third (20) feedback resistors, the amplitude of the input differential signal is amplified, which together with the common-mode signal that is unchanged with respect to the input, and also error the zero bias voltage of the first (18) and second (19) operational amplifiers is fed to the output (4) and output (5) of the input two-channel precision transducer (1) and then to the inverting (7) and non-inverting 8 inputs of the active adder (6) and , respectively, to the inverting and non-inverting inputs of the second (12) voltage-current converter. As can be seen from the structure of the DUT, in the circuit, the incoming signals are subtracted according to the “current summation” principle, which can significantly reduce the error introduced in the final result by the zero bias voltage of the first (18) and second (19) operational amplifiers that are part of the input precision converter (1) subject to its implementation within a single crystal. In this case, the incoming common-mode voltage is suppressed due to the implementation of a high attenuation coefficient of the input common-mode signal in the input differential stages of the second (12) voltage-current converter. As a result, at the output of the active adder 6 and the instrumental amplifier as a whole, an amplified differential signal appears with an error introduced by the zero bias voltage of the active adder 6.

However, the inefficiency of using the amplitude characteristics of the first (18) and second (19) operational amplifiers in the structure of Fig. 2, which is associated with the presence of an in-phase component in their output signal, leads to inefficient use of the input boundary voltage range of the second (12) voltage-current converter ". This reduces the dynamic range of the DUT implemented by the circuit. Therefore, it is necessary to suppress the in-phase component of the input signal in the structure of the input precision transducer (1).

Consider the operation of the DUT 3.

A signal containing in-phase and differential components is fed to input 1 (Vx.1) and input 2 (Vx.2) of the input precision transducer (1) and is fed to the non-inverting input of the third (28) voltage-current converter, which inverts the fifth ( 36) voltage-current converter and to the inverting input of the third (28) voltage-current converter, non-inverting input of the fifth (36) voltage-current converter, respectively, due to the implementation of a high attenuation coefficient of the input common-mode signal in the input differential cial third stages (28) and fifth (36) transducer "voltage-current" is suppressed inphase signal component. At the same time, amplification of the amplitude of the differential component of the signal is performed according to the selected ratio of the first (44) and third (46) feedback resistors and the second (45) and third (46) feedback resistors. The amplified differential signal component together with the common-mode signal component weakened with respect to the input, as well as the error introduced by the zero drift voltage at the output (35) of the second (32) current-voltage converter and at the output (43) of the third (40) converter current-voltage ”is fed to the output (4) and output (5) of the input precision transducer (1) and then to the inverting (7) and non-inverting 8 inputs of the active adder (6), where the input signals are subtracted. This makes it possible to significantly reduce the error introduced in the final result by the zero-drift voltage of the DUT and increase the attenuation coefficient of the input common-mode voltage, provided that the input precision converter (1) is implemented within a single crystal and the amplified differential voltage is summed. As a result, at the output of the active adder 6 and the instrumental amplifier as a whole, an amplified differential signal appears with an error determined by the zero bias voltage of the active adder 6.

We show analytically that the above properties of the instrumental amplifier are implemented in the inventive scheme of figure 3.

Indeed, using the methods of analysis of electronic circuits, it can be shown that the proposed instrumental amplifier (figure 3) is characterized by the following parameters.

Differential signal gain:

$$\begin{array}{c}{K}_D=2\cdot \left(\mathrm{one}+\frac{R_{\mathrm{four}\mathrm{four}}}{R_{\mathrm{four}6}}+\frac{R_{\mathrm{four}5}}{R_{\mathrm{four}6}}\right)\cdot K_{d.6},\\ K_{d.6}=\frac{S_{\mathrm{one}2}K}{\mathrm{one}+S_{\mathrm{one}\mathrm{one}}K}=\frac{S_{\mathrm{one}2}}{S_{\mathrm{one}\mathrm{one}}},\end{array}(\mathrm{four})$$

where K _d.6 is the gain of the differential signal of the active adder (6), S ₁₂ is the slope of the second (12) voltage-current converter, S ₁₁ is the slope of the first (11) voltage-current converter (figure 3) . Therefore, in the circuit design of the input voltage-current converters of the active adder (6) (Fig. 3), it is necessary to provide, due to the integrated technology, a high accuracy ratio S ₁₂ and S ₁₁ . In this regard, with the correct implementation of the operating modes of the active elements of the circuit, the equality K _d.6 = R _and1 / R _{and2 is} almost exactly realized, which, if necessary, can be equal to unity. Then, the choice of the nominal value of the third (46) additional feedback resistor R ₄₆ , taking into account the resistance of the first (44) additional feedback resistor R ₄₄ and the second (45) additional feedback resistor R ₄₅ , sets the value of the parameter K _{d of the} instrument amplifier of Fig. 3. In particular, the equality R ₄₄ = R ₄₅ = R can be used. In this case, the common-mode voltage transfer coefficient of the instrumental amplifier:

$$K_{\mathrm{from}n}=\frac{R}{R_{\mathrm{four}6}}\cdot \left(K_{\mathrm{about}ccn.28}-K_{\mathrm{about}\mathrm{from}\mathrm{from}n.36}\right)\cdot K_{\mathrm{about}\mathrm{from}\mathrm{from}n.6},(5)$$

where K _{osn. 28} , K _{osn. 36} - attenuation coefficients of the input common-mode signal implemented in the input stages of the third (28) and fifth (36) voltage-current converters, respectively, K _{osn. 6} - attenuation coefficient of the input common-mode signal of the active adder (6). In this case, the voltage zero drift of the instrumental amplifier is determined by the ratio:

$$U_{dp.\mathrm{AND}\mathrm{At}}=\left(U_{dp.\mathrm{four}3}-U_{dp.35}\right)+U_{dp.6},(6)$$

where U _dr.IU - zero drift of the instrument amplifier, U _{dr. 35} , U _{dr. 43} - zero drift voltage at the output (35) of the second (32) current-voltage converter and at the output (43) of the third (40) converter "Current-voltage", respectively, U _dr.6 - drift voltage of the active adder (6). Given the implementation of the input precision Converter (figure 3) in a single technological process within the framework of a single crystal U _dr . ₃₅ = U _{dr. 43} , therefore:

$$U_{dp.\mathrm{AND}\mathrm{At}}=U_{dp.6}=K_{d.6}\cdot E_{\mathrm{from}m.\mathrm{one}2},(7)$$

where K _d.6 - gain of the differential signal of the active adder (6), E _{see 12} - EMF bias in the input circuits of the second (12) Converter "voltage-current" (figure 4). Since for the active adder of signals (6) K _d . _OUZ = 1:

$$U_{dp.\mathrm{AND}\mathrm{At}}=E_{\mathrm{from}m.\mathrm{one}2}.(8)$$

Thus, the voltage of the zero drift of the instrumental amplifier is determined by the emf bias of the active signal adder (6).

The voltage at the output (35) and output (43) of the second (32) and third (40) current-voltage converters are determined respectively by the formulas:

$${\mathrm{<figure-callout\; id="3"\; label="U"\; filenames="00000021.png,00000022.png"\; state="\{\{state\}\}">U</figure-callout>}}_{35}=K_{\mathrm{about}\mathrm{from}\mathrm{from}\mathrm{<figure-callout\; id="2"\; label="n"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">n</figure-callout>}28}\cdot U_{\mathrm{from}n}-\frac{R_{\mathrm{four}\mathrm{four}}}{R_{\mathrm{four}6}}\cdot U_d,$$

$$U_{\mathrm{four}3}=K_{\mathrm{about}\mathrm{from}\mathrm{from}\mathrm{<figure-callout\; id="3"\; label="n"\; filenames="00000021.png,00000022.png"\; state="\{\{state\}\}">n</figure-callout>}36}\cdot U_{\mathrm{from}n}-\frac{R_{\mathrm{four}5}}{R_{\mathrm{four}6}}{U}_d.(9)$$

Thus, when K _osni << 1, the efficiency of using the amplitude characteristics at the output (35) and output (43) of the second (32) and third (40) current-voltage converters increases, respectively (Fig. 3). Indeed, in the prototype device

$$U_{\mathrm{one}8}=U_{\mathrm{from}n}-\frac{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{20}}{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{22}}{U}_d,$$

$$U_{\mathrm{one}9}=U_{\mathrm{from}n}+\frac{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{2\mathrm{one}}}{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{22}}{U}_d.(\mathrm{one}0\text{)}$$

These voltages are determined by the common-mode voltage at the input of the instrument amplifier U _sn .

In addition, in the prototype

$$U_{dR.\mathrm{AND}\mathrm{At}}=E_{\mathrm{from}m.27}\left(\mathrm{one}+\frac{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{2\mathrm{four}}}{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{23}}\right)=2E_{\mathrm{from}m.27},(\mathrm{one}\mathrm{one}\text{)}$$

where E _{see 27} - voltage zero offset of the third (27) operational amplifier (figure 1), therefore, the zero drift of the prototype is almost 2 times greater than the zero drift of the claimed device.

The decrease in K _sn (5) due to K _osni << 1 is explained by the differential properties of the input main and additional differential stages.

Thus, the influence of technological errors in the manufacture of resistors Θ _Ri applies only to the differential gain (4) and does not affect the transfer coefficient of the common-mode signal (5) and its zero drift (8), which is also seen in the drawing in Fig. 11.

Indeed, as follows from relations (4), (5) and (8)

$$\delta K_d=\frac{\mathrm{one}}{K_d}\left(2\frac{R{l}_{\mathrm{four}\mathrm{four}}}{R_{\mathrm{four}6}}{\mathrm{<figure-callout\; id="2"\; label="\Theta \; R"\; filenames="00000021.png,00000024.png"\; state="\{\{state\}\}">\Theta \; </figure-callout>}}_{R_{}}+2\frac{R{l}_{\mathrm{four}5}}{R_{\mathrm{four}6}}{\Theta }_{R_3}-2\frac{R_{\mathrm{four}\mathrm{four}}+R_{\mathrm{four}5}}{R_{\mathrm{four}6}}{\Theta }_{R_{\mathrm{one}}}\right),(\mathrm{one}2)$$

$$\begin{array}{c}\Delta K_{\mathrm{from}n}=\frac{R}{R_{\mathrm{four}6}}\cdot \left(K_{\mathrm{about}\mathrm{from}\mathrm{from}n.28}-K_{\mathrm{about}\mathrm{from}\mathrm{from}n.36}\right)\cdot K_{\mathrm{about}\mathrm{from}\mathrm{from}n.6}\cdot \left({\Theta }_{R_{\mathrm{four}\mathrm{four}}}+{\Theta }_{R_{\mathrm{four}5}}\right)-\\ -\frac{R}{R_{\mathrm{four}6}}\cdot \left(K_{\mathrm{about}\mathrm{from}\mathrm{from}n.28}-K_{\mathrm{about}\mathrm{from}\mathrm{from}n.36}\right)\cdot K_{\mathrm{about}\mathrm{from}\mathrm{from}n.6}\cdot {\Theta }_{R_{\mathrm{four}6}}\end{array},(\mathrm{one}3)$$

$$\Delta U_{dR.\mathrm{AND}\mathrm{At}}=\Delta E_{\mathrm{from}m.\mathrm{ABOUT}\mathrm{At}3},(\mathrm{one}\mathrm{four})$$

the influence of Θ _{Ri is} attenuated by the introduced channels into the circuit of the claimed device. Moreover, ΔK _sn (13) is the limiting value of the transmission coefficient of the input common-mode signal.

In the circuit of the instrumental amplifier of FIG. 2 (claim 1), these parameters are determined by the following relations

$${\mathrm{TO}}_{\mathrm{from}n}={\mathrm{TO}}_{\mathrm{about}\mathrm{from}\mathrm{from}n.\mathrm{one}2}(\mathrm{one}5)$$

$$U_{dR.\mathrm{AND}\mathrm{At}}=U_{dR.6}=E_{\mathrm{from}m.\mathrm{one}2}+U_{\mathrm{from}n}\cdot K_{\mathrm{about}\mathrm{from}\mathrm{from}n.\mathrm{one}2}(\mathrm{one}6)$$

those. the common-mode voltage transfer ratio (ratio 15) is determined by the attenuation coefficient of the input common-mode voltage of the second (12) voltage-current converter, which also determines zero drift (ratio 16).

Thus, the proposed instrumental amplifier compares favorably with the prototype and analogs in that it has a higher (maximum achievable) common-mode signal attenuation coefficient, independent of resistive elements in the adder structure and weakly dependent on the error of resistive elements in the circuit in general, which avoids costly precision laser settings of these resistors. In addition, the proposed instrument amplifier is characterized by a more efficient use of the amplitude characteristic at the outputs of the input precision transducer (1) of Fig. 3, due to the suppression of the in-phase component in its structure, as well as by the low voltage of the zero drift of the instrument amplifier, which has a low dependence on the error of the resistive elements.

These theoretical conclusions are confirmed by the graphs of FIG. 6, FIG. 7, FIG. 8, FIG. 9, FIG. 10, FIG. 11, FIG. 12, FIG. 13.

Bibliographic list

1. Patent US20100259323 A1 fig. 1

2. Patent US 20110043281 A1

3. Patent US 20110043280 A1

4. Patent US 20070260150 A1

5. Patent US 20060267987 A1

6. Patent US 20050275460 A1

7. Patent US 20020163383 A1

8. Patent US20020113651 A1

9. Patent US 00008138830 B2

10. Patent US 00007952428 B2

11. Patent US 00007880541 B1

12. Patent US 00007728947 B2

13. Patent US 00007719351 B2

14. Patent US 00004490682

15. Patent US 00004206416

16. Patent US 00003453554

Claims (3)

1. An instrument amplifier for a precision analog-to-digital interface, comprising an input precision converter (1) of the first (2) and second (3) input voltage sources with the first (4) and second (5) antiphase outputs, an active adder (6), inverting whose input (7) is connected to the first (4) output of the input precision transducer (1), and the non-inverting input (8) is connected to the second (5) output of the input precision transducer (1), the output of the device (9) associated with the output of the active adder (6), with the first (2) and second (3) input voltage sources are supplied to the inputs of the input precision transducer (1) relative to the common bus of power supplies (10), characterized in that the active adder (6) contains the first (11) voltage-current converter, the inverting input of which is connected to the output of the device (9), and the non-inverting input is connected to a common bus of power sources (10), the second (12) voltage-current converter, the inverting input of which is connected to the inverting input (7) of the active adder (6), and the non-inverting input is connected to non-invert the input input (8) of the active adder (6), the first (13) current output of the first (11) voltage-current converter is connected to the first (14) current output of the second (12) voltage-current converter and is connected to a non-inverting current the input of the first (15) current-voltage converter, the second (16) current output of the first (11) voltage-current converter is connected to the second (17) current output of the second (12) voltage-current converter and is connected to inverting current input of the first (15) output current-voltage converter moreover, the signal at the first (13) current output of the first (11) voltage-current converter is out of phase with the signal at the second (16) current output of the first (11) voltage-current converter, and the signal is at the first (14) current output of the second (12) the voltage-current converter is out of phase with the signal at the second (17) current output of the second (12) voltage-current converter and is in phase with the signals at the first (13) current output of the first (11) voltage-current converter. 2. The instrumentation amplifier for the precision analog-to-digital interface according to claim 1, characterized in that the input precision converter (1) contains the first (18) and second (19) operational amplifiers, the non-inverting input of the first (18) operational amplifier is connected to the first ( 2) an input voltage source, the non-inverting input of the second (19) operational amplifier is connected to the second (3) input voltage source, between the output of the first (18) operational amplifier, connected to the first (4) output of the precision input converter (1) and the inverting input of the first (18) operational amplifier includes the first (20) feedback resistor between the output of the second (19) operational amplifier connected to the second (5) output of the input precision transducer (1) and the inverting input of the second (19) the operational amplifier includes a second (21) feedback resistor, and a third (22) feedback resistor is included between the inverting inputs of the first (18) and second (19) operational amplifiers. 3. The instrumentation amplifier for the precision analog-to-digital interface according to claim 1, characterized in that the input precision converter (1) contains a third (28) voltage-current converter, the non-inverting input of which is connected to the first (2) input voltage source, the fourth (29) voltage-current converter, the non-inverting input of which is connected to the common bus of power sources (10), the first (30) current output of the third (28) voltage-current converter is connected to the first (31) current output of the fourth ( 29) converter “Voltage-current” and is connected to the inverting current input of the second (32) current-voltage converter, the second (33) current output of the third (28) voltage-current converter is connected to the second (34) current output of the fourth (29 ) a voltage-current converter and is connected to a non-inverting current input of the second (32) current-voltage converter, and the signal at the first (30) current output of the third (28) voltage-current converter is out of phase with the signal at the second (33 ) current output of the third (28) converter Voltage-current, and the signal at the first (31) current output of the fourth (29) voltage-current converter is out of phase with the signal at the second (34) current output of the fourth (29) voltage-current converter and is in phase with the signals at the first ( 30) the current output of the third (28) voltage-current converter, and the output (35) of the second (32) current-voltage converter is connected to the first (4) output of the input precision converter (1), the fifth (36) converter voltage-current ”, the non-inverting input of which is connected to the second (3) input source voltages and is connected to the inverting input of the third (28) voltage-current converter, the sixth (37) voltage-current converter, the non-inverting input of which is connected to the common bus of power supplies (10), the first (38) current output of the sixth (37 ) the voltage-current converter is connected to the first (39) current output of the fifth (36) voltage-current converter and is connected to the non-inverting current input of the third (40) current-voltage converter, the second (41) sixth current output (37) voltage-current converter n with the second (42) current output of the fifth (36) voltage-current converter and is connected to the inverting current input of the third (35) current-voltage converter (1), and the signal is at the first (38) current output of the sixth ( 37) the voltage-current converter is out of phase with the signal at the second (41) current output of the sixth (37) voltage-current converter, and the signal at the first (39) current output of the fifth (36) voltage-current converter is out of phase with the signal at the second (42) current output of the fifth (36) voltage-current converter and common mode the signals at the first (38) current output of the sixth (37) voltage-current converter, wherein the output (43) of the third (40) current-voltage converter is connected to the second (5) output of the input precision converter (1), the inverting input of the fifth (36) voltage-current converter is connected to the inverting input of the third (28) voltage-current converter, the output (35) of the second (32) output current-voltage converter is connected to the non-inverting input of the fourth (29) voltage-current converter through the first (44) add feedback resistor, the output (43) of the third (40) output current-voltage converter is connected to the inverting input of the sixth (37) voltage-current converter through the second (45) additional feedback resistor, and between the inverting inputs of the sixth ( 37) and the fourth (29) voltage-to-current converter, a third (46) additional feedback resistor is included.

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