RU2335841C1 - High-voltage dc voltage converter with filter-compensating circuit and method of controlling its output power - Google Patents

Abstract

FIELD: electrical engineering.

SUBSTANCE: invention relates to power sources incorporating a converter of DC voltage into direct voltage for supplying loads requiring primarily high-voltage of some kV to several hundreds kV at power consumption of one to several hundreds kW. The said power sources can be used for supplying high-power tube amplifier in transmitting hardware, electronic beam smelters, electric filters intended for coal powder sedimentation at thermal electric power stations, charging devices etc. the known DC voltage converter the LC-circuit inductor is connected in series to the said transformer primary circuit while the LC-circuit capacitor is connected in parallel to the transformer secondary. The LC-circuit parameters are selected so that the relation between resonance frequency (Fres) of LC-circuit to frequency (f) of the inverter output voltage makes 1.8...3, with Fres/f=2 being the optimum value. The relation of the LC-circuit wave resistance approximates to the load base resistance varies from 1 to 1.5 with the load base resistance is determined as the relation between the rated voltage across the load and the load rated current. The aforesaid proposed circuit can be modified to incorporate several inverters with throttles and several transformers. Depending upon required output voltage, the number of open-output rectifiers connected in series and the number of secondaries of the transformers with speed-up capacitances. The open-output rectifier can be connected as a bridge rectifier, a rectifier with a doubler etc.

EFFECT: LC-circuit smaller size, higher efficiency of converter, lower production and operating, longer life.

2 cl, 7 dwg

Description

The claimed technical solution relates to electrical engineering, namely to power sources, incorporating a constant voltage converter of an electric power source to a constant voltage for supplying electric consumers, mainly requiring high-voltage power supply with voltage from units of kilovolts to several hundred kilovolts with power consumption from units to several hundreds kilowatt. Mentioned sources can be used for power supply: high-power tube amplifiers in transmission equipment, electron beam melting plants, electrostatic precipitators for the deposition of coal dust at thermal stations, chargers and other devices.

The structure of the DC / DC converter (DC / DC converter) is as follows: input capacitor - inverter - step-up (step-down) transformer - rectifier - output filter.

Typically, DC / DC converters for low voltages (up to approximately 1000 V) are designed with a choke connected between the output rectifier and the load. This allows you to get certain advantages, for example, improved harmonic composition of the voltage at the load.

However, for relatively high voltages from approximately 3 kV to 30 kV (as well as for higher voltages), the use of the aforementioned rectifier circuit with a choke is associated with a number of difficult to solve problems. The defining among them is the issue of overvoltage on rectifier valves. With the high-frequency conversion characteristic of DC / DC converters and the high-voltage version of the transformer (thick insulation, which leads to an increase in stray capacitance in the transformer), stray capacitance of the transformer begins to play a large role, causing oscillatory processes that accompany each switching of the keys of the input inverter. Since the output choke prevents the passage of high-frequency oscillations into the load, noticeable additions to the voltage of the secondary winding of the transformer appear, i.e. overvoltage. This is especially manifested in the intermittent current mode, when the so-called "ringing" occurs, caused by overcharging the stray capacitance of the secondary winding of the transformer.

The voltage level at the output rectifier in the circuit with the output inductor is additionally affected by changes in the input mains voltage (directly proportional).

It should be noted that in accordance with regulatory documents for electrical devices with rated voltages up to 1000 V, overvoltages equal to (1.5 ... 3) times the value of the nominal voltage are allowed. However, with an increase in the nominal voltage to several tens of kilovolts, the permissible overvoltage level decreases significantly, reaching a value of 5%.

The aforementioned circumstance prompts the use of a circuit with a rectifier shunted to the output by a capacitor (capacitor) for high-voltage DC / DC converters, in parallel with which a load is connected. This allows you to significantly reduce the level of overvoltage on the rectifier. Such a rectifier with a capacitor connected in parallel to the output is often referred to in the technical literature as an “open output rectifier”. This term is used hereinafter in the present description and claims.

A known Converter (1) DC voltage obtained by rectifying the voltage of an industrial network, into a constant high voltage voltage containing a bridge transistor inverter, the output of which is connected in series with a resonant oscillatory LC circuit connected to the primary winding of a step-up transformer. A rectifier with an open output is connected to the secondary winding of the said transformer, i.e. the rectifier output is bridged by a capacitor. The converter also contains a control system. The disadvantage of this device is the high reactive power of the elements of the resonant oscillatory LC circuit, as well as the aforementioned “ringing" caused by overcharging of the parasitic capacitance of the secondary winding of the transformer, which occurs in the mode of intermittent currents.

There is also a technical solution (2) and its further development - a DC-voltage converter (3), containing a bridge transistor inverter, to the output of which a primary winding of a step-up transformer is connected in series with a resonant oscillatory LC circuit. A rectifier with an open output is connected to the secondary winding of the said transformer, i.e. the rectifier output is bridged by a capacitor. The converter also contains a control system.

This technical solution (3) is the closest to the claimed technical solution in its essence and technical result.

The disadvantage of the DC-voltage converter (3) is the high reactive power of the elements of the resonant oscillatory LC circuit, as well as the large (more than 3) current ratio (the ratio of the current amplitude to the average current value over a half-cycle), which complicates and increases the cost of an LC circuit, and the device as a whole. In addition, as will be shown below, in the known converter (3), when switching transistors of a bridge inverter in the high load power mode, there are additional losses in the transistors due to the fact that they are switched on to the counter-conducting reverse diode of the transistor of the same arm, the switching current of which passing through an included transistor and causes the aforementioned losses.

The problem to which the claimed technical solution is directed is to reduce the reactive power of the LC circuit elements and the current multiplicity with respect to the technical solution (3), reduce the current load of the converter elements and losses in them.

In solving this problem, the technical result achieved is to reduce the dimensions of the resonant oscillatory LC circuit, increase the efficiency of the converter, reduce its cost and the level of costs for ongoing maintenance, increase the service life of the converter.

In accordance with the proposed technical solution, this problem is solved by the fact that in the known high-voltage DC-voltage converter, comprising: an inverter, an oscillatory resonant LC circuit, a transformer, a rectifier with an open output and an inverter control device containing current and voltage sensors; said inverter, which converts an unstable, widely varying DC voltage of an electric power source into an alternating voltage of increased frequency (inverter output voltage) is made, for example, according to a single-phase bridge circuit, each of whose two arms is formed by upper and lower controlled valves with reverse conductivity, for example IGBTs connected in series, the contact connecting the upper and lower valves of the first arm of the inverter forms the first, and a similar contact of the second The inverter’s cha L) the oscillatory resonant LC circuit is connected in series to the primary circuit of the said transformer, and capacitively element (C) LC-oscillating resonant circuit is connected parallel to the secondary winding of said transformer;

the parameters of the vibrational resonant LC circuit are such that the ratio of the resonant frequency (Fres) of the vibrational resonant LC circuit to the frequency (f) of the inverter output voltage is 1.8 ... 3, and the value Fres / f = 2 is optimal, and the wave the resistance of the vibrational resonant LC circuit to the base load resistance is in the range 1 ... 1.5, and the base load resistance is calculated as the ratio of the rated voltage on the load to the rated current of the load.

In accordance with the proposed technical solution, this problem is also solved by the fact that in the known method of controlling the output power of the aforementioned high-voltage DC-voltage converter, at which the output current frequency is kept constant and approximately equal to half the resonant frequency of the vibrational resonant LC circuit, and the output current half-cycle is formed by pulse width modulation using the current recovery state or current zero state, then determine the moment the output current drops to zero, after which the current pause is maintained and then the next half-cycle of the output current is formed, performing actions similar to those described for the previous half-cycle by symmetry, according to the claimed technical solution, the half-period of the inverter output current is generated by pulse-width modulation as follows: include the upper valve of the first shoulder and the lower valve of the second shoulder, after a period of time determined by the control system depending on the required equal power, turn off the upper valve of the first shoulder (or the lower valve of the second shoulder), turning the circuit into a state of zero current, then calculate the duration of the current pause, i.e. the time interval between the beginning of the next half-cycle of the output current and the moment the output current drops to zero, if the duration of the calculated current pause is shorter than the "dead time", then the lower valve of the second arm (or the upper valve of the first arm) is turned off at the instant of time calculated by the control system ), transferring the circuit to the recovery state to increase the duration of the current pause to a predetermined value, and then form the next half-cycle of the output current, including the upper valve of the second arm and lower the first valve of the first arm and then performing symmetrical actions similar to those described above for the previous half-period of the output current.

Figure 1 presents the electric circuit of the DC-DC Converter, made in accordance with the United States Patent No. 4680693 (1) and in accordance with the application WO 2006/079985 A2 (3).

Figure 2 presents the electrical circuit of the inventive DC-DC converter with filter compensating circuit.

On figa, 3b, 3C presents a diagram illustrating the state of inclusion of the elements of the Converter, made in accordance with the application WO 2006/079985 A2 (3).

On figa, 4b, 4C shows the timing diagram of the output voltage and current of the inverter, as well as the voltage across the capacitor of the resonant oscillating LC circuit in the Converter, made in accordance with the application WO 2006/079985 A2 (3).

On figa, 5b, 5c, 5d presents a timing diagram of the signals that control the on and off semiconductor switches of the inverter, as well as timing diagrams of the output voltage and current of the inverter, as well as the voltage across the capacitor of the LC circuit in the inventive filter-compensated converter circuit.

On figa, 6b, 6c, 6d presents a diagram illustrating the state of the inclusion of elements and the path of the flow of currents in the inventive Converter with filter filter circuit.

In FIG. 7a and 7b show two of the many possible embodiments of a filter-compensating converter.

Below is a brief description of the device and the operation of the Converter, made in accordance with the application WO 2006/079985 A2 (3).

The circuit of the DC-DC converter, made in accordance with the technical solution (3) is presented in figure 1.

The converter of direct voltage of an electric power source to direct voltage for power supply of electric consumers, made in accordance with technical proposal (3), includes: a device for converting direct voltage of an electric power source to alternating voltage of increased frequency f, containing a bridge inverter 1 made according to a single-phase bridge circuit , each of the two shoulders of which is formed by upper and lower controlled valves with reverse conductivity, for example IGBT, connected by flax. In Fig. 1, the inverter valves are represented as semiconductor switches (transistors) 2, 3, 4, and 5. The filter capacitor 6 is connected to the DC terminals “-” and “+” of the inverter 1 for connecting to the aforementioned electric power source. The primary terminals of the transformer 9 are connected to the terminals of the alternating current of the inverter 1 in series with a resonant oscillatory LC circuit consisting of a reactor 7 (inductive element) and a capacitor 8 (capacitive element). A high-voltage bridge rectifier 10 with an open output is connected to the secondary winding of the transformer 9 consisting of valves 11, 12, 13, 14 and an output filter capacitor 15 connected to the DC outputs of the rectifier 10.

Each of the valves 11, 12, 13, 14 may consist of several semiconductor diodes connected according to the value of the output voltage (usually in series). The capacitor 15 can also be formed by the corresponding connection of several capacitors. There are control devices in the circuit (not shown in the diagram). The circuit is designed and operates in such a way that the oscillatory resonant LC circuit is tuned to the second harmonic frequency of the aforementioned increased frequency f.

When operating the inverter 1 with a circuit connected to it, consisting of a series-connected inductor 7, a capacitor 8 (resonant oscillating LC circuit) and the primary winding of the transformer 9, in accordance with technical solutions (2) and (3), to generate the primary current I windings of the transformer 9 uses three states of inclusion of the elements of the inverter 1.

The first state, conventionally called its “conducting state” (see figa), involves the inclusion of keys 2 and 4 (conductive state) for a positive half-wave of the average current value I of the primary winding of the transformer 9 and keys 3 and 5 for the negative half-wave of the average current value I of the primary winding of the transformer 9. The marked state is shown in the time diagram of FIGS. 4a and 4b and occupies the time period T1 at the half-cycle equal to 1 / (2 · f) corresponding to the switching frequency f of the key elements of the inverter 1.

The second state, let's call it “current recovery state” (see fig. 3b), is realized when the keys 2 and 4 for the positive half-wave of current I (or 3 and 5 for the negative half-wave of current I) are turned off. It corresponds to the length of time T2 in figa. The current I of the primary winding of the transformer 9 in this state charges the capacitor 6, passing through the reverse diodes of the transistors 3 and 5 for the positive half-wave of the current I (or through the reverse diodes of the transistors 2 and 4 for the negative half-wave of the current I).

The third state, let's call it “current zeroing state” (it corresponds to the time interval T2 in Fig. 4b), occurs when one of the transistors, for example, transistor 2, is turned off. Since inverter 1 is a converter made in accordance with technical solutions (2) and (3), the current during the half-cycle is oscillatory in nature, to ensure that the current flows in the negative direction, some time after the transistor 2 is turned off, the transistor 3 is turned on. The current circuit in its positive direction is shown in FIG. 3 . The current passes through the transistor 4 and the reverse diode of the transistor 3. The current loop in its negative direction (not shown in FIG. 3c) coincides with the current loop in its positive direction, with the difference that the current passes through the transistor 3 and the reverse diode of the transistor 4.

In accordance with technical solutions (2) and (3), when using a bridge inverter 1 with variable output power, various combinations of the above states are used. On figa, 4b, 4C shows the timing diagram of the output voltage Uab and current I of the inverter 1, as well as the voltage Uc on the capacitor 8 in modes with different levels of output power of the Converter.

In the mode of small output power, the duration of the conductive state T1 of the keys 2 and 4 (or 3 and 5) is regulated. So, for example, in Fig. 4a, at the end of the time interval T1, the keys 2 and 4 (or 3 and 5 for the negative half-wave) are turned off. The recovery state begins with a duration of T2. The current I of the primary winding of the transformer 9 in this state, decreasing, charges the capacitor 6, passing through the reverse diodes of the transistors 3 and 5 for the positive half-wave of the current I (or through the reverse diodes of the transistors 2 and 4 for the negative half-wave of the current I). At the moment the current I passes through the zero value, the recovery state begins with the reverse direction of the current I (due to the oscillation of the process) with a duration of T3. In this state, the current I has the form of a half-sine wave and passes through the inverse diodes of transistors 2 and 4 (or 3 and 5 for the negative half-wave). At the end of the interval T3, the current I drops to zero and the half-cycle ends with a current pause of duration T4. Similar processes occur in another half-cycle, differing in reverse polarity.

In all such modes (with a small output power), the duration of the conducting state T1 of the keys 2 and 4 (or 3 and 5) is controlled at a constant switching frequency f of the key elements of the inverter 1, approximately equal to half the resonant frequency of the LC circuit.

In phase T2, instead of the recovery state (as in FIG. 4a), a grounding state as shown in FIG. 4b can be used.

When the bridge inverter 1 is operating with a high level of output power, in accordance with technical solutions (2) and (3), the duration of the conducting state T1 of keys 2 and 4 (or 3 and 5) is kept at maximum, and the level of output power is controlled by changing the frequency f. Fig. 4c shows timing diagrams of the output voltage Uab and current I of the inverter 1, as well as the voltage Uc on the capacitor 8 in the mode with a high level of converter output power. Fig. 4c shows two half-periods of the output current I of inverter 1, the positive half-cycle of the current being provided by switching on the keys 2 and 4, and the negative by turning on the keys 3 and 5. As can be seen from Fig. 4c, at the end of the positive half-period of the current I of inverter 1, the formation of a negative half-period the current I of the inverter 1 (i.e., the inclusion of keys 3 and 5) begins at the moment when the current I has a negative value within the positive half-cycle. This means that immediately before the keys 3 and 5 (or one of them) are turned on, the current I flows in the inverter 1 (see Figs. 1 and 3, where Fig. 3 shows the path of the positive currents) through the inverse diodes of the transistor switches 2 and 4 (if the current recovery state is used) either through the diode of the transistor switch 2 and the transistor 5 or through the diode of the transistor switch 4 and the transistor 3 (if the current ground state is used). When the transistors 3 and 5 (or one of them) are subsequently turned on, which is necessary for the formation of a negative half-period of the current I of inverter 1, the transistor 3 is turned on to the conducting reverse diode of transistor 2 and the transistor 5 is turned on to the conducting reverse diode of transistor 4 of the same arm, t. e. when the transistor is turned on, the off current (reverse current) of the diode flows through it. This causes additional losses when the transistors are turned on ("ON"), which degrades the efficiency of the device.

The disadvantage of the technical solution (3) is that during each half-cycle 1 / (2 · f) corresponding to the switching frequency f of the key elements of the inverter 1, the output current curve I of the inverter 1 along with the positive half-wave also contains a negative half-wave. To ensure the necessary level of the average current value for a half-period, the positive half-wave must contain a component that compensates for the effect of the negative half-wave. This results in increased current loading of semiconductor switches 2, 3, 4, 5 and an increased value of reactive power passing through elements 7 and 8 of the LC circuit. In addition, as can be seen from the diagrams in figure 4, at the end of the positive half-wave of the current I there is a rise in voltage Uc on the capacitor 8, due to the oscillatory process in the LC circuit.

Additional losses, increased values of currents, voltages and powers in the key elements 2, 3, 4, 5 of inverter 1 and elements 7 and 8 of the resonant LC circuit lead to the need to increase their installed capacities, which negatively affects the overall dimensions and cost performance of the converter.

To overcome these shortcomings contributes to the claimed technical solution.

The device of the claimed technical solution in its static state can be described according to the scheme in figure 2. As can be seen from the comparison of the circuits in Figs. 1 and 2, the main and, as shown below, significant difference of the claimed technical solution is the connection of the capacitor 8 of the resonant resonant LC circuit parallel to the secondary winding of the transformer 9. With this connection, the possibility of high-frequency oscillations of arbitrary frequency is effectively eliminated and the amplitude is “ringing”, since in parallel with a small parasitic capacitance 16 (in FIG. 1), the value of which depends on the insulation state of the transformer 9, a larger Shai capacitor 8, which determines the operation mode of the secondary winding circuit of the transformer 9. In this case, as shown below, LC-circuit can be characterized by the term "filter-circuit". In addition, a significant feature of the claimed technical solution is that the parameters of the vibrational resonant LC circuit are such that the optimum value of its resonant frequency is equal to the second harmonic frequency of the aforementioned alternating voltage of increased frequency, and the wave resistance of the vibrational resonant LC circuit ensures the flow of the rated load current at minimum voltage source of electricity.

The operation of the inventive converter with a filter compensating circuit is illustrated using FIGS. 5 and 6. FIG. 5 shows timing diagrams of the signals g2, g3, g4, g5 controlling the on and off of the corresponding semiconductor switches 2, 3, 4, 5 of inverter 1 (FIG. 5a and 5c), as well as timing diagrams of the output voltage Uab and current I of the inverter 1, as well as the voltage Uc on the capacitor 8. FIGS. 6a, 6b, 6c, 6d are diagrams illustrating the switching states of the elements and the current paths in the converter.

In the inventive converter with a filter compensating circuit, the switching states of the inverter elements are similar to those in the inverter of the technical solution (3), however, the electromagnetic processes are different.

The inventive device operates as follows. The semiconductor switches of the inverter 1 operate at a frequency f below the resonant frequency of the LC circuit. The frequency f is approximately half the resonant frequency of the LC circuit. At the beginning of the half-cycle equal to 1 / (2 · f), the transistors (for example, 2 and 4) of inverter 1 (the interval t1 in Fig. 5b, 5d) open. The current I of the inverter 1 increases and energy is accumulated in the inductor 7. At the beginning of the interval t1, the current increases forcibly because the voltage of the capacitor 8 to which it was charged at the end of the previous period coincides in sign with the input voltage. The capacitor 8 is recharged. The voltage on it is less than the output voltage of the rectifier 10, determined by the capacitor 15, and the diodes 11, 12, 13 and 14 of the rectifier 10 are closed. The contours of the flow of currents in the Converter for the time interval t1 shown in figa. At the end of the time interval t1, the voltage across the capacitor 8 becomes equal to (slightly more by the magnitude of the drop across the diodes of the rectifier 10) the output (voltage across the capacitor 15). The diodes of the rectifier 10 corresponding to the sign of the voltage across the capacitor 8 open and current flows into the load. During the time interval t2, the current I of the inverter 1 remains constant or increases somewhat (see Fig. 5b, 5d) depending on the accepted design parameters of the circuit. Accordingly, the energy stored in the inductor 7 remains constant or increases. The current flow paths in the converter for the time interval t2 are shown in FIG. 6b. At the end of the time interval t2, one of the transistors (for example, transistor 2) is turned off, the circuit goes into a state of zero current, which continues for the time interval t3 (see fig. 5b, 5d). The current paths in the converter for the time interval t3 are shown in FIG. 6c. The energy in the load comes from the energy stored in the inductor 7.

Depending on the operating mode of the converter, determined by the load power level, intermittent current mode or boundary mode is possible.

At the beginning of the interval t3, the circuit goes into a state of zero current, then the control system, acting according to the method of “conditional prediction of mismatch” (USSR AS No. 1480066, 05/15/89, Bull. No. 18), determines the duration of the time interval t4, during of which the current I of the inverter 1 must drop to zero (see Fig. 5d), so that at the end of the period there is "dead time" to exclude the through-loop mode of the inverter 1 arm.

With a small output power at the beginning of the interval t3, the circuit is transferred to the current zero state, which remains until the end of the interval t3, the current I of the inverter 1 drops to zero and a current pause of duration t4 occurs (see Fig. 5b), significantly exceeding the dead time time ”(intermittent current mode).

With a significant output power, if the control system determines that the time interval t4 is too long and there is no margin for “dead time”, the control system calculates the point in time at which the circuit must be transferred to the recovery state and at that moment turn off all transistors . This leads to a sharper decrease in current I and a shortening of the time interval t4, which allows providing a margin of time ("dead time") for switching off the semiconductor switches that form the previous half-cycle of current I before turning on the semiconductor switches that form the next half-cycle of current I. In Fig.5d the time interval corresponding to dead time is not shown.

The contours of the flow of currents in the Converter for the time interval t4 shown in Fig.6d.

Similar processes occur in another half-cycle, differing in reverse polarity.

As can be seen from FIGS. 5b and 5d, during the half-cycle, a change in the direction of current flow I of the inverter 1 and overcharging of the capacitor 8 is excluded. This provides a significant reduction in the values of currents, voltages and powers in the key elements 2, 3, 4, 5 of inverter 1 and elements 7 and 8 resonant LC circuit in comparison with the technical solution (3).

In the proposed technical solution, cases of turning on an inverter 1 transistor to a diode conducting towards it are excluded, as is the case in the known technical solution (3).

Thus, the proposed converter with a filter-compensating circuit can significantly reduce the installed power of the converter elements and, as a result, reduce the overall dimensions and cost parameters of the converter.

For specific applications, the filter-compensating converter circuit shown in FIG. 2 may be modified in various ways depending on the values of the output voltage and power. Depending on the power, a different number of inverters 1 with reactors 7 and transformers 9 can be used. Depending on the required output voltage, the number of rectifiers 10 with an open output connected in series can vary and the number of secondary windings of transformers 9 with boost capacitors 8 can vary. Also can vary rectifier circuit 10 with open output: bridge rectifier (as in FIG. 2), rectifier according to the scheme of doubling, quadrupling, etc. Two of the many mentioned embodiments of the filter-compensating converter are shown in FIGS. 7a and b.

Information sources

1. United States Patent No. 4680693, Jul., 14, 1987, Int. Cl. ⁴ H02P 13/20. US C1. 363/98, 363/132. Continuous high DC voltage supply particularly for an X-ray e mitter tube. (A source of continuous constant high-voltage voltage mainly for an x-ray tube).

2. Application WO 01/37416 A2, (43) International Publication Date 25 May 2001, Int. C1. ⁷ H02M 7/00, Power supply unit including an inverter.

3. Application WO 2006/079985 A2, (43) International Publication Date 3 August 2006, (51) Int. Patent Classification: Not classified, Power supply and method for operating a power supply.

Claims (2)

1. A high voltage DC / DC converter, comprising: an inverter, an oscillatory resonant LC circuit, a transformer, an open-output rectifier, and an inverter control device comprising current and voltage sensors; said inverter, which converts an unstable, widely varying DC voltage of an electric power source into an alternating voltage of increased frequency (inverter output voltage) is made, for example, according to a single-phase bridge circuit, each of whose two arms is formed by upper and lower controlled valves with reverse conductivity, for example IGBTs connected in series, the contact connecting the upper and lower valves of the first arm of the inverter forms the first, and a similar contact of the second the inverter forms the second AC terminals intended for connecting the elements of the resonant resonant circuit and the primary winding of the said transformer, to the secondary winding of which a high-voltage rectifier with an open output is connected (a rectifier with a capacitor connected to the output terminals of the rectifier), characterized in that the inductive element ( L) the oscillatory resonant LC circuit is connected in series to the primary circuit of the said transformer, and the capacitive element (C) oscillates a LC resonant circuit is connected parallel to the secondary of said transformer; the parameters of the vibrational resonant LC circuit are such that the ratio of the resonant frequency (Fres) of the vibrational resonant LC circuit to the frequency (f) of the inverter output voltage is 1.8 ... 3, and the value Fres / f = 2 is optimal, and the wave the resistance of the vibrational resonant LC circuit to the base load resistance is in the range 1 ... 1.5, and the base load resistance is calculated as the ratio of the rated voltage on the load to the rated current of the load. 2. The method for controlling the output power of the high-voltage DC-voltage converter according to claim 1, wherein the frequency of the output current is kept constant and approximately equal to half the resonant frequency of the oscillating resonant LC circuit, and the half-period of the output current is generated by the pulse width modulation method using the recovery state current or state of zero current, then determine the moment the output current drops to zero, after which they withstand a current pause and then form the following the half-period of the output current, performing symmetrical actions similar to those described for the previous half-period, characterized in that the half-period of the inverter output current is generated by the pulse-width modulation method as follows: turn on the upper valve of the first arm and the lower valve of the second arm, after a period of time determined by the system control depending on the required power level, turn off the upper valve of the first shoulder (or the lower valve of the second shoulder), putting the circuit in a state of zero current, then m calculated current pause duration, i.e. the time interval between the beginning of the next half-cycle of the output current and the moment the output current drops to zero, if the duration of the calculated current pause is shorter than the "dead time", then the lower valve of the second arm (or the upper valve of the first arm) is turned off at the instant of time calculated by the control system ), transferring the circuit to the recovery state to increase the duration of the current pause to a predetermined value, and then form the next half-cycle of the output current, including the upper valve of the second arm and lower the first valve of the first arm and then performing symmetrical actions similar to those described above for the previous half-period of the output current.

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