RU2262078C2 - Method and device for neutralizing capacitance of coupling of differential shift transformer with inaccessible movable electrode - Google Patents

Abstract

FIELD: measurement technology.

SUBSTANCE: method and device can be used in measuring circuits of float-type rotametric liquid and gaseous media consumption transformers with capacitance differential float shift transformer where float has to be movable electrode and is inaccessible for direct connection to measuring circuit. Capacitance of capacitive differential shift transformer is charged from dc sources with currents being equal in value but having different polarities. Equality of charging currents is kept non-changeable during whole working cycle of transformation. Due to this, complete compensation of charging currents is performed in capacitance of coupling of capacitive differential transformer, which results in elimination of effect of capacitance on result of transformation in circuits with pulse-width transformation.

EFFECT: improved efficiency of operation.

3 cl, 3 dwg

Description

The invention relates to measuring technique and can be used to build measuring circuits of float rotametric flow transducers of liquid and gaseous media with a capacitive differential transducer displacement of the float.

Capacitive differential displacement transducers with a movable common electrode are widely used in measurement technology due to their high sensitivity, good linearity of the conversion function, low power consumption and negligible efforts arising on the movable electrode due to electrostatic forces [1-3].

However, in a number of practically important cases (float level gauges, rotametric flow sensors of liquids and gases with a capacitive differential transducer for displacing the float, etc.), the movable electrode of the capacitive differential transducer is not available for connecting a current lead to it. In addition, the presence of a current lead would inevitably exert an additional parasitic force on the movable electrode, which reduces the accuracy of the converter. In these cases, the electrical connection with the movable electrode is due to the communication capacitance formed by the lateral surface of the electrically conductive float and the electromagnetic shield (transducer housing). In particular, a similar design of a capacitive differential converter is described in [1] on p.47. An equivalent circuit of such a converter is shown in FIG. On it, differential capacitances, depending on the position of the movable electrode, are designated as C1 and C2, and the communication capacitance is C3. Moreover, the available electrode of the communication capacitance is usually grounded, simultaneously performs the functions of an electromagnetic screen and a converter housing. When powering such a converter from a harmonic current source and including it in a bridge measuring circuit (as shown in FIG. 2), the absence of galvanic contact with the movable electrode and the presence of a communication capacitance do not cause problems. However, such a measuring circuit is very sensitive to stray distributed capacitances, especially for small values of the differential capacitance of a capacitive differential transducer, it requires very fine individual tuning (using the tuning capacitor C5 included in the bridge circuit) and is inconvenient for using digital measurement methods. If you try to include such a converter in the measuring circuit with the conversion of the measured value into the time interval (circuits with pulse-frequency or pulse-width conversion), the presence of the communication capacitance C3 leads to serious complications. This is due to the fact that the potential of the common point of such a converter (point C in FIG. 2) does not remain unchanged during the conversion cycle, which leads to strong negative feedback through this capacitance. Therefore, changes in the differential capacitances C1 and C2 have practically no effect on the output signal of the measuring circuit. Similar problems arise when trying to use self-generated measuring circuits with a frequency output, since the communication capacitance C3 is included in the frequency-setting circuits of both self-oscillators (in series with one of the differential capacities). This leads to such a strong connection between the oscillators, in the circuits of which the capacitances C1 and C2 of the capacitive differential converter are included, that they are mutually synchronized and their frequency practically does not depend on the ratio of the capacities C1 and C2.

The technical problem to which the invention is directed is to neutralize the effect of the coupling capacitance of a capacitive differential displacement transducer with an inaccessible movable electrode when using measuring circuits with pulse-width conversion.

This problem is solved by providing full compensation of charging currents on the communication capacitance when changing the magnitude of the charges of differential capacities. This is achieved by the fact that the charge of the differential capacitances C1 and C2 is equal in magnitude and opposite in sign to the currents, which ensures their mutual compensation in the communication capacitance C3, and therefore the absence of charge on it during the formation of the duration of time intervals depending on the values of the differential capacities of the transducer C1 and C2. This method is implemented as follows. At the initial moment, all capacities of the converter must be completely discharged. This can be easily done using analog keys connected to points A and B of the capacitive differential transducer, and when they open these points short to ground. Further, these keys are simultaneously closed and the differential capacitance C1 and C2 of the capacitive differential converter starts charging from the DC sources of the same magnitude but of opposite polarity connected to points A and B of this converter. In this case, the voltages at the differential capacitances C1 and C2 will linearly change in opposite directions with speeds inversely proportional to the capacitances C1 and C2. The voltage at the communication capacitance C3 will remain zero, because equal in magnitude currents of opposite polarity will flow through it, compensating each other. The voltages at points A and B using comparators are compared with threshold values given from sources of a reference voltage of equal magnitude and opposite polarity (the same sources of reference voltage can be used to form the aforementioned current sources). The first to work is the comparator that is connected to the differential capacitance, which is currently of lesser importance. (The polarity of the threshold voltages supplied to the comparators must correspond to the polarity of the voltages at the corresponding differential capacitors when they are charged). When the threshold voltage value is reached at the second differential capacitance (which is currently of greater importance), a differential pulse is generated that opens the analog keys connected to points A and B of the capacitive differential converter and at the same time the formation of the output pulse ends, the width of which will be determined by the difference of moments triggering comparators. To preserve the conditions for compensation of currents in the communication capacitance C3, the linear region of the charge of the smaller of the differential capacitances at its minimum value should be greater than the region of the linear charge of the larger of the differential capacitances (which will have the maximum possible value). In this case, the current flowing through the smaller of the differential capacitances will remain unchanged even after the voltage on it reaches a threshold value. This condition will be met when choosing the absolute value of the thresholds of the comparators much less than the maximum possible (in absolute value) output voltage of the current sources. While maintaining the constancy of charging currents, the time from the initial moment to the operation of the comparator connected to a differential capacitance with a lower value of T ₁ and the duration of the period T _p determined by the charge time of a larger capacity can be calculated by the formulas

where C _m - the smaller of the differential capacitance;

C _b - the largest of the differential capacitance;

U ₀ - the absolute value of the reference voltage;

I _C - the absolute value of the charging currents of the current sources.

If we connect the outputs of the comparators with the inputs of the trigger, then the pulse duration T _and generated by this trigger will be equal to:

Denoting by C _min and C _{max the} minimum and maximum possible values of the differential capacitances C1 and C2 and by C _{cp the} average value of the differential capacitances corresponding to their equality

C1 = C2 = C _cp ,

we get

T _{and min} = 0 for C1 = C2 = C _cp ;

T _{and max} = (C _max -C _min ) U ₀ / I _s .

To avoid ambiguity frame (T _and will be the same with opposite parities differential capacitance: C1 / C2 = C2 / C1) is sufficient to determine which of the comparators load the first - with a positive or negative threshold voltage, that is simple to implement both hardware and software ( the latter is possible when constructing a measuring circuit based on a microprocessor).

This method allows you to measure the relative difference of the differential capacitance

It is sufficient to determine the ratio of pulse duration T _and duration T _P for the period defined by (2), i.e. time from the beginning of the charge of the tanks to the operation of the comparator connected to the larger of the differential tanks.

This method can be implemented both by traditional hardware and using a microprocessor using a pulse-counting method of measuring time intervals. A microprocessor-based measuring circuit is more economical, flexible, and convenient to use, therefore, we will consider it as an example of the implementation of this method.

The microprocessor-based block diagram implementing the described method is shown in FIG. 3. Capacitive differential transducer 1, consisting of differential capacitances C1 and C2 and communication capacitance C3 is highlighted by a dashed rectangle. The inputs of the differential capacitances C1 and C2 are connected to analog switches 2 and 3 and to current sources 4 and 5 of the same magnitude but of different polarity, as well as to the inputs of the respective comparators 7 and 8. The accessible electrode of the communication capacitance C3 is grounded. The second inputs of comparators 7 and 8 are connected with reference voltages + U ₀ and -U _{0 of the} same magnitude, but of different polarity from the source of reference voltages 6. The same reference voltages are used to form sources of bipolar current 4 and 5, which in this case are amplifiers with deep negative current feedback. The outputs of the comparators 7 and 8 are connected to the two-input circuit “And” 9 and to the inputs of the microprocessor 11. The output of the two-input circuit “And” is connected to the input of the single-shot 10, the output of which is connected to the control inputs of the analog keys 2 and 3.

The scheme operates as follows. In the initial state, all capacitances of the capacitive differential converter 1 are completely discharged, since the analog keys 2 and 3 are open. At the initial moment of time, the analog switches 2 and 3 are closed and the differential capacitances C1 and C2 of the capacitive differential transducer 1 begin to be charged from bipolar sources of direct current 4 and 5. Since the charging currents are equal in magnitude but have opposite polarity, then summing up on the communication capacitance C3 capacitive differential transducer 1, they cancel each other, which means that the capacitance C3 remains completely discharged. At the same time, the differential capacitances C1 and C2 are charged with these currents linearly with speeds inversely proportional to the values of these capacities, with capacitance C1 being charged with positive polarity, and capacitance C2 with negative. The charge voltage at the capacitor C1 is compared by the comparator 8 with the reference voltage + U ₀ , and the charge voltage at the capacitor C2 is compared by the comparator 7 with the reference voltage -U ₀ . The first to work is a comparator connected to a smaller capacity, because It will charge faster. But even after the comparator is activated, this capacity should continue to be charged linearly until the second comparator is connected, connected to a larger capacity. Only when the second comparator works, the two-input circuit “I” (9) will open and the signal will go to a single-shot 10, producing a short rectangular pulse, which is fed to the control inputs of analog keys 2 and 3. Opening, analog keys 2 and 3 close the differential capacitances C1 and C2 to the ground, leading to their full discharge. In this case, both comparators return to their original state. At the end of the control pulse, the analog keys 2 and 3 are closed, and the entire cycle of the measuring circuit is repeated. Since the signals from the outputs of both comparators are fed to the microprocessor, it is possible to programmatically determine both the moment of the beginning of the cycle, the moment of operation of the first comparator (connected to the smaller of the differential capacitors), and the moment of operation of the second comparator (time of the end of the cycle). Accordingly, it is not difficult to programmatically determine T ₁ , T _and , T _p and their ratio, which makes it possible to determine C _m , C _b , ΔC = C _b -C _m and ΔC / C _b in accordance with expressions (1-4). And by the sequence of operation of the comparators, it is easy to determine which of the differential capacitances is larger and which is smaller. Having the conversion function C = f (X) of a specific capacitive differential displacement transducer, it is not difficult to determine the measured displacement X.

The proposed method and the measuring circuit that implements it are especially convenient for constructing a gas flow meter based on a rotametric float primary transducer with a capacitive differential transducer for displacing the float, since it does not require individual adjustments, while at the same time it is easy to individually calibrate the entire counter for direct readout in units of volumetric gas flow.

Literature

1. Turichin A.M. Electrical measurements of non-electrical quantities. - M. - L.: Energy, 1966 .-- 690 p.

2. Levshina E.S., Novitsky P.V. Electrical measurements of physical quantities (measuring transducers). - L .: Energoatomizdat, 1983 .-- 320 p.

3. Farzane N.G., Ilyasov L.V., Azim-zade A.Yu. Technological measurements and instruments. Textbook for high schools. - M .: Higher school. 1989 .-- 456 p.

Claims (2)

1. The method of neutralizing the coupling capacitance of a differential motion transducer with an inaccessible movable electrode when using the time-domain conversion method, which consists in simultaneously charging the differential capacitance of a capacitive differential motion transducer from DC sources and comparing the voltage values of the capacitance of these capacitors with the set threshold values, characterized in that differential capacitances are charged bipolar, but equal in absolute value tnoj magnitude of currents being equal charging currents is maintained during the entire conversion cycle, whereby a capacitance connection of the capacitive differential transducer made their full mutual compensation. 2. A measuring circuit that implements the method according to claim 1, comprising a capacitive differential displacement transducer with an inaccessible movable electrode connected to the measuring circuit using a communication capacitance, two analog switches connected to the differential capacitances of this current source transducer, and two analog switches connected to the same points two comparators, the second inputs of which are supplied with reference voltages from a source of reference voltages, and the outputs of which are connected to a microprocessor, characterized in that they are used once non-polar sources of reference voltages, the polarity of which corresponds to the polarity of the current source used, and the outputs are connected to the second inputs of the comparators, the outputs of which are connected in parallel to the inputs of the microprocessor and the two-input I circuit, the output of which is connected to the input of a single-shot, and the output of the latter is connected to the control inputs of analog keys, moreover, throughout the entire conversion cycle, the equality of the absolute values of the charging currents is maintained by choosing the necessary ratio between the threshold voltage values yazheny and maximum possible values of the output voltages of the current sources, depending on the ratio of the minimum and maximum possible values of differential capacitances of capacitive differential transducer.

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