RU2260917C1 - Method for normalization of composite phase-manipulated signal - Google Patents

Abstract

FIELD: radio engineering, applicable in antiference radiolinks.

SUBSTANCE: the method is featured by the fact that the pseudorandom sequence with clock pulse f_p and for expansion of the spectrum is divided into two orthogonal sequences, one of which contains only even harmonics of the initial pseudorandom sequence, and the other - only the odd ones, then each of the obtained sequence is multiplied with a simple phase-manipulated signal, then the upper side band is separated from the spectrum of one obtained signal, and the lower side band - from the spectrum of the other signal, these unlike side bands are summed up, in each side band two narrow sections of the spectrum symmetrical relative to frequency f₀+1/2f_p, in the upper side band and relative to frequency f₀-1/2f_p in the lower side band, one of the separated sections of the spectrum in each side band of the separated spectrum sections is amplified to the known magnitude, and the other, symmetrical to it, is inverted, after which the separated and remained non-separated sections of the spectrum in both side bands are summed up, the separated narrow spectrum sections in each side band are altered according to the pseudorandom law.

EFFECT: enhanced anti-interference of the radiolink is attained due to the fact that in the method of normalization of the composite phase-manipulated signal consists in expansion of the spectrum of the simple phase-manipulated signal obtained by multiplication of the carrying sinusoidal oscillation with frequency f₀ and the binary information signal.

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Description

The present invention relates to the field of radio engineering and can be used in jamming radio links.

It is known that an increase in the noise immunity of radio links is achieved both due to the secrecy of the transmission, and due to the noise immunity of the reception. Secrecy determines the probability of detecting a signal, its identification (belonging), time for evaluating the parameters, type and effectiveness of organized interference.

Therefore, when choosing the type of signal in interference-protected radio lines, it is necessary to predict (provide) the enemy’s actions for reconnaissance and electronic suppression.

For electronic suppression, barriers and frequency-targeted interference are usually used. If the obstruction has a sufficiently large power, it is able to suppress both broadband and narrowband signals. However, the installation of obstruction interference is possible and not always advisable, since it suppresses its own radio lines operating in this frequency range. In addition, for the establishment of effective obstructive interference requires a large transmitter power. Under these conditions, to suppress a narrow-band signal or a signal with a slow pseudo-random tuning of the operating frequency (MHF), the adversary will most likely put an interference aiming in frequency than a barrage.

Widespread use in modern noise-protected radio links are found with frequency hopping signals. However, they have low energy secrecy. Higher energy secrecy is provided by complex phase-manipulated signals (SPS), the method of formation of which is described in the book (L.E. Varakin. Communication systems with noise-like signals. - M .: Radio and communications, 1985, p.16, Fig. 1.7) adopted for the prototype.

The essence of the prototype method is to expand the spectrum of a simple phase-shifted signal (FMNS) obtained by multiplying the carrier sinusoidal oscillation with a frequency f _o and a binary information signal. The expansion of the spectrum is carried out by multiplying a simple FMNS and a pseudo-random sequence (PSP) with a clock frequency f _t .

The prototype method is implemented in the device, a functional diagram of which is shown in figure 1, where it is indicated:

1 - carrier oscillation generator;

2 ¹ , 2 ² - multipliers;

3 - PSP generator.

However, the SPSK formed in this way has insufficient structural secrecy. Indeed, when such a signal is squared, discrete spectral lines are formed at frequencies:

f is 0; f _t ; 2f _about ; 2f _o ± f _t

where f _{about the} frequency of the carrier oscillation;

f _t - the clock frequency of the SRP.

The appearance of spectral lines at frequencies 2f _о , 2f _о ± f _t is the main unmasking sign of the binary SPSK, and facilitates its detection, identification and estimation of parameters.

The formation of these spectral lines is due to the antisymmetry of the phase spectrum of the SPSK relative to the frequencies f _о and f _о ± 1 / 2f _t (figure 2), which can be written as

φ (ff _о ) = - φ (f _о -f),

φ (ff _o ± 1 / 2f _t) = - φ (f _o ± 1 / 2f _t -f),

When squaring, the products of the frequency components are formed:

- upper and lower side bands, which give a spectral line at a frequency of 2f _about ;

- bands (f _o -f _o + 1 / 2f _t ) and (f _o + 1 / 2f _t -f _o + f _t ), which give a spectral line at a frequency of 2f _o + f _t ;

- bands (f _o -f _t -f _o -1 / 2f _t ) and (f _o -1 / 2f _t -f _o ), which give a spectral line at a frequency of 2f _o -f _t ;

- bands (f _o -f _t -f _o -1 / 2f _t ) and (f _o -f _o + 1 / 2f _t ), as well as (f _o -1 / 2f _t -f _o ) and (f _o + 1 / 2f _t -f _o + f _t ), which give a spectral line at a frequency f _t .

The most likely obstacle that an adversary can deliver with such a signal is a dangerous impulse obstruction with a band equal to the SPSK band (Addressable control and communication systems. Optimization issues. / Ed. By G.I. Tuzov. - M.: Radio and communications 1993, pp. 123-127). Such interference can bring the probability of error in the radio line to an unacceptably large value.

These shortcomings can be eliminated,

- firstly, the destruction of the symmetry of the amplitude spectrum, which prevents the formation of a spectral line at a frequency of 2f _o ;

- secondly, the compensation of spectral lines at frequencies 2f _o ± f _t , and the formation of the compensating voltage is ensured by radiation with increased power of narrow sections of the spectrum of a simple SPSK in each of the side bands in combination with a change in sign (inversion) of parts of the spectrum that are symmetrical with respect to the frequencies f _o + 1 / 2f _t and f _o -1 / 2f _t, respectively;

- thirdly, by changing, according to the pseudo-random law, the selected narrow sections of the spectrum emitted with increased power, which complicates the identification of the signal and reduces the likelihood of setting the most dangerous interference.

The problem that the present invention solves is to increase the noise immunity of the radio line and is achieved by the fact that in the method of generating a complex phase-shift key signal, which consists in expanding the spectrum of a simple phase-shift key signal obtained by multiplying the carrier sinusoidal wave with a frequency f _о and a binary information signal, used according to the invention spectrum expansion pseudo-random sequence (PSP) with a clock frequency f _{t is} divided into two orthogonal sequences singularities, one of which contains only even harmonics of the original SRP and the other only odd ones, then each of the obtained sequences is multiplied with a simple phase-manipulated signal, then the upper sideband is extracted from the spectrum of one received signal, the lower sideband is from the spectrum of the other signal the opposite side bands are folded, in each side strip two narrow sections of the spectrum are distinguished, symmetrical with respect to the frequency f _o + 1 / 2f _t in the upper side band and with respect to the frequency f _o -1 / 2f _t in the lower In each side band, one of the selected parts of the spectrum is amplified to a known value, and the other, symmetrical to it, is inverted, after which the selected and remaining unselected parts of the spectrum in both side bands are added up, and the narrow sections of the spectrum in each of the side bands change according to a pseudo-random law.

Graphic materials presented in the application:

Figure 1. - Functional diagram of the device that implements the prototype method.

Figure 2. - An example of the symmetry of the phase spectrum of the SPSK.

Figure 3. - Functional diagram of the device that implements the proposed method.

Figure 4. - The shape of the amplitude spectrum of the output signal.

Figure 5. - An example implementation of a switched inverter.

6. - Functional diagram of the control unit.

The proposed method is implemented in a device whose functional diagram is shown in figure 3, where it is indicated:

1 - pseudo-random sequence generator (GPSP);

2 - block delay for half the period of the SRP;

3 ¹ , ... 3 ^N - bandpass filters;

4 - block synchronization;

5 ¹ , 5 ² , 5 ³ - adders;

6 - block subtraction;

7 ¹ , ... 7 ^N - adjustable amplifiers;

8 - control unit;

9 - carrier oscillation generator;

10 ¹ , 10 ² , 10 ³ - multipliers;

11 ¹ , ..., 11 ^N - switched inverters;

12 - filter of the upper (lower) side strip;

13 - filter of the lower (upper) side strip.

The device contains a generator PSP 1, the first output of which through the synchronization unit 4 is connected to the input of the control unit 8, and the second output is connected to the first inputs of the adder 5 ¹ , the subtraction unit 6 and the input of the delay unit 2, the output of which is connected to the second inputs of the adder 5 ¹ and subtracting unit 6. The output of the adder May ¹ through series-connected multiplier ¹⁰ February filter and the upper (lower) side panel 12 is connected to a first input of an adder March ^5, the second input of which is connected in series through the filter bottom (top) of the side strips 13 and ¹⁰ March multiplier connected to the output of the subtracting unit 6. The apparatus further comprises series-connected carrier wave generator 9 and the multiplier ¹⁰ January, a second input which is the input binary information signal, and an output connected to second inputs of the multipliers ^February 10th and ^March 10th. The output of the adder 5 is connected to the input of each of the N band-pass filters 3 ¹ , ..., 3 ^N , the outputs of which through the corresponding series-connected adjustable amplifiers 7 ¹ , ..., 7 ^N and switched inverters and 11 ¹ , ..., 11 ^{N are} connected to the corresponding N inputs of the adder 5 ² , the output of which is the output of the device. The first N outputs of the control unit 8 are connected to the second (control) inputs of the respective adjustable amplifiers 7 ¹ , ..., 7 ^N , and the second N outputs of the control unit 8 are connected to the second (control) inputs of the corresponding switched inverters 11 ¹ , ..., 11 ^N.

The device that implements the proposed method is as follows.

A periodic pseudo-random sequence with a clock frequency f _t generated by the generator 1 is added up in the adder 5 ¹ and subtracted in block 6 with its copy delayed by half a period from the output of block 2.

As a result, at the output of adder 5 ¹ a sequence is formed whose spectrum consists only of even (2,4,6 ...) harmonics of the original SRP, and at the output of subtraction block 6 another sequence is formed whose spectrum consists only of odd ones (1,3 , 5 ...) harmonics of the original SRP.

In block 10 ^{1, a} carrier sinusoidal oscillation having a frequency f _o generated by the generator 9 is multiplied and a binary information signal is obtained, resulting in a simple phase-shifted signal whose spectrum is expanded by multiplying this signal in blocks 10 ² and 10 ³ with the corresponding orthogonal sequences arriving from the outputs of the adder 5 ¹ and the subtraction block 6.

The sideband filter 12 selects the upper (or lower) band of the signal spectrum from the output of block 10 ² , and the sideband filter 13 selects the lower (or upper) spectrum band of the signal from the output of multiplier 10 ³ , respectively.

As a result of the addition of these opposite sidebands, the output of the adder 5 ³ produces a two-band signal with an asymmetric amplitude spectrum (the presence of harmonics in the upper sideband corresponds to the absence of the corresponding harmonic in the lower sideband).

Next, from the spectrum of the received signal in each sideband, for example, two narrow sections of the spectrum are symmetric with respect to the frequency f _o + 1 / 2f _t in the upper sideband and relative to the frequency f using a comb of band-pass filters 3 ¹ , ... 3 ^N _about -1 / 2f _t in the lower side strip. In each of the side bands, one of the selected sections of the spectrum is amplified in the corresponding adjustable amplifier 7 ¹ , ..., 7 ^N to a known calculated value, and the other selected section, symmetrical to it, is inverted (the sign of the input voltage is changed) in the corresponding switched inverter 11 ¹ , ..., 11 ^N by commands from the corresponding output of the control unit 8. The gain of each of the amplifiers 7 ¹ , ..., 7 ^N depends on the level of the control voltage coming from the corresponding output of block 8.

In adder 5 ^2, all selected and remaining unselected parts of the spectrum in both side bands are added up.

Thus, at the output of the device there is a complex combined phase-shift signal, consisting of broadband and four narrowband components, the amplitude spectrum of which is presented in figure 4, where

Δf is the width of the spectrum of a simple FMNS, which can be taken equal to the frequency band of the broadband component;

ΔF - frequency band of one narrowband component;

K is the gain of the selected narrow section of the spectrum;

.N _c - the spectral density of a simple FMNS power p _with equal

.

This signal can be considered as the sum of two signals: a broadband signal with high energy structural secrecy and a signal with slow frequency hopping, which has reduced energy secrecy.

The possibility of forming a discrete spectral line at a frequency of 2 f _{about is} excluded due to the destruction of the symmetry of the amplitude spectrum with respect to the frequency f _o .

The exclusion of the possibility of the formation of discrete spectral lines at frequencies 2f _o ± f _{t is} achieved by compensation, which is carried out by adding the components with different signs due to the broadband and narrowband components, as can be seen from figure 4.

To compensate for discrete lines at frequencies 2f _o ± f _t provided that the spectrum of the simple SPSK is uniform, the equality

where is the gain

The above equation reflects the equality of the amplitudes of discrete lines at frequencies 2f _o ± f _t generated in each sideband when multiplying a narrow-band component with a symmetric part of the spectrum and the rest of the spectrum of a simple FMNS.

The frequency change of narrow-band components and symmetric parts of the spectrum in the side bands is carried out according to the pseudo-random law. In this way, the frequency hopping signal is simulated. Narrowband components obviously have lower energy stealth, so they are detected faster than the broadband component. It is natural to assume that narrowband components will be suppressed in the first place, and narrowband frequency-targeted interference will most likely be used. These narrowband interference when receiving a signal can be rejected. In this case, the broadband component can be taken in the same way as a conventional binary SPSK.

It should be emphasized that the proposed method allows you to generate a complex signal that can be received by the receivers of the old park, designed for ordinary SPSK.

Thus, the use of the proposed complex combined phase-shift signal

- firstly, it complicates the detection, identification and disclosure of its structure by typical SPSK reconnaissance receivers;

- secondly, it reduces the likelihood of the enemy putting the most dangerous interference;

- thirdly, it can be carried out by receivers of the old park without modification.

It follows that the use of such a signal makes it possible to increase the noise immunity of radio lines compared to the use of typical (binary) SPSKs.

The elements that make up the device, the functional diagram of which is presented in figure 3, are mainly typical.

An example implementation of a switched inverter 11 is presented in figure 3, where indicated:

11.1 - non-inverting repeater;

11.2 - inverting repeater;

11.3 ¹ , 11.3 ² - keys;

11.4 - scheme \ NOT \;

11.5 - adder.

Each of the switched inverters 11 ¹ ... 11 ^N (Fig. 3) contains a series-connected non-inverting repeater 11.1 and a key 11.3 ¹ , the output of which is connected to the first input of the adder 11.5, the output of which is the output of the inverter 11, as well as a series-connected inverting repeater 11.2 11.3 and key ^2, the output of which is connected to the second input of adder 11.5, which repeaters inputs 11.1 and 11.2 are combined and the first input of the inverter 11. The second input of the switchable inverter 11 is a control (3) and connected to the second input of Cl 11.3 ca ² and through the circuit \ NO \ 11.4 - a second input key ¹ 11.3.

The keys 11.3 ¹ , 11.3 ^{2 are} unlocked when applied to the second (control) inputs of a logical unit and pass the input signal with or without inversion depending on the control voltage.

The control unit 8 at its outputs generates pulses that control the gain of the adjustable amplifiers 7 ¹ , ..., 7 ^N and the switched inverters 11 ¹ , ..., 11 ^N. In the simplest case, it can be performed on the basis of a trigger pulse counter and a group of decoders configured for certain states of this counter.

Functional diagram of the control unit 8 is presented in (Fig 6).

When the state of the counter changes, the numbers of the adjustable amplifiers 7 ¹ , ... 7 ^N (Fig. 3) with a higher gain (K) change, and therefore, the frequency jump of the narrowband component occurs. In this case, the corresponding inverter 11 ¹ , ..., 11 ^N (Fig. 3) of the symmetric portion of the signal spectrum is turned on.

. Тогда число триггеров в счетчике должно быть равно 6 (2⁶=64).Let's pretend that

. Then the number of triggers in the counter should be equal to 6 (2 ⁶ = 64).

At the outputs of the counter triggers six-digit numbers are formed, starting from 000000 and ending with 111111. Using six-input circuits AND and connecting their outputs with the corresponding direct and inverse outputs of the counter triggers using a switching matrix, 64 decoders can be arranged. In this case, each decoder will be tuned to the corresponding number.

The first set of 64 decoders will change the gain of the respective adjustable amplifiers 7 ¹ , ..., 7 ^N (figure 3). The second set of 64 decoders will control the corresponding inverters 11 ¹ , ..., 11 ^N (figure 3).

The synchronization unit 4 can be made in the form of a decoder, the output of which appears when a certain combination of states of the triggers of the shift register appears, on the basis of which the generator PSP 1 is built.

Claims (1)

A method of forming a complex phase manipulated signal comprising the spread spectrum prime phase manipulated signal obtained by multiplying the carrier sine wave of frequency f _o and the binary information signal, characterized in that used for spread-spectrum pseudorandom sequence (PRS) with a frequency f _r is separated into two orthogonal sequences, one of which contains only even harmonics of the original SRP, and the other contains only odd harmonics, then each of them obtained by been consistent multiplied with a simple phase-shift keyed signal, hereinafter from the spectrum of the received signal recovered upper sideband of the spectrum of another signal - the lower sideband, these differently constituted sidebands folded, in each sideband separated by two narrow spectrum which is symmetric with respect to the frequency f _o + 1 / 2f _into an upper sideband and relatively frequency f _of -1 / 2f _t in the lower sideband, each of the side bands one of the selected portions of the spectrum to enhance the known value and the other, the symmetry ary him, inverted, and then allocated the remaining unselected portions of the spectrum in both sidebands are summed, and the narrow portions of the spectrum allocated to each of the side panels change according to a pseudorandom.

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