RU2438250C1 - Method of transmitting and receiving signals - Google Patents

Abstract

FIELD: information technology.

SUBSTANCE: method involves simultaneous emission of several amplitude- and phase-shift keyed useful signals and a masking signal, summation thereof and suppression of the masking signal during reception, wherein the masking signal emission used is not only similar to the useful signal on polarisation, spatial, time and frequency parameters, but also overlaps the frequency variation range of each useful signal emission; two reference signals with equal initial phases are generated, wherein all parameters of one of them coincide with parameters of the emitted useful signal, and the second reference signal is distinguished only by carrier frequency; a mutual correlation function is generated between the receiving additive family of the masking signal and the corresponding reference signal, and the second mutual correlation function is subtracted from the first.

EFFECT: higher security of transmitting information.

3 cl, 12 dwg

Description

The invention relates to the field of radio engineering and can be used in radio communications and other systems to increase the secrecy of information transfer.

The reliability of the transmission of confidential information substantially depends on the energy and structural secrecy of the signals used. Currently widely used methods of transmitting digital information using noise-like binary signals, manipulated in amplitude, phase or frequency. Such methods of transmitting digital information are widely used in the creation of communication systems, mobile telephony, access loops for radiotelephone systems, the Internet, wireless local area networks, long-distance space communications, multi-access systems and other purposes.

A known method of digital communication with an expanded spectrum of the signal by modulation using complementary Golei sequences, as well as a transmitter and a receiver for its implementation [RF Patent No. 2280957, IPC Н04К 27/18, publ. July 27, 2006].

However, this method and the devices used for its implementation do not provide reliable secrecy of information transfer, because the useful signal (PS) used at its zero level of the side lobes of the correlation function does not involve the use of a misinforming masking signal (MS).

A known method of transmitting a masked signal, which uses two-channel transmission of encoded chaotic oscillations [E.V. Kalyanov. Information transfer using chaotic oscillation coding. - Radio engineering and electronics, 2002, v. 47, fig. 1, p. 469]. The first channel transmits the sum of the PS and the MS, but the MS is converted in amplitude and phase, and on the second, the non-transformed MS. At the input of its own receiver, the MS received on the second channel is subjected to conversion similar to that in the transmitter and subtracted from the total signal received on the first channel. As a result of this, after subtraction, only the PS remains.

The method does not provide reliable secrecy of information transfer, because does not exclude the possibility of compensation of MS and allocation of PS. Due to the separate transmission of the converted and not converted masking signals, it becomes possible to separately receive not only the sum of the PS and the MS converted in amplitude and phase, but also the non-converted MS. This facilitates the determination of the laws of MS modulation, obtaining from an unconverted MS transformed by the amplitude and phase of the MS, and opening the PS structure.

There is a method of increasing the secrecy of the transmission of a group of narrow-band signals, in which a masking (misinforming) signal is emitted regardless of the PS group [RF Patent No. 2232475, IPC Н04К 1/02, publ. 07/10/2004]. In accordance with this method, MSs are converted into spread spectrum MSs; in the receiver, simultaneously with compression of the spread spectrum MSs, the spectra of each PS are expanded. The initial MS compressed in the spectrum is rejected, and then each of the spread spectrum PSs is compressed in the spectrum with restoration of the input PSs.

The method does not provide reliable secrecy of the transmission of information and the reliability of the allocation of PS, because does not exclude the possibility of highlighting narrowband PS against the background of broadband MS in the process of decryption [Widrow B., Stirnz S. Adaptive signal processing / Transl. from English - M.: Radio and Communications, 1989. pp. 320-322, Fig. 12.36 - 12.38]. In addition, it is known that frequency rejection does not suppress all spectral components of a spectrum-squeezed MS, which leads to the appearance of uncompensated MS residues, a decrease in the MS compensation efficiency and the reliability of PS isolation [Adaptive interference compensation in communication channels / Yu.I. Losev, A G.G. Berdnikov, E.Sh. Goikhman et al. / Ed. Yu.I. Loseva. - M .: Radio and communications, 1988, p. 13, 72].

Closest to the claimed method in its technical essence is a way to increase the secrecy of the transmission of a group of binary PS, manipulated in amplitude, phase or frequency [RF Patent No. 2282941, IPC Н04К 1/02, publ. 08/27/2006]. In accordance with this method, when transmitting a PS group, not only each PS, manipulated in phase or frequency, but also a broadband MS, the spectrum of which covers the frequency range of the entire PS group, is independently emitted. The value of the carrier frequency MS is selected as a multiple of the reciprocal duration of one PS element, and the period is a multiple of the duration of one PS element. Useful digital information is transmitted using manipulated amplitude, frequency or phase PS. The carrier frequencies of useful signals in the group are multiples of the reciprocal of the duration of one PS element. Upon receipt of the additive mixture of the PS group and broadband MS, the MS is compensated by subtracting from the unrestricted additive mixture of the PS and MS the additive mixture of the PS and MS delayed for the period of the masking signal. The distortions of the useful signals resulting from compensation are eliminated in the PS recovery blocks included at the outputs of the receivers.

The method does not provide reliable secrecy of the transmission of information, as well as sufficient compensation efficiency of the MS and the reliability of the allocation of PS. This is due to the fact that with a frequency difference in the spectral widths of PS and MS, there are methods for isolating narrow-band PS against the background of broadband MS [Widrow B., Stirnz S. Adaptive signal processing / Transl. from English - M.: Radio and Communications, 1989. pp. 320-322, Fig. 12.36-12.38]. In addition, the periodic repetition of the same structure of the MS allows it to be compensated (i.e., to open the PS) not only when received in its own radio channel, but also in the means of radio reconnaissance using inter-period compensation.

The patent did not analyze the effectiveness of inter-period compensation for MS depending on the multiplicity of inter-period subtraction. However, analysis of the compensation efficiency of interfering signals depending on the subtraction factor shows that the worst noise suppression is realized with a single subtraction of the MS, which is used in the prototype method [Radio systems: Textbook. for universities / Yu.P. Grishin, V.P. Ipatov, Yu.M. Kazarinov et al. / Ed. Yu.M. Kazarinova. - M .: Higher. school, 1990, pp. 265-268, Fig. 12.3]. Therefore, a single compensation does not provide reliable suppression of all spectral components of the MS, which reduces the effectiveness of the compensation of the MS, and therefore reduces the reliability of the allocation of PS.

The claimed invention is aimed at increasing the secrecy and reliability of the reception of useful radio signals.

The technical result to which the invention is directed is to form a radio signal structure for which the masking and useful signals do not differ in statistical, structural, polarizing, spatial, frequency and temporal characteristics.

This technical result in the claimed method is achieved by the fact that in the known method of transmitting and receiving binary signals (which consists in simultaneously emitting the amplitude and phase of the useful and masking signals, forming their additive mixture at the receiver input, and then suppressing the masking signal when receiving) use the radiation of a masking signal, which not only does not differ from the useful signal in polarization, spatial, temporal and frequency parameters, but also overlaps the range of variation of the radiation frequency of each of the desired signal. To achieve the desired technical result, two reference signals with equal amplitudes and initial phases are additionally formed. In this case, the parameters of the first of them coincide with the parameters of the emitted useful signal. The second of them has all the parameters except the carrier frequency, which is mirror relative to the carrier frequency of the masking signal, also coincide with the parameters of the emitted useful signal. Using these reference signals, the cross-correlation functions between the received additive mixture of the useful and masking signals and the corresponding reference signals are determined, and then the second is subtracted from the first cross-correlation function. The result of the subtraction, which is the autocorrelation function of the useful signal, will be the output of the receiver. The formation of two reference signals, two cross-correlation functions and their subsequent subtraction implements a new method of interspectral post-correlation compensation of the masking signal.

To create two reference signals with equal parameters, in addition to the carrier frequency, we consider the frequency conversion. As follows from well-known literature sources, in order to shift the spectrum of the useful signal both to the region of the total frequency of oscillations involved in this transformation and to the region of the difference frequency, it is necessary to carry out the frequency conversion. In the region of the total frequency of oscillations, the phases of the oscillations involved in this transformation are summed up, and in the region of the difference frequency, they are subtracted [Gonorovsky IS, Demin MP Radio engineering circuits and signals: Textbook. benefits for universities. - 5th ed., Revised. and add. - M .: Radio and communications, 1994, pp. 240-243, (8.51) - (8.54 '); Radio engineering circuits and signals: Textbook. benefits for universities. / Ed. K.A. Samoilo. - M.: Radio and Communications, 1982, p. 245-250, (9.25) - (9.31)].

The prior art currently allows you to convert the phase of the signal to the opposite or when using an additional signal generator with a frequency twice the frequency of the signal, which phase is converted [Radio-electronic systems: the basics of construction and theory. Directory. / Shirman Y.D., Losev Yu.I., Minervin N.N. and etc.; Ed. J.D. Shirman. - M .: ZAO Makvis, 1998. p. 740, fig. 25.21; Radio-electronic systems: the basics of construction and theory: a Handbook. Ed. 2nd, rev. and add. / Ed. J.D. Shirman. - M .: Radio engineering, 2007, p. 450, fig. 25.34], or when using the well-known quadrature signal processing [Sergienko AB Digital Signal Processing: A Textbook for High Schools. 2nd ed. - St. Petersburg: Peter, 2007, pp. 168-170, Fig. 3.14] using an additional harmonic signal generator [Radio engineering circuits and signals: Textbook. benefits for universities. / Ed. K.A. Samoilo. - M.: Radio and Communications, 1982, pp. 249-250]. When using an additional harmonic signal generator, the phase difference of the quadrature components will be opposite to the phase difference obtained as a result of the classical frequency conversion. The restoration of the radio signal from these quadrature components will make it possible to obtain a heterodyne signal with the opposite initial phase. Thus, at first two heterodyne signals of the same frequency are formed, but with opposite phases.

In the subsequent frequency conversion from the original useful signal using these heterodyne signals, two reference signals are formed, respectively, at the total and difference frequencies, in which the initial phases are equal. Since two reference signals at different carrier frequencies are obtained by converting the frequency of the original useful signal using two heterodyne signals of equal amplitude, the complex envelopes of the reference signals will be equal.

An essential aspect of increasing the efficiency of using various systems of radio electronics is the use of the same radio frequency resource, i.e. common frequency band for various radio signals. To increase not only the reliability of the transmission of the primary signal (signal carrier of the information message), but also reduce the inter-channel interference of signals in radio communications, codes are used with the expansion of the signal spectrum. To isolate each of the signals simultaneously present in the common frequency band, it is necessary to use radio signals orthogonal either in form or in time of their existence [Gepko I.A., Oleinik V.F., Chaika Yu.D., Bondarenko A.V. . Modern wireless networks: state and development prospects. - K .: Ekmo, 2009, p. 73-74].

Widespread, for example, spread-spectrum signals that are phase-manipulated according to orthogonal codes, for example, based on the Walsh-Hadamard matrix [A.M. Shloma, M.G. Bakulin, V.B. Kreidelin, A.P. Shumov. New algorithms for generating and processing signals in mobile communication systems. - M .: Hot line - Telecom, 2008, p. 43; Gepko I.A., Oleinik V.F., Chaika Yu.D., Bondarenko A.V. Modern wireless networks: status and development prospects. - K .: Ekmo, 2009, pp. 165-167].

Along with the given codes, orthogonal codes of variable length, the so-called OVSF codes [A.M. Shloma, M.G. Bakulin, V.B. Kreidelin, A.P. Shumov. New algorithms for generating and processing signals in mobile communication systems. - M .: Hot line - Telecom, 2008, p. 43; Gepko I.A., Oleinik V.F., Chaika Yu.D., Bondarenko A.V. Modern wireless networks: state and development prospects. - K .: Ekmo, 2009, pp. 170-172].

The use of orthogonal signals in the same frequency range leads to the fact that the formation of an additive mixture of useful and masking signals at the input of the receiver is carried out by simultaneously emitting the useful and masking signals, and the phase manipulation of the useful signals is determined by orthogonal both short and long codes. In order to distinguish the autocorrelation functions of the corresponding useful signals in the received additive mixture, cross-spectral post-correlation compensation of the masking signal for each useful signal is necessary.

Although Walsh codes are complete systems of functions, they are “orthogonal only at a point”, i.e. in the absence of mutual temporary mismatch of signals. As a result of this, the mutual correlation function (VCF) of signals using orthogonal Walsh codes has sufficiently large side lobes (correlation noises), which lead to a decrease in the reliability of the transmission of information of other signals found in these side lobes of the VKF [I. Gepko, Oleinik V.F., Chaika Yu.D., Bondarenko A.V. Modern wireless networks: status and development prospects. - K .: Ekmo, 2009, p.167].

Thus, the goal of using orthogonal signals, which would ensure the receipt of side lobes equal to zero, was not achieved [V. Lityuk, L. V. Lityuk Methods of digital multiprocessor processing of ensembles of radio signals. - M.: SOLON-PRESS, 2007, p. 495; Gepko I.A., Oleinik V.F., Chaika Yu.D., Bondarenko A.V. Modern wireless networks: status and development prospects. - K .: Ekmo, 2009, p. 165].

The solution to this problem was the use of complementary signals. A characteristic feature of complementary signals is the summation of the correlation functions (CF) of complementary signals (most often pairs) of a group. The found new properties of the group of complementary signals allow the side lobes of the CF group to be equal to zero (CFG). In this case, the main lobes of CF complementary signals of the group will be summed. In addition, equality to zero of the total VKF of complementary signals in the group is achieved. The latter, as shown by well-known studies [Lityuk V.I., Lityuk L.V. Methods of digital multiprocessor processing of ensembles of radio signals. - M.: SOLON-PRESS, 2007, p. 403; M.V. Litvin. The property of selective quasi-splitting of the Doppler frequency of complementary signals // Questions of radio electronics, ser. RLT, 2009, issue 1, pp. 139-146; Gepko I.A., Oleinik V.F., Chaika Yu.D., Bondarenko A.V. Modern wireless networks: status and development prospects. - K .: Ekmo, 2009, pp. 175-178], can be achieved only if the condition of "orthogonality only at a point" of each of the signals of the group, ie in the absence of mutual temporary mismatch of signals.

To achieve the required technical result when using complementary useful signals, the results of the corresponding inter-spectral post-correlation compensations of the masking signal, included in one group, must be summed. This will allow, after summing, to obtain CFG of complementary useful signals, which will be the output for the corresponding group.

Thus, the essence of the method lies in the fact that the suppression of the masking signal is carried out by the method of cross-spectral post-correlation compensation, for the implementation of which the correlators use the new principle of generating reference useful signals with the same initial phase at different carrier frequencies.

The list of graphic materials illustrating the invention is shown in figures 1-12.

Figure 1 presents the structural diagram of a communication system that implements the proposed method using a useful signal, manipulated in amplitude and phase.

Figure 2 presents the structural diagram of a communication system that implements the proposed method using useful signals, phase manipulation of which is determined by the corresponding orthogonal or short or long codes.

Figure 3 presents a structural diagram of a communication system that implements the proposed method when using useful signals, in which the phase manipulation of the corresponding signal from the group of useful signals is determined by the corresponding complementary code from the group of codes.

Figure 4 presents a temporary plot of PS.

Figure 5 presents the amplitude-frequency spectra of the PS.

Figure 6 presents the modules of the autocorrelation functions of the PS.

Figure 7 presents the time plots of the additive mixture of PS and MS and their amplitude-frequency spectra at zero frequency offset.

On Fig presents time plots of the additive mixture of PS and MS and their amplitude-frequency spectra when the carrier frequency of the PS is shifted by half the width of the MS spectrum.

Figure 9 shows the time plots of the additive mixture of PS and MS and their amplitude-frequency spectra when the carrier frequency of the PS is shifted by the width of the MS spectrum.

Figure 10 presents the time plots of the additive mixture of PS and MS and their amplitude-frequency spectra when the carrier frequency of the PS is shifted by one and a half spectral widths of the MS.

Figure 11 shows the modules of the autocorrelation functions of the PS after the interspectral post-correlation compensation of the MS when the carrier frequency of the PS is shifted by half the width, width and one and a half widths of the MS spectrum, respectively.

On Fig presents the modules of the autocorrelation functions of complementary PS without MS and after interspectral post-correlation compensation of the MS when the carrier frequency of the PS is shifted by half the width, width and one and a half widths of the MS spectrum, respectively.

Structural diagrams of the communication system for the relevant claims that implement the proposed method are shown in figures 1, 2, 3, respectively, and contain:

1ps, 1ms - transmitters of useful and interfering signal with antennas, respectively;

2ps, 2ms - communication lines (channels) of corresponding signals;

3 - a receiving antenna of its radio link;

4 - receiver PS;

5 - correlator receiver PS;

6 - driver of reference signals with equal amplitudes and phases;

7 - the second correlator of the receiver PS;

8 is a subtraction scheme;

9 - adder;

IPC - inter-spectral post-correlation compensation of the masking signal, including the formation of two reference signals, two cross-correlation functions and their subsequent subtraction.

In these structural diagrams, a double arrow indicates several signals.

To explain the operation of the system, reconnaissance by the enemy, we consider a block diagram explaining the first paragraph of the formula of the proposed method. This block diagram shown in Fig. 1 consists of transmitters of useful 1ps and interfering 1ms signals, communication lines (channels) 2ps, 2ms, receiving antenna 3, receiver PS 4, correlator of receiver PS 5, a reference signal shaper with the same phases 6, the second correlator of the receiver PS 7, the subtraction scheme 9.

Using transmitters of useful and interfering signals 1ps and 1ms, PS and MS are simultaneously emitted. The carrier frequency of the emitted useful signal ω _pc0 is shifted relative to the frequency of the interfering ω _ms0 by _ΔΩ , i.e. PS is emitted at a frequency (ω _ms0 - _ΔΩ ). In the process of passing the air 2ps and 2ms, an additive mixture of emitted PS and MS is formed, which is received by the receiving antenna 3, and then goes to the receiver PS 4, and then to the correlators of the receiver PS 5 and 7. At the first inputs of the first and second correlators the output signal the receiver, which is an additive mixture of emitted PS and MS, and reference signals with equal amplitudes and phases, i.e., amplitudes and phases, are fed to the second inputs of the first and second correlators with equal complex amplitudes, at the fundamental (ω _ms0 - _ΔΩ ) and "mirror" (ω _ms0 - (- ΔΩ)) frequencies, respectively.

The definition of both ACF and VKF F (τ) signals is greatly simplified if we use analytical signals [Radio engineering circuits and signals: Textbook. benefits for universities. / Ed. K.A. Samoilo. - M .: Radio and communications, 1982, p.112-116; Gonorovsky I.S., Demin M.P. Radio engineering circuits and signals: Textbook. for universities. - 5th ed., Revised. and add. - M .: Radio and communications, 1994. pp. 105-107], which are associated with the correlation function of the original real signal Φ _s (τ) by the following relation

Φ _s (τ) = 1/2 Re [Φ (τ)].

To create reference signals with equal amplitudes and phases, i.e. with equal complex envelopes, at first two heterodyne signals of the same frequency ΔΩ, but with opposite phases, are formed in the driver of the reference signals 6

- амплитуда гетеродинных сигналов.Where

- the amplitude of the heterodyne signals.

In a subsequent frequency conversion from the original wanted signal

snc (t) = S _ps (t) exp (jω _ms0 t)

Using these heterodyne signals, two reference signals are formed, respectively, at the total and difference frequencies, in which the initial phases are equal. As a result of these frequency conversions, the reference signals are as follows

s _on1 (t) = S _nc (t) exp [j (ω _ms0 -ΔΩ) t]

and

s _on2 (t) = S _nc (t) exp [j (ω _ms0 + _ΔΩ ) t].

комплексная огибающая опорных сигналов.Here

complex envelope of reference signals.

Since two reference signals at different carrier frequencies are obtained by converting the frequency of the original useful signal snc (t) = S _ps (t) exp (jω _ms0 t) using two heterodyne signals of equal amplitude, the complex envelopes of the reference signals son1 (t) and son2 (t) will be equal.

Adopted additive mixture of useful snc (t) (PS)

ω

ω

and masking sms (t) (MC)

, начальные фазы

и законы фазовой манипуляции θnc(t), но разные несущие частоты (ω_мс0-ΔΩ) и (ω_мс0+ΔΩ)of signals [snc (t) + sms (t)] after receiver 4 is supplied to the first inputs of the first 4 and second correlators 7, and the second inputs receive the first son1 (t) and second son2 (t) reference signals, respectively. Reference signals have equal amplitudes

initial phases

and the laws of phase manipulation θnc (t), but different carrier frequencies (ω _ms0 -ΔΩ) and (ω _ms0 + _ΔΩ )

The signals at the outputs of the correlators for analytical signals are determined by the expressions of the following type [Gonorovsky I. S., Demin MP Radio engineering circuits and signals: Textbook. benefits for universities. - 5th ed., Revised. and add. - M.: Radio and Communications, 1994, p. 65-71].

Here, the superscript “*” means a complex conjugate.

As proposed in the invention, to implement the new IPC method, it is necessary to subtract the signal of the second (6) from the signal at the output of the first correlator (5)

after which the ACF of the useful signal should remain.

The signal at the output of the correlator in the case of processing one PS in the absence of MS can be represented as follows [Gonorovsky IS, Demin MP Radio engineering circuits and signals: Textbook. benefits for universities. - 5th ed., Revised. and add. - M.: Radio and Communications, 1994, p. 65-71].

The maximum signal at the output of correlator (8) is achieved at τ = 0, i.e. in the absence of a temporary mismatch between the useful snc (t) and the reference son1 (t) signals [Radio-electronic systems: the basics of construction and theory: Reference. Ed. 2nd, rev. and add. / Ed. J.D. Shirman. - M .: Radio engineering, 2007, p. 234., Honorovsky I.S., Demin M.P. Radio engineering circuits and signals: Textbook. benefits for universities. - 5th ed., Revised. and add. - M.: Radio and Communications, 1994, p. 65-71]. It follows that the signal difference at the outputs of the correlators (7) must be equal to the integral (8), which means that all other terms in (7) must be equal to zero.

To explain the compensation of the masking signal, we first consider the second term of the integral (5)

Then, taking into account (2) and (3), expression (9) can be written

We express the complex conjugate envelope of the useful snc (t) in terms of its spectral density

and substitute in the formula (10)

We replace the variable in the last integral and get

Now we turn to the second term of the integral (6)

To obtain the VKF F ₂₂ (τ) of the masking signal and the second reference signal, we perform transformations similar to the calculation of the VKF F ₁₂ (τ), but taking into account the mirror frequency shift relative to the carrier frequency of the MS. You can finally record

As was noted, the difference [Φ _s12 (τ) _{-Φ s22} (τ)], which determines the quality of the inter-spectral post-correlation compensation of the MS, should be equal to zero. Meanwhile, using expressions (12) and (13), it is rather difficult to conclude that expression (7) is equal to zero. Therefore, you need to convert these VKF.

In the work [Radio-technical circuits and signals: Textbook. benefits for universities. / Ed. K.A. Samoilo. - M .: Radio and communications, 1982, pp. 114-116; Varakin L.E. Communication systems with noise-like signals. - M .: Radio and communications, 1985, p. 39, (3.1) - (3.3)] it is shown that the complex envelope of the phase-shift key signal can be represented as

, θ_n - модуль и аргумент комплексной огибающей дискреты фазоманипулированного сигнала; N - количество дискрет фазоманипулированного сигнала; τ_д - длительность дискреты. Для случая бинарных фаз θ_n=±π можно также ввести последовательность C_n=exp[j θ_n], элементы которой принимают ±1 в зависимости от значения фазы θ_n. Тогда ВКФ маскирующего и первого опорного сигналов можно записатьHere

, θ _n is the modulus and argument of the complex envelope of the discrete of the phase-manipulated signal; N is the number of discrete phase-shifted signal discrete; τ _d - the duration of the discrete. For the case of binary phases θ _n = ± π, one can also introduce the sequence C _n = exp [j θ _n ], whose elements take ± 1 depending on the value of the phase θ _n . Then, the VKF of the masking and first reference signals can be written

Since the interspectral post-correlation MS compaction is carried out using simultaneously present discrete reference and masking signals, it is possible to limit, without loss of generality, the time interval of one discrete. Then

Performing transformations similar to the derivation of relation (12), we obtain

- амплитудно-частотный спектр МС.Here

- the amplitude-frequency spectrum of the MS.

We represent the frequency spectrum of the MS discretes in the form of the sum of the main G _{d msn0} and the side lobes of the spectrum G _{d msnK} .

because they are orthogonal. Here K is a discrete variable that determines the number of the side lobe of the spectrum

будет четной функцией, а аргумент θ _мсnK - нечетной:

, θ_мсnK=-θ_мсn-K. При этом следует обратить внимание, θ_мсnK - кратно (-π), θ_мсn-K - кратно (+π) [Радиотехнические цепи и сигналы: Учеб. пособ. для вузов. / Под ред. К.А.Самойло. - М.: Радио и связь, 1982, стр.56-57, рис.2.9, рис.2.10; Гоноровский И.С., Демин М.П. Радиотехнические цепи и сигналы: Учеб. пособ. для вузов. - 5-е изд., перераб. и доп. - М.: Радио и связь, 1994. стр.21-35, рис.2.9, рис.2.13].Similar to harmonic signal analysis, the module of the lobes of a rectangular discrete

will be an even function, and the argument θ _{msnK will} be odd:

, θ _msnK = _{-θ msn-K} . It should be noted that θ _msnK - multiple (-π), θ _msn-K - multiple (+ π) [Radio circuits and signals: Textbook. benefits for universities. / Ed. K.A. Samoilo. - M .: Radio and communications, 1982, p. 56-57, fig. 2.9, fig. 2.10; Gonorovsky I.S., Demin M.P. Radio engineering circuits and signals: Textbook. benefits for universities. - 5th ed., Revised. and add. - M.: Radio and Communications, 1994. pp. 21-35, Fig. 2.9, Fig. 2.13].

All this analysis allows for any Kth lobe of the spectrum, the MS discretes to represent the VKF (15) and (16) in the following form

A comparison of expressions (15) and (16) shows that the integrands will be equal to each other, which we denote

Here, φ _nmK is the phase R _{nmK (τ)} .

This equality follows from the equality of the mirror side lobes of the amplitude spectrum of the discrete complex envelope of the MS, which is multiplied by the same spectrum of the discrete of the complex envelope of the PS. The same spectrum discrete complex envelope PS was achieved due to the equality of the complex envelopes of the reference signals at the mirror carrier frequencies. At the same time, as already noted, the phase-frequency spectrum of MS discrete will be odd.

Let us turn to the correlation functions of real signals, which will be equal

Ф _s12nm-K (τ) = 1/2 Re [Ф _12nm-K (τ)], Ф _22nmK (τ) = 1/2 Re [Ф _22nm (τ)].

The signs in the expressions “±” are due to the consideration of the even or odd side lobe of the MS, i.e. parity or oddness K.

As a result of this, when implementing the new IPC method in the process of subtracting the signal of the second correlator from the signal at the output of the first (7), we obtain the following difference for one discrete of the Kth side lobe of the MS

which is zero.

Similar differences, equal to zero, will be for other side lobes and discrete MS.

Last integral in expression (6)

This follows from the well-known time-frequency mismatch functions of Woodworth [Radio-electronic systems: the fundamentals of construction and theory: A Handbook. Ed. 2nd, rev. and add. / Ed. J.D. Shirman. - M .: Radio engineering, 2007, p. 269-270].

Thus, after the cross-spectral post-correlation compensation of the masking signal, the ACF of the useful signal is released.

To clarify the second paragraph of the claims of the present invention, consider the structural diagram shown in figure 2. This block diagram consists of 1ms interfering signal transmitters and 1ps useful signals, moreover, the phase shift of the corresponding useful signals is determined by the corresponding orthogonal or short or long codes of the 2ps, 2ms communication lines (channels), receiving antenna 3, receiver 4, as well as corresponding inter-spectral post-correlation masking signal compensators.

The principle of operation of the system for each useful signal, the phase shift of which is determined by the corresponding orthogonal either short or long code, is similar to the principle discussed above.

The output of each of the cross-spectral post-correlation compensators of the masking signal, which ensures the separation of the autocorrelation function of only the corresponding orthogonal useful signal, since the WKF of the orthogonal signals are zero, will be the receiver output for the corresponding orthogonal useful signal.

To explain the third paragraph of the claims of the present invention, consider the structural diagram shown in figure 3. This block diagram consists of transmitters of the interfering signal 1ms and 1ps useful signals, and the phase manipulation of the corresponding useful signals is determined by the corresponding complementary codes from the group of codes, 2ps, 2ms communication lines (channels), receiving antenna 3, PS 4 receiver, and also corresponding inter-spectral post-correlation masking signal compensators. The results of the corresponding inter-spectral post-correlation compensations of the masking signal included in one group are summed, which forms the autocorrelation function of the corresponding group of complementary useful signals and is the output of the receiver for the corresponding group of useful signals. The principle of the system for each useful signal, the phase shift of which is determined by the corresponding complementary codes from the group of codes, is similar to the principle discussed above.

To confirm the practical possibility of obtaining the technical result indicated by the applicant, FIGS. 4 to 12 show the results of mathematical modeling of the claimed method.

As signals, phase-manipulated binary codes (PMS) signals whose phase structure was determined by modified Hadamard matrices [Lituk V.I., Lityuk L.V. Methods of digital multiprocessor processing of ensembles of radio signals. - M.: SOLON-PRESS, 2007, pp. 389-407]. As a conventional designation for the FMS, we will use the generally accepted designation: plus (+) and minus (-), when the phase of the code is zero and one hundred and eighty degrees, respectively. During the simulation, without loss of generality, eight-element FMS were studied, of which PS and MS were composed [Lityuk V.I., Lityuk L.V. Methods of digital multiprocessor processing of ensembles of radio signals. - M.: SOLON-PRESS, 2007, p. 400]. The following FMS were chosen as PS, the numbering of which was preserved according to the source: S1 = [++ - + --- +]; S6 = [- ++++ - ++], and as an MS, S7 = [- ++ --- +++]; S4 = [+++ - ++ - +], orthogonal to the corresponding PS. Temporal diagrams of the PS are shown in Fig. 4, and the corresponding amplitude-frequency spectra are shown in Fig. 5. For the simultaneous transmission of two complementary PSs on different frequency channels, the mismatch of their carrier frequencies was chosen, which is inverse to one time FMS discrete.

The normalized result of the processing module by the correlator of the corresponding PS is shown in Fig.6.

To ensure the secrecy of the transmitted information, it is necessary to form powerful orthogonal MSs that provide masking of PSs even with a frequency detuning exceeding the MS spectrum width, i.e. in generated out-of-band radiation. To do this, we considered an MS whose power is 34 dB higher than the power of the corresponding PS. The time plots of the MS and the corresponding amplitude-frequency spectra are shown in Fig.6. Temporal plots of the additive mixture of MS with PS with a shift of the carrier frequency of the PS by half-width, width and one and a half widths of the MS spectrum and the corresponding amplitude-frequency spectra are shown in Figs. 8-10, respectively. Comparison of the time and spectral diagrams of the additive mixture of weak PS and powerful MS (Figs. 8-10) shows their complete coincidence with the time and spectral diagrams of one MS (Fig. 7), i.e. PSs against the background of powerful MSs are not detected either in the time or in the spectral regions. Figure 11 shows the modules of the autocorrelation functions of the PS after the interspectral post-correlation compensation of the MS with the frequency offset between the MS and the PS by half-width, width and one and a half widths of the spectrum of the MS, respectively. Comparison of the modules of the autocorrelation functions of the PS after the interspectral post-correlation compensation of the MS when the carrier frequency of the PS is shifted by half the width, width and one and a half widths of the spectrum of the MS, respectively, shows complete suppression of the interfering signal. Some differences in the autocorrelation functions of the PS after the inter-spectral post-correlation compensation of the MS with a detuning that is not equal to the width of the interference spectrum are due to the presence of correlation noises of useful signals when they are processed using the first and second reference signals. These correlation noises of useful signals can be determined using the time-frequency mismatch functions of Woodworth [Radio-electronic systems: the fundamentals of construction and theory: Handbook. Ed. 2nd, rev. and add. / Ed. J.D. Shirman. - M .: Radio engineering, 2007, p. 269-270]. From the time-frequency mismatch functions of Woodworth, it follows that with a frequency detuning equal to the width of the MS spectrum, there are no correlation noises of useful signals. This effect can be observed in the second plot of Fig. 11, because The ACF of the useful signal in Fig. 6 completely coincides with the data. However, even with a frequency detuning exceeding one and a half the width of the MS spectrum, the distortions of the autocorrelation functions of the PS can be neglected.

Therefore, the claimed method provides not only an increase in the secrecy of the transmission of binary useful signals with an extended spectrum through modulation using orthogonal or complementary codes due to the simultaneous emission of powerful orthogonal masking signals, but also reduces the ability to decipher the structure of the PS.

Claims (3)

1. A method of transmitting and receiving binary signals, which consists in the simultaneous emission of the amplitude and phase manipulated useful and masking signals, forming their additive mixture at the input of the receiver, and subsequent suppression of the masking signal during reception, characterized in that they use the radiation of a masking signal that does not not only differs from the useful signal in polarization, spatial, temporal and frequency parameters, but it also covers the range of variation of the radiation frequency of each useful signal additionally form two reference signals with equal amplitudes and initial phases, all parameters of the first of them coinciding with the parameters of the emitted useful signal, while all parameters of the second of them, except the carrier frequency, which is mirror relative to the carrier frequency of the masking signal, also coincide with the parameters radiated useful signal, form cross-correlation functions between the received additive mixture of useful and masking signals and the corresponding reference signals, and then from the first th cross-correlation functions of the second subtracted, the subtraction result, which is the autocorrelation of the desired signal, the receiver will yield, thus forming two reference signals, the two cross-correlation functions and their subsequent subtraction is compensated mezhspektralnoy poslekorrelyatsionnoy masking signal. 2. The method according to claim 1, characterized in that the formation of an additive mixture of useful and masking signals at the input of the receiver is carried out by simultaneously emitting the useful and masking signals, and the phase manipulation of the useful signals is determined by orthogonal, either short or long codes, while for the corresponding useful signal, the corresponding interspectral post-correlation compensation of the masking signal is carried out, and the result of the corresponding subtraction, which is the output of the receiver as for the respective orthogonal useful signal is the autocorrelation function of the corresponding orthogonal useful signal. 3. The method according to claim 2, characterized in that the phase manipulation of the corresponding signal from the group of useful signals is determined by the corresponding complementary code from the group of codes, and the results of the corresponding inter-spectral post-correlation compensations of the masking signal included in one group are summed, which forms the autocorrelation function of the corresponding group complementary useful signals and is the output of the receiver for the corresponding group of useful signals.

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