RU2248102C1 - Method for autocorrelation receiving of noise-like signals - Google Patents

Abstract

FIELD: radio engineering.

SUBSTANCE: method includes multiplying received signal with support signal, duration of received signal is measured, frequency detection of received signal is performed, to separate moments of jump-like change of phase, number and value of clock periods is determined, support signal is formed by delaying received signal for time τ_d1=K₁τ_c divisible by period τ_c, total voltage is singled out, multiplied with received signal, delayed for time τ_d2=K₂τ_c, divisible by clock period τ_c, voltage of frequency difference is singled out, multiplied with received signal, delayed for time τ, which is periodically changed on basis of linear law, low-frequency voltage is singled out, proportional to autocorrelation function, and compared to threshold level, if threshold level is exceeded, cyclic displacement is measured, on basis of which code structure of received signal is determined.

EFFECT: higher efficiency.

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Description

The proposed method relates to radio engineering and can be used in digital communication systems, in particular, in devices for synchronizing and receiving phase-shifted (PSK) signals.

Known methods and devices for receiving noise-like signals (ed. Certificate of the USSR No. 177471, 451166, 491187, 543194, 860276, 1417206; RF patents No. 2085036, 2011299, 2106068, 2168869, 2121756; US patents No. 4146841, 4811363, 4912422; FR patents No. 2646255, 3935911; Petrovich N.P. et al. Communication system with noise-like signals. - M .: Sov. Radio, 1969, p. 94, Fig. 38a; J. Spilker. Digital satellite communications. - M. : Communication, 1979, p. 281; Varakin L.E. Communication systems with noise-like signals .-- M .: Communication, 1985, p. 18, fig. 1.9.c and others).

Of the known methods closest to the proposed is the "Method of autocorrelation reception of noise-like signals" (RF patent No. 2121756, H 04 B 1/20, 1988), which is selected as a prototype.

This method consists in multiplying the received signal with the reference signal and integrating the resulting product, the reference signal being formed by pairwise multiplying the delayed elements of the received signal and the received products, the delay of the elements of the received signal is carried out at time intervals that are multiples of the clock period, and the multiplicity coefficients correspond to the summed digits in a recurrence ratio, and each delayed value or the resulting product is used as a multiplier only once. The method provides the reception of noise-like signals only with a known recurrence ratio.

However, the implementation of the known method is possible only with an a priori knowledge of the recurrence ratio and the clock period of the received signals.

An object of the invention is the provision of receiving noise-like signals with a priori unknown internal structure.

The problem is solved in that according to the method of autocorrelation receiving noise-like signals, which consists in multiplying the received signal with a reference signal, measure the duration of the received signal, carry out frequency detection of the received signal, thereby highlighting the moments of phase jump, determine the number and size of clock periods, the reference signal formed by delaying the received signal by time τ _P1 = K ₁ τ _e multiple of the clock period τ _e, allocate total power, eremnozhayut it with the received signal delayed by time τ _s2 = R ₂ τ _e multiple of the clock period τ _e is isolated voltage of the frequency difference, multiply it with the received signal delayed by time τ, which is periodically changed in a linear fashion, emit a low-frequency voltage, proportional to the autocorrelation function, compare it with a threshold level, if the threshold level is exceeded, the cyclic shift will be changed, by which the code structure of the received signal is determined.

The structural diagram of a device that implements the proposed method is presented in figure 1. Timing diagrams explaining the essence of the proposed method are depicted in figure 2.

The device comprises a frequency detector 2, a pulse counter 3, a first arithmetic unit 4, the second input of which is connected to the input of the device through a signal duration meter 1, a first scaling multiplier 5, a first delay line 7, the second input of which is connected to the input of the device , the first multiplier 8, the second input of which is connected to the input of the device, the first band-pass filter 9, the second multiplier 11, the second input of which through the second delay line 10 is connected to the input of the device and the second scaling multiplier 6, the second band-pass filter 12, the second input of which through the third delay line 14 is connected to the input of the device and the output of the sawtooth generator 13, a low-pass filter 16, a threshold block 17, a key 18, the second input of which is connected to the output of the delay line 14 , the second arithmetic unit 19, the second input of which is connected to the output of the first arithmetic unit 4, and the registration unit 20, the second and third inputs of which are connected to the output of the meter 1 signal duration and arithmetic unit 4. Essentially The method of autocorrelation reception of noise-like signals is as follows.

Suppose that a pseudo-random sequence (PSP) is used as the modulating function, the symbols of which are described by the recurrence relation

x _i = a ₁ x _i-1 ⊕ a ₂ x _i-2 ⊕ ... ⊕ a _m x _im ,

where a _i = {0,1} are the coefficients of the forming polynomial

A (X) = X ⁰ ⊕ a ₁ X ¹ ⊕ a ₂ X ² ⊕ ... ⊕ a _m X ^m ,

⊕ is the addition sign modulo two, m is the bit depth of the pseudo-random sequence, the period of which is defined as N = 2 ^m -1.

To transmit such a sequence M (t) through the communication channels (FIG. 2, b), the high-frequency harmonic oscillation is manipulated in phase (FIG. 2, a)

U _c (t) = V _c × cos (ω _c t + φ _c ), 0≤ t≤ <T _c

where V _c , ω _c , φ _c , T _c - amplitude, carrier frequency, initial phase and duration of high-frequency oscillations;

the result is a phase-shift (QPSK) signal (noise-like signal) (figure 2, c)

U ₁ (t) = V _c × cos (ω _c t + φ _c (t) + φ _c ), 0≤ t≤ T _c

where φ _k (t) = {0, π} is the manipulated phase component that displays the phase manipulation law in accordance with the modulating code M (t) (PSP) (Fig. 2, b), and φ _k (t) = const for kτ _e <t <(k + 1) τ _e , and can change stepwise at t = kτ, i.e. at the borders between elementary premises (k = 1,2 ... N-1); τ _e , N - the duration and number of chips that make up the signal with a duration of T _c (T _c = Nτ _e ).

At the receiving location, the QPSK signal is supplied to the inputs of the meter 1 of the signal duration, frequency detector 2, multiplier 8, delay lines 7, 10, 14.

At the output of the frequency detector 2, short bipolar pulses are generated (Fig. 2, d), the temporary position of which corresponds to the moments of the phase change of the received PSK signal U ₁ (t) (Fig. 2, c). These pulses are fed to the input of a 3-pulse counter, where the number v of phase jumps is calculated. The following relationship exists between the number of phase jumps v and the number N of chips

ν = 0.5 × (N-1).

The number of phase jumps ν, counted by the counter 3, is fed to the first input of the arithmetic unit 4, the second input of which is the signal duration T _c measured by meter 1. In the arithmetic unit 4, the duration of elementary parcels τ _e (the clock period) is determined

At the same time, the received QPSK signal U ₁ (t), (0 (Fig. 2, c) is supplied to the first input of the multiplier 8. The value of τ _e , through the scaling multipliers 5 and 6, is supplied to the control inputs of the delay lines 7 and 10, respectively, where delays:

τ _z1 = k ₁ × τ _e , τ _z1 = k ₂ × τ _e ,

multiples of the clock period τ _e At the second input of the multiplier 8, the received QPSK signal is delayed by a value of τ _s1 (Fig.2, d), which is a reference signal

U ₂ (t) = U ₁ (t-τ _z1 ) = V _c × cos [ω _c (t-τ _z1 ) + φ _k (t-τ _z1 ) + φ _c ], 0≤ t≤ T _c .

The output of the multiplier 8 forms the following oscillation

U ₃ (t) = U ₃ × cos [2ω _c t-ω _c τ _z1 + φ _k (t) -φ _k (t-τ _z1 ) + 2φ _c ] + U ₃ × cos [ω _c τ _z1 + φ _k (t) -φ _k (t-τ _z1 )], 0≤ t≤ T _c

;Where

;

K is the transmission coefficient of the multiplier from which the band-pass filter 9, tuned to 2ω _c , the total voltage is allocated (figure 2, e)

UΣ t) = U ₃ × cos [2ω _c t-ω _c τ _z1 + φ _k (t) -φ _k (t-τ _z1 ) + 2φ _c ], 0≤ t≤ T _c ,

which is fed to the first input of the multiplier 11, to the second input of which the received QPSK signal is delayed by a value of τ _s2 (FIG. 2, g) by the delay line 10

U ₄ (t) = U ₁ (t-τ _z2 ) = V _c × cos [ω _c (t-τ _z2 ) + φ _k (t-τ _z2 ) + φ _c ], 0≤ t≤ T _c .

The output of the multiplier 11 forms the following oscillation

U ₅ (t) = V _{5 × cos [3ω c t-ω} c (τ _P1 + τ _{s2) + φ k (t) +} φ k (t-τ _P1) + φ _k (t-τ _s2) + 3φ _c] + V _{5 × cos [ω c t + ω} c (τ _{s2 P1} -τ) + 3φ _c] + V _{5 × cos [ω c t + ω} c (τ _{s2 P1} -τ) + φ _k (t) + φ _k (t-τ _z1 ) -φ _k (t-τ _z2 )], 0≤ t≤ T _c ,

;Where

;

from which the band-pass filter 12 tuned to ω _c , the voltage of the frequency difference is allocated (figure 2, h)

U _p (t) = V _{5 × cos [ω c t + ω} c (τ _{s2 P1 -τ) + φ k (t) +} φ k (t-τ _P1) -φ _k (t-τ _s2)], 0≤ t≤ T _c ,

the manipulated phase of which is as follows

φ _kp (t) = φ _k (t) + φ _k (t-τ _z1 ) -φ _k (t-τ _z2 ) = φ _k (t-θ τ _e ), where θ is the cyclic shift expressed by the number of clock periods (elementary premises).

The voltage U _p (t) from the output of the band-pass filter 12 is supplied to the first multiplier 15, to the second input of which the received PSK signal is delayed by a value of τ using the delay line 14, which is periodically tuned according to a linear law using a sawtooth voltage generator 13,

U ₆ (t) = U ₁ (t-τ) = V _c × cos [ω _c (t-τ) + φ _k (t-τ) + φ _c ], 0≤ t≤ T _c ,

where τ is a variable value of the delay value of the delay line 14. The output of the multiplier 15 produces the following voltage

₇ U (t) = V _{7 × cos [2ω c t + ω} c (τ _s2 -τ _{P1 + τ) + φ k (t} ) + φ k (t-θ τ _e) + 2φ _c] + V × ₇ cos [ω _c (τ _s2 -τ _{P1 + τ) + φ k (t} -θ τ _{e) -φ k (t-τ)} ], 0≤ t≤ T c,

.Where

.

The low-pass filter 16 emits a low-frequency voltage proportional to the autocorrelation function,

U _n (t) = V _n × cos [ω _c (τ _s2 -τ _{P1 + τ) + φ k (t} -τ)], 0≤ t≤ T c.

which is compared with the threshold level of V _pores in the threshold unit 17. The threshold voltage of V _{pores is} exceeded only at the maximum voltage value U _n (t), which is obtained when the following condition is met

τ ₀ = θ τ _e, ω _c × (τ _s2 -τ _P1 + θ τ _e) = 2π K, K = 1, 2, 3 ...

If the threshold level V threshold is exceeded _{, the} threshold unit 17 generates a constant voltage, which is supplied to the control input of the sawtooth voltage generator 13, stopping its adjustment, and to the control input of the key 18, opening it. In the initial state, the key 18 is always closed.

With this value of the delay value τ ₀ = θ τ _e , which corresponds to the maximum of the autocorrelation function R (τ), through public key 18 it enters the arithmetic block 19, where the value of the duration τ _{e of the} elementary messages from the output of the arithmetic block 4 is also received. In the arithmetic block 19 cyclic shift is determined

,

,

which is fixed by the registration unit 20, where the measured values of the duration τ _{e of} elementary packets and the duration T _{c of the} received FMN signal are also recorded. The specified shift establishes a unique correspondence between the code structure of the received QPSK signal and the conversion function, which is specified by the parameters τ _z1 and τ _z2 :

θ ↔ θ [A (X), B (X),]

where A (X) is the generating polynomial that determines the code structure of the received PSK signal;

и

, а коэффициент В₀=1.B (X) = B ₀ X ⁰ + B ₁ X ¹ + ... + B _n X ⁿ is a transformation function whose numbers of zero coefficients are defined as

and

, and the coefficient B ₀ = 1.

For example, for τ = 2τ _{P1 e} and τ _s2 = 3τ _e (θ = 8)

A (X) = X ⁰ ⊕ X ² ⊕ X ⁵ ;

B (X) = X ⁰ ⊕ X ² ⊕ X ³ .

By measuring the cyclic shift θ, the code structure (the law of phase manipulation) of the received PSK signal can be determined from the correspondence table.

Thus, the proposed method in comparison with the prototype provides the reception of noise-like signals with a priori unknown code structure. This is achieved by measuring the cyclic shift θ, which establishes a unique correspondence between the code structure of the received PSK signal and the conversion function.

Claims (1)

The method of autocorrelation receiving noise-like signals, which consists in multiplying the received signal with a reference signal, characterized in that the duration of the received signal is measured, frequency of the received signal is detected, thereby highlighting the moments of phase jump, determine the number and magnitude of the clock periods, the reference signal is formed by delay the received signal for a time τ _Z1 = K ₁ τ _e multiple of the clock period τ _e , the total voltage is isolated, it is multiplied with the received with a signal delayed for a time τ _Z2 = K ₂ τ _e , a multiple of the clock period τ _e , isolate the voltage of the frequency difference, multiply it with a received signal delayed for a time τ, which is periodically changed according to the linear law, isolate a low-frequency voltage proportional to the autocorrelation function compare it with a threshold level; when the threshold level is exceeded, a cyclic shift is measured, which determines the code structure of the received signal.

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