MXPA97005401A

MXPA97005401A - Satellite communications system that uses concatenated paral use

Info

Publication number: MXPA97005401A
Application number: MXPA/A/1997/005401A
Authority: MX
Inventors: Michael Hladik Stephen; Alan Check William; James Glinsman Brian; Fleming Fleming Robert Iii
Original assignee: General Electric Company
Priority date: 1996-07-17
Filing date: 1997-07-16
Publication date: 1998-04-01

Abstract

The present invention relates to a very small aperture terminal communications system for satellite communication, comprising: a plurality of very small aperture terminals, each comprising: a parallel concatenated encoder comprising a plurality of component encoders connected in a parallel concatenation, the parallel concatenated encoder applying a concatenated code parallel to a block of data bits received from a source, and generating component keywords therefrom, the parallel concatenated encoder comprises a keyword formatter for formatting the bits of the component keywords, to provide a composite keyword, a packet formatter for assembling data packets for transmission, each data packet comprises bits of at least one composite keyword, a modulator for receiving the data packets and providing modulated signals from the same, an up-converter to transfer the modulated signals to a carrier frequency, an interconnection to connect each terminal of very small opening to an antenna to transmit modulated signals to the satellite, and receive signals modulated from the satellite, a down-converter to transfer each signal received from the carrier frequency at an intermediate frequency, a demodulator for synchronization and demodulation of the received signals, a packet-to-keyword formatter for forming composite keywords received from the demodulated signals, and a composite decoder comprising a plurality of component decoders to decode composite keywords received

Description

SATELLITE COMMUNICATIONS SYSTEM USING PARALLEL CONCATENATED CODING Field of the Invention The present invention relates generally to satellite communications systems and, more particularly, to a very small aperture terminal satellite communications system employing parallel concatenated encoding on its internal or external links, or both. .

Background of the Invention There is an emerging market for satellite multimedia communications using very small aperture terminals (VSAT's), low cost. The advantages of using a smaller antenna than what is currently used in general practice in the industry today, include reduced reflector cost, lower boarding costs, reduced hardware and assembly work, and greater customer acceptance due to a less protruding appearance However, the use of a dish antenna with a smaller aperture may cause an undesirable reduction in the capacity of the network. This is due to many causes related to the reduced size of the antenna: (1) decreased power of the received and transmitted signals caused by the associated decrease in directional antenna performance; and (2) regulations of the Federal Communications Commission (FCC) that limit the power transmitted by a very small aperture terminal that uses an antenna smaller than a specified size, in order to limit the flow density of interference power in orbital slots of adjacent satellites. The use of a very small aperture terminal power amplifier with the same power output or less, in order to reduce the cost of the very small aperture terminal, also contributes to the decrease in network capacity, due to to power limitations. Unfortunately, it is difficult to obtain the desired large coding gain in short blocks of data (which are typical of some types of very small aperture terminal transmissions) to solve these problems, with the required bandwidth efficiency and decoder complexity. , using conventional coding techniques. In accordance with the foregoing, it is desirable to provide a satellite communications system that increases the capacity of the network when using very small aperture terminals with reduced antenna apertures, by decreasing the spectral density ratio of E./N. power-by-bit-to-noise required, with spectrally efficient techniques.

SUMMARY OF THE INVENTION In accordance with the present invention, a very small aperture terminal satellite communications network uses parallel concatenated encoding on its internal or external links, or both. In one embodiment, for short blocks of data that are typical of packet transmissions, credit card transactions, and compressed voice communications, convolutional codes joined in the queue, systematic, non-repetitive, are used as the component codes in such a case. Parallel concatenated coding scheme. For longer data blocks, which are typical of file transmission, the very small aperture terminal and the cube terminal of the network use repetitive systematic convolutional codes. In a preferred embodiment, the parallel concatenated coding techniques mentioned above are used in conjunction with spread spectrum modulation, resulting in a system that complies with the Federal Communications Commission's regulations regarding the total power spectral density of the signals transmitted, and that mitigates interference from adjacent satellites.

BRIEF DESCRIPTION OF THE DRAWINGS The features and advantages of the present invention will be apparent from the following detailed description of the invention, when read with the accompanying drawings in which: Figure 1 is a simplified block diagram illustrating a communications system of very small aperture terminal employing parallel concatenated coding, in accordance with the present invention. Figure 2 is a simplified block diagram illustrating a cube terminal of the very small aperture terminal satellite communication system employing parallel concatenated coding in accordance with the present invention. Figure 3 is a simplified block diagram illustrating a programmable encoder, useful in a very small aperture terminal communications system, in accordance with the present invention. Figure 4 is a simplified block diagram illustrating a programmable decoder, useful in a very small aperture terminal communications system, in accordance with the present invention.

Detailed Description of the Invention The invention described herein is a very small aperture terminal satellite communications system, which uses parallel concatenated encoding techniques, which involves, for example, parallel concatenated convolutional codes in parallel, and codes systematic repetitive concatenated parallel convolutions (that is, the so-called "turbo codes"), and their respective decoders. In particular, for concatenated parallel concatenated tail codes, a decoder comprising circular MAP decoding is employed, as described in Patent Application of the United States of America Number 08 / 636,742 by Stephen M. Hladik and John B Anderson, commonly assigned, co-pending, filed on April 19, 1996, and incorporated herein by reference. Parallel concatenated encoding is used in internal link transmissions (very small-to-cube aperture terminal) or in external link transmissions (very small aperture cube-to-terminal), or both links of a communications network Very small aperture terminal satellite. In addition, parallel concatenated coding can be used to provide error correction / detection coding for direct para-pair transmissions (very small aperture terminal-to-terminal aperture very small). In one embodiment, for short blocks of data that are typical of packet transmissions, credit card transactions, and compressed voice communications, non-repetitive systematic joined-in-the-tail codes are used, such as component codes in a schema of parallel concatenated coding. For longer data blocks, which are typical of file transmission, the very small aperture terminals and the cube terminal of the network use parallel concatenated coding comprising repetitive systematic convolutional codes. In accordance with the present invention, the use of these parallel concatenated coding techniques, in conjunction with extended spectrum modulation, provides a very effective solution to facilitate compliance with the Federal Communications Commission regulations mentioned above on satellite interference. adjacent, by decreasing the effective radiated power (ERP) required and the power spectral density of the transmitted signal. In addition, this combination mitigates interference from adjacent satellites. Figure 1 is a block diagram of a very small aperture terminal satellite communications system employing parallel concatenated coding in accordance with the present invention. This system essentially comprises a number of very small aperture terminals 10, a satellite 12 with a transmitter-receiver, and possibly a cube terminal 14. Communication within the very small aperture terminal network can be either in a single sense or in two directions, and can travel in a variety of trajectories: (1) very small aperture terminal-to-terminal aperture very small directly (ie, mesh connectivity) and (2) very small aperture terminal- A-terminal cube and / or cube-to-terminal terminal with very small aperture (ie, star connectivity). As shown in Figure 1, a very small aperture terminal 10 comprises signal processing of the transmitter 20, signal processing of the receiver 22 and an antenna 24. In accordance with the invention described herein, signal processing of the transmitter of the very small aperture terminal comprises the following: an input port 25 for accepting data from an information source 25; an encoder 28 that applies a concatenated code parallel to the blocks of data bits received from the source; a packet formatter 30 for generating a data packet (comprising one or more key words of the encoder 28), a synchronization bit pattern and control signaling bits; a modulator 32; an up converter 34 for transferring the modulated signal to the carrier frequency; a power amplifier 36; and a connection to the antenna 24 through an appropriate interconnection (e.g., a switch or duplex filter coactor). The signal processing of the very small aperture terminal receiver comprises: a low noise amplifier 40, a down converter 42 for transferring the received signal from the carrier frequency to an intermediate frequency, a demodulator 44 for synchronization and demodulation, a formatator packet-to-keyword 46, a decoder 48 suitable for the parallel concatenated code used by the transmitter, and an output port 49 for transferring received messages (i.e., blocks of data bits) to an information seat 50 For brevity, a detailed block diagram for only a very small aperture terminal is shown in Figure 1. The synchronization functions performed by the demodulator 44 include carrier frequency synchronization, frame synchronization, symbol synchronization, and, if necessary, carrier phase synchronization. Symbol synchronization is the process of estimating the best sampling per(ie, epoch of the symbol) for the output of the demodulator, in order to minimize the probability of a symbol decision error. Frame synchronization is the process of estimating the epoch of the symbol for the first symbol in a frame (for continuous transmissions) or packet (for non-continuous transmissions) of received data. For the case where the spread spectrum signals are transmitted by the very small aperture terminal, the very small aperture terminal modulator shown in Figure 1 includes the extension function; and the demodulator shown in Figure 1 includes the de-exten- sion function. Extended-spectrum techniques increase the bandwidth of the signal relative to the bandwidth of the modulated data signal by imposing an extension signal comprising chips (in the case of the direct-sequence extended spectrum) or jumps (in the case of the extended spectrum of frequency hopping) that are pseudorandom and independent of the data signal. In the extended direct sequence spectrum, the data signal is multiplied by a signal corresponding to a pseudorandom sequence of chips having the values of +1 or -1. The duration of the pulses of the chip is smaller than the symbol interval of the modulated data signal; consequently, the bandwidth of the resulting signal is greater than that of the original modulated signal. In the extended spectrum of frequency hopping, the carrier frequency of the modulated signal is percally changed, in accordance with a pseudorandom pattern. Again, the bandwidth of the distributed signal is greater than that of the original modulated signal. The de-spread in the demodulator is the process of removing the extension of the received signal. Typically, the demodulator correlates the received signal with a replica of the extension waveform, to de-spread a direct-sequence spread spectrum signal, whereas in an extended-frequency hopping system, it jumps the frequency of an oscillator in the downstream converter of the receiver, using the same pattern used by the transmission terminal to de-spread a frequency-extended extended spectrum signal. Typically, a filter is applied to the signal received after de-expansion to attenuate wideband noise and interference components in the recovered signal. In Figure 2 a block diagram of the cube terminal is presented. In accordance with the invention described herein, it comprises: input ports 51 for accepting data from one or more information sources 52; output ports 53 for transferring received messages (i.e., blocks of data bits) to one or more information entries 54; a bank of transmitter channel processors 56; a bank of receiver channel processors 58; a switch 60 for connecting each active source to a transmitter channel processor and for connecting each active receiver channel processor to the appropriate information seat or to a transmit channel processor; a memory 62; a controller 64 for controlling the flow of data through the switch; a combiner 66 for combining the signals generated by each transmitter channel processor in a signal; an up converter 68 for transferring the combined signals to the carrier frequency; a power amplifier 70 connected to the antenna by an appropriate interconnection (e.g., a duplex filter switch or communicator); an antenna 72; a low noise amplifier 74 which is coupled to the antenna by the aforementioned interconnection; a down converter 76 for transferring the received signal from the carrier frequency to an intermediate frequency (IF); and a signal splitter 78 for providing the received intermediate frequency signal, or possibly a filtered version of the received intermediate frequency signal to the receiver channel processor bank. The transmit channel processor shown in Figure 2 comprises: an encoder 80 which applies a concatenated code parallel to blocks of data bits received from a source; a packet formatter 82 for generating a data packet (comprising one or more key words of the encoder 80), a pattern of synchronization bits and control signaling bits; and a modulator 84. As with the very small aperture terminal, cube modulators include the extension function for the case in which the spread spectrum signals are transmitted through the cube. The receiver channel processor of Figure 2 comprises a demodulator 86, a packet-to-keyword converter 88 for selecting samples from the demodulator output to form the received received key words to the decoder for parallel concatenated codes, and a decoder 90 suitable for the parallel concatenated code used by the transmitter. The cube demodulators include many functions: synchronization, demodulation, and, for the case in which the cube receives extended spectrum signals, de-exhilaration. One function of the cube memory is to temporarily store data received from information sources or receiver channel processors, in case all the transmitter channel processors or output ports are busy when a message arrives at the switch 60. The memory also stores configuration parameters of the network and necessary operational data. In an alternative embodiment of the present invention, an outer code is used in concatenation in series with the parallel concatenated code (PCC) (internal); an associated external decoder is also connected in concatenation in series with the decoder for the inner parallel concatenated code. Additionally, the very small aperture and cube terminal equipment can use a flexible encoder / decoder system, which can be programmed, to implement many options: (1) parallel concatenated encoding, as described hereinabove; (2) an outer code in concatenation in series with an internal parallel concatenated code (PCC), as described above in the present; (3) serial concatenated encoding comprising an outer encoder and only one component encoder of a parallel concatenated code encoder; (4) a conventional convolutional code or block code only (that is, without serial or parallel concatenation). Figure 3 illustrates a block diagram of a flexible, programmable coder that implements these four coding options. As shown, the flexible encoder, which can be programmed, comprises an encoder 100 for parallel concatenated codes, an encoder 102 for an external code, and five switches S1-S5. The encoder 100 for parallel concatenated codes comprises N encoders, N-l interpaging, and a keyword formatter 106. Table I, below, summarizes the positions of the switches for different modes of operation of the encoder.

Table I Figure 4 is a flexible, programmable decoder block diagram that implements the decoders for the four encoder modes presented hereinabove. This composite, programmable decoder comprises a decoder 110 for parallel concatenated codes, a threshold decision device 112 for implementing a decision rule, a decoder 114 for an external code, and six switches S1-S6. Assuming that the output of the decoder 110 is the probability that the value of the decoded bit is equal to zero, an important decision rule is: If the output is greater than 1/2, then decide that the decoded bit is zero; if it is less than 1/2, then assign the value one; if it is equal to 1/2, then arbitrarily assign a value. The decoder 110 for parallel concatenated codes also comprises a composite keyword for the keyword converter component 116, N component decoders, N1 interpaging and two identical de-interpaging 118. Each de-interpager has a reordering function that returns a sequence of data elements that have been altered by the interpaging Nl connected in series to their original order. Table II, below, summarizes the positions of the switches for different modes of the decoder operation. (In the table, X denotes the condition "it does not matter", that is, the switch can be in any position).

Table II The very small aperture terminal uses different codes (for example, PCCC codes, PCCCs joined in the queue, systematic repetitive convolutional, systematic non-repetitive, block convolutional) in different combinations (for example, modes 1, 2, 3, and 4), depending on the communication application and the required transmission speeds. When convolutional codes are used in any of the modes described hereinabove, the programmable encoder of Figure 3 may also include drilling by a known pattern, to increase the speed of the resulting code, and the decoder that can be programmed. program of Figure 4 may also include the associated function of defoaming. When perforated convolutional codes are used as the component codes in the parallel concatenated coding, the keyword formatter of Figure 3 removes the code bits from the component keywords, in accordance with the desired puncturing patterns. In this case, the keyword composed of the decoder of the parallel concatenated code for the component keyword converter inserts neutral values for the punched bits in the component keywords to which it outputs the component decoders. Note that in Mode 3 or Mode 4, the encoder switches S4 and S5, and the decoder switches SI and S2 are all set to position 0. Therefore, Figures 3 and 4 show the drilling unit 140 , and the spoilage unit 142, respectively, in phantom to implement these piercing and wiping functions, respectively, when a convolutional code punched in Mode 3 or Mode 4 is used. In a preferred embodiment of this invention, codes are used. convolutional as the component codes in an internal parallel concatenated code, and a block code (for example, a Reed-Solomon code or BCH code) is used as an outer code in serial concatenation. In a preferred embodiment in which the spread spectrum signals are transmitted by very small aperture terminals, a random channel access protocol such as ALOHA is used, in conjunction with code division multiple access. The cube receiver uses a number of demodulators for each extension code, in order to receive time overlap signals that use delayed versions in time of the same extension sequence. Each demodulator for a given extension sequence modulates a signal using a time change different from that extension sequence. Also in the preferred modality, one or more extension sequences are reserved for use by very small aperture terminals for specified periods of time on an assigned basis, in order to provide higher quality channels with higher performance. The reservation requisitions of very small aperture terminals and assignments are processed by a network controller that is connected to the cube terminal. In a preferred embodiment using spread spectrum signals and the programmable encoder and decoder described hereinabove, the system associates an extension sequence with a particular error correction code to allow different signals to use different error correction coders simultaneously . Since each sequence of extension of the detected signal is identified by a corresponding demodulator, the receiver can appropriately configure the programmable decoder for each detected signal. This mode of network operation is useful for supporting many applications that have different error correction coding requirements without the need for additional control signaling. A circular MAP decoder useful as the decoder of the component in the commonly assigned, co-pending US Patent Application No. 08 / 636,742 is described in Figure 4. The circular MAP decoder can release both an estimate of the encoded data block and reliability information to a data seat, for example, a speech synthesis speech processor for use in transmission error concealment or a protocol processor for data packet as a measure of block error probability for use in repeated demand decisions. As described in commonly assigned U.S. Patent Application No. 08 / 636,732, co-pending from Stephen M. Hladik and John B. Anderson, filed on April 19, 1996 and incorporated herein by reference, the decoder Circular MAP is useful for decoding convolutional codes joined in the queue, particularly when these are used as component codes in a parallel concatenated coding scheme. A circular MAP decoder for interleaved error correction codes employing queuing in accordance with U.S. Patent Application Number 08 / 636,742 produces indistinct output output information. The circular MAP decoder provides an estimate of the probabilities of the states in the first stage of the interleaving, which are likely to replace the a priori knowledge of the start date in a conventional MAP decoder. The circular MAP decoder provides the probability extension of the start state in one of two ways. The first includes a solution to a specific value problem for which the resulting specific value is the probability extension of the desired start state; with knowledge of the start setting, the circular MAP decoder performs the rest of the decoding in accordance with the conventional MAP decoding algorithm. The second is based on a recursion for which the evasive approximations converge to an extension of the start state. After sufficient successive approximations, a state is known in a circular sequence of states with a high probability, and the circular MAP decoder performs the rest of the decoding according to the conventional MAP decoding algorithm which is set in "Optimal Decoding of Linear Codes • for Minimizing Symbol Error Rate, "by Bahl, Cocke, Jelinek and Raviv, IEEE Transactions on Information Thssory, pp. 284-287, March 1974. The objective of the conventional MAP decoding algorithm is to find the conditional probabilities: P { state m at time t | receives channel outputs y_ r • • • r Y ^} * The term L in this expression represents the length of the data block in unidadee of the number of encoder symbols. (The encoder for a code (n, k) operates on the k-bite input symbol to generate n-bit output symbols.) The term t. is the output of the channel (symbol) at time t. The MAP decoding algorithm really first finds the probabilities:? ^ M) = P { St = m; YL1}; (l) that is, the link probability that the encoder establishes at time t, »S, is m and the set of channel outputs y 1 = is received. { and ?, ..., and L} . These are the desired probabilities multiplied by a constant (P. {Y. L}, the probability of receiving the set of channel outputs { Y1 / ..., yL.}. Now define the elements of a matrix r by r Xr (ij) = P. {state j at time t, and t | state i at time t-1.} Calculate the matrix r as a function of the transition probability of the R channel ( Yt, X), the probability p (m / m ') that the encoder makes a transition from state m to m' at time t, and the probability q (X / m '/ m) that the symbol output of the encoder is X taking into account that the previous state of the encoder is m 'and that the current state of the encoder is M. In particular, each element of r is calculated by adding all the possible X outputs of the encoder as follows: yt <m », m) =? p. (m / m ') q. (X / m', m) R (Y., X). (2) X t t * The MAP decoder calculates the L of these matrices, one for each interlacing stage. These are formed from the received channel output symbols and the nature of the interleaving branches for a given code.

Then define the probability elements of the link M as a vector in a row < * t using < * t (j) = P { state j at time t; and? J • Y > (3) and the conditional probability elements M of a column vector Bt by ß (j) = p. { yt + 1, ..., and / state j at time t} (4) for j - 0,1 ..., (M-l) where M is the number of states of the encoder. (Note that matrices and vectors are denoted in the present by using bold.) The steps of the MAP decoding algorithm are as follows: (i) Calculate l? by recursion forward: at = ott_1rt, t = 1, ..., L (5) (ü) Calculate Rl t. . . , R by recuring backwards: ßt = rt + lßt-l 't = L-l, ... l (6) (iii) Calculate the elements of? by:? fc (i) = at (i) ßt (i) all the i, t = l, ..., L (7) (iv) Find the related amounts as necessary. For example, let AtD be the set of states St =. { 8 ^, 8 ^, ..., Skmt} such that the element jth of St, S t, is equal to zero. For a conventional non-recursive interleaving code, SDt = dD, the data bit jth at time t. Therefore, the indietinta output of the decoder is: where P { YL1} = S? T (m) and m m is the index corresponding to a state St < The firm decision of the decoder or the output of the decoded bit is obtained by the application of s ?. { 3t - OlY1 ',} to the following decision rule:? dj = 0 Pid ^ = 0 J Y ^} > 1; < 2-d3t = 1 E > 1/2, then? Dt = 0; Yep { dD. = AdDt = 1; otherwise, arbitrarily assign to dD the value 0 or 1. As another example of a related quantity for the previous (in) pae (iv), the probability matrix s. It comprises elements defined as follows: st (i / j) = p < st-? = i '' st = j; ? L? = at -? (i) Yt (i'i ßt (j) These probabilities are useful when determining the a posteriori probability of the output bits of the encoder.These probabilities are also useful in the deciphering of recursive convolutional codes. the standard application of the MAP decoding algorithm, ee initializes the forward recursion by means of the vector aQ = (1,0, ..., 0), and the backward recursion is initialized by fiL = (1,0, .. .0) T. These initial conditions are based on the assumptions that the initial state of the encoder S. = 0 and that its final state SL = 0. A modality of the circular MAP decoder determines the probability extension of the initial state by means of to solve a specific value problem as follows: Let a, R., T and? remain as before, but take the aQ and ßQ as follows: Place ß to the column vector (111 ... 1) T. Let o_ is an unknown variable (vector).

Then, (i) Calculate r for t = 1,2, ... L in accordance with equation (2). (ii) Find the largest specific value for the product of the matrix r? G, ... GL. Normalize the corresponding specific value so that its components add up to one unit.

This vector is the solution for a_. The specific value is P { YL ?} - (üi) Form the successive a through the forward recursion established in equation (5). (iv) Starting from ßt, initialize as in the previous one, form the fit by means of the backward recursion established in equation (6). (v) Form the? t as in (7), as well as other desirable variables, such as, for example, the indistinct output P { d? 3Jt =? | YL ,} or the sfc probability matrix described hereinabove. The unknown variable aQ satisfies the equation of the matrix Based on the fact that this formula expresses a relationship between probabilities, the product of matrices r. on the right it has the largest specific value equal to P { x ..}. , and that the corresponding specific vector must be a probability vector. With the initial ß = (111 ... 1) T, equation (6) gives ß .. In this way, the repeated applications of this backward recursion give all the ß. Once Q is known and it is placed ß, all calculations in the circular MAP decoder follow the conventional MAP decoding algorithm. An alternative modality of the circular MAP decoder determines the probability distributions of state by means of a recursion method. In particular, in one modality (the dynamic convergence method), the recursion continues until the convergence of the decoder is detected. In this method of recursion (or dynamic convergence), steps (ii) and (iii) of the specific vector method previously described in the preend are replaced as follows: (ii.a) Starting with an aQ equal to (1 | M) , ..., 1 | M), where M is the number of states in the interlace, calculate the recursion forward L times. Normalize the results so that the elements of each new add up to the unit. Hold all vectoree L at. (ii.b) Leave a_ equal to a from the previous point and, starting at t = 1, calculate the first probability vectors L in a again. * Ml This is, calculate o (m) = S at l (i)? T (i, m) for = i = 0 0,1 ..... M-1 and t = 1/2 / .... L - "where L _, - ee a minimum number of interlaced stages. Normalize as before. Retain only the most recent set of L a. found through recureion in steps (ii.a) and (ii.b) and G¡L -n previously found in the paeo (ii.a). (ii.c) Compare the < * L-n of paeo (ii.b) with the previously found set of paeo (ii.a). If the correspondence element M of the new and previous mail is within tolerance ranges, proceed to the payment (iv) set forth hereinabove. Otherwise, continue to step (ü.d). (ii.d) Let t = t + 1 and calculate t = c ___ t't. Normalize as before. Retain only the most recent set of L calculated and the a previously found in step (ii.a). (ii.e) Compare the new ones to the previously found set. If the Ms of the new and previous t are within a tolerance range, proceed to step (iv). Otherwise, continue with step (ii.d) if the two most recent vectors are not within the tolerance range and if the number of recursions does not exceed a specified maximum (typically 2L); proceed to paeo (iv) and i is not like that. This method continues later with steps (iv) and (v) given earlier in the preamble with respect to the specific vector method for producing the indietintae output and decoded output loe of the circular MAP decoder. In another alternative embodiment of the circular MAP decoder described in the North American Patent Application Number 08 / 636,742, the reclosure method is modified so that the decoder need only process a predetermined, fixed number of interleaving steps for a second time, that is, a previously determined depth of envelopment. This is advantageous for implementation purposes because the number of calculations required for decoding is the same for every code message block. As a result, the complexities of hardware and software are reduced. One way to estimate the envelopment depth required for MAP decoding of a convolutional code attached to the queue is to determine it from hardware and software experimentation, requiring that a circular MAP decoder with a wrapping depth be required. variable and that experiments are conducted to measure the error of decoded bits versus Et, / No For depths of envelopment that increase successively. The minimum enclosure depth of the decoder that provides the minimum probability of decoded bit error is found for an Eb / Not specified when additional increases in wrapping depth do not decrease the probability of error. If a decoded bit error rate that is greater than the minimum achievable at a specified E / N is tolerable, it is possible to reduce the required number of interleaving steps processed by the circular MAP decoder. In particular, the wrapping depth described hereinabove can be terminated simply when the desired average probability of the bit error is obtained. Another way to determine the wrapping depth for a given code is by using the distance properties of the code. For this purpose, it is necessary to define two different decoder decision depths. As noted in the present, the term "correct trajectory" refers to the sequence of states for a path through interleaving that results from coding a block of data bits. The term "incorrect subset of a node" refers to the set of all incorrect branches (interlaces) outside of a correct path node and its descendants. The two deflection depths defined subsequently depend on the convolutional encoder. The decision depths are defined as follows: (i) Define the forward decision depth for an error correction and, LF (e), to be the first depth in the interlacing in which all the trajectories in the incorrect subset of an initial node of correct trajectory, whether they join later towards the correct trajectory or not, are more from a Hamming distance 2e from the correct trajectory. The significance of LF (e) is that if there are fewer errors forward of the initial node, and you know that the coding has started there, then the decoder must decode correctly. J.B. Anderson and K. Balachandran provided a formal tabulation of forward decision depths for convolutional codes in "Decision Depths of Convolutional Codes," IEEE Transactions on Information Theory, volume IT-35, p. 455-59, March 1989. Many properties of LF (e) are decribed in this reference and also J.B. Anderson and S. Mohán in Source and Channel Coding - An Algorithmic Approach, Kluwer Academic Publishers, Norwell, MA, 1991. The main thing among these properties is that there is a simple linear relationship between LF and e; for example, with a range of 1/2 codes, LF is approximately 9.08e. (ii) Then define the decision depth without joining for the correction of the error e, LU (e), so that it is the first depth in the interlace where all the paths in the interlace that never touch the correct path are at least of a Hamming diet of 2e of the correct trajectory. The significance of Lü (e) for decoding Circular MAP of indistinct decision is that the probability of identifying a state in the actual transmitted path is high after the decoder processes the interleaving steps Li (e). Therefore, the minimum envelopment depth for circular MAP decoding is LU (e). The calculations of the depth LU (e) show that it is always larger than LF (e) but that it obeys the approximate law. This implies that the minimum envelopment depth can be estimated as the forward decision depth LF (e) if the decision depth is unknown without joining a code. By finding the decision depth without joining a minimum for a given encoder, we find the smallest number of interleaving stages that must be processed by a practical circular decoder that generates indistinct outputs. J.B. Anderson and K. Balachandran gave an algorithm to find LF (e), the forward decision depth, in "Decision Depths of Convolutional Codes", cited earlier in this. To find LU (e): (i) Extend the interleaving of the code from left to right, starting with all the interlacing nodes simultaneously, except for the zero state. (ii) At each level, delete any trajectories that join the correct path (all zero); do not extend any path outside the correct node (zero). (iii) At the level, find the smallest Hamming distance, or weight, between the trajectories that end at nodes at this level. (iv) If this minimum distance exceeds 2e, stop. Then, LU (e) = k. As described in the Patent Application of the United States of America Number 08 / 636,742, experimentation through computer simulation led to ineffective results: (1) env envelopment processing improves decoder performance; and (2) the use of a wrapping depth of LU (e) + LF (e) »2 LF (e) improves performance in a meaningful way. Hence, a preferred modality of the circular MAP decoder algorithm based on recursion comprises the following steps: (i) Calculate r. for t = 1/2 / ... L in accordance with equation (2). (ii) Starting with an a_ equal to (1 | M, ... / l | M), where M is the number of states in the interlace, calculate the forward recursion of equation (5) (L + L ) times for u = 1/2 / ... (L + L) where L is the enclosure depth of the decoder. The interlace level index t takes the values ((u-1) mod L) + 1. When the decoder wraps around the received sequence of the symbol of the channel, the a is treated as a_. Normalize the results so that the elements of each ot. add up to the unit. Retain the vectoree to the most recent L found by this recureion. (iii) Starting with an initial ß equal to T (1 /.../ 1), calculate the backward recursion of the equation (6) (L + Lw) times for u = 1,2 / ... (L + Lw). The interlaced level index t takes the values L- (u mod L). When the decoder wraps around the received sequence, ß1, it is used as R, l? - í, and is used as rtIj- ±, when the new ß_ is calculated. Normalize the results so that the elements of each ß. add up to the unit. Again, retain the most recent vectors of L found by this recursion. The next step of this recursion method is the same as the paeo (v) set forth hereinabove with respect to the specific vector method for producing the decisiee indietintae and ealidae of bits decoded by the circular MAP decoder. While the preferred embodiments of the present invention have been shown and described herein, it will be obvious that such modalities are provided by way of example only. It is read many variations, changes and replacements will occur and be skilled in the art and move away from the invention in the present. Accordingly, it is intended that the invention be limited only by the spirit and scope of the appended claims.

Claims

1. A very small aperture terminal communication system for satellite communication, comprising: a plurality of very small aperture terminals, each comprising: a parallel concatenated encoder comprising a plurality of component encoders connected in a parallel concatenation, the parallel concatenated encoder applying a concatenated code parallel to a block of data bits received from a source, and generating component keywords from it, the parallel concatenated encoder comprises a keyword formatter to format the bits of the component key words , to provide a composite keyword; a packet formatter for assembling data packets for transmission, each data packet comprising bits of at least one compueeta keyword; a modulator for receiving the data packets and providing modulated signals therefrom; an up-converter to transfer the modulated signals to a carrier frequency; an interconnection for connecting each respective very small aperture terminal to an antenna for transmitting modulated signals to the satellite, and receiving signals modulated from the satellite; a down converter to transfer each signal received from the carrier frequency to an intermediate frequency; a demodulator for synchronization and demodulation of the received signals; a packet-to-keyword formatter to form composite keywords received from the modulated signals; and a composite decoder comprising a plurality of component decoders for decoding received key words.

2. The communication system of claim 1, wherein the component encoders comprising the parallel concatenated encoder apply convolutional codes to the data bite block.

3. The communication system of claim 2, wherein the parallel concatenated convolutional code comprises repetitive systematic codes. The communication seventh of claim 2, wherein the parallel concatenated convolutional code comprises non-repetitive systematic codes and joined in the queue. The communications system of claim 4, wherein the component decoders comprise MAP decoders. 6. The communication system of claim 1, wherein the modulator comprises an extended spectrum modulator, and the deemodulator comprises a demixing demodulator. The communication system of claim 1, wherein: the parallel concatenated code comprises an internal parallel concatenated code, connected in concatenation in series with an outer code; and the decoder comprises an internal decoder associated with the internal parallel concatenated code, and further comprises an external decoder associated with the external serial concatenated code. The communication seventh of claim 1, wherein the encoder and the decoder comprise a programmable encoder / decoder system, comprising a plurality of encoding options that can be selected by switches. The communications system of claim 8 comprising four coding / decoding options: (1) parallel concatenated coding; (2) an outer code in concatenation in series with an internal parallel concatenated code; (3) serial concatenated encoding comprising an outer encoder and a single component encoder; Y (4) a unique code, such that only one component encoder is used. The communication seventh of claim 8 which also comprises at least one cube terminal; The modulator of each very small aperture terminal comprises an extended-spectrum modulator to apply one of a plurality of extension sequences to each packet of data to be transmitted, the extension sequences being grouped into sets, each set comprising at least an extension sequence, each set of extension sequences being associated with one of the coding options; the cube terminal comprising at least one de-scrambling demodulator for each extension sequence, and a plurality of decoder, said cube terminal, modulating and deepening the signals received from the satellite, which are transmitted in time lapse intervals, and Each one uses one of the coding options and one of the extension sequences associated therewith, the decoders being configured for each received signal based on the extension frequency identified by the de-extention modulator. The communications system of claim 1, which also comprises at least one cube terminal to provide star connectivity. The communication system of claim 1, wherein the parallel concatenated encoder also comprises a drilling function to suppress code bits of the co-ponentee key words, in accordance with a pre-determined drilling pattern, and the composite decoder It includes a function of defoaming to insert neutral values for the bits drilled in the word key component.