KR890000971B1 - Power converting system - Google Patents

Abstract

The system converts electric power between ac and dc and includes a chopping circuit which is connected to the dc output side of a full- wave thyristor bridge circuit. The system is controlled to reduce the output voltage from a maximum to a zero value by first reducing the duty cycle of the chopping circuit until a minimum pulse width is reached and then reducing the phase angle of firing of the thyristors of the bridge circuit. This system is capable of effecting stable control over the entire controlling range.

Description

Power converter

1 to 4 are diagrams illustrating the principle of operation of the present invention, and FIG. 1 is a main circuit diagram of a power converter.

2 is a waveform diagram for explaining the operation of the circuit of FIG.

3 is a characteristic diagram of pulse width and phase angle with respect to the voltage command.

4 is a function generating circuit diagram for obtaining the characteristics of FIG.

5 is an overall configuration diagram showing a specific embodiment of the power conversion apparatus according to the present invention.

6 is a waveform diagram for explaining the operation of FIG.

7 is a control circuit diagram of a DC motor to which the power conversion device of the present invention is applied.

8 is a main circuit diagram of a power conversion device according to another embodiment of the present invention.

9 is an operational waveform diagram of FIG.

10 is a main circuit diagram of a power conversion device according to another embodiment of the present invention.

11 is an operating waveform diagram of FIG.

12 is a main circuit diagram of a power conversion device according to another embodiment of the present invention.

13 is an operating waveform diagram of FIG.

14 is a main circuit diagram of a power conversion device according to another embodiment of the present invention.

The present invention relates to an apparatus for converting electric power between alternating current and direct current using a control rectification circuit, for example, a full-wave bridge circuit.

Background Art Conventionally, a single phase propagation bridge circuit or a three phase propagation bridge circuit has been used for power conversion between single phase alternating current or three phase alternating current and DC. In this way, the DC output voltage of the full-wave bridge circuit is controlled by varying the phase controllable control rectifying means connected to each arm of the bridge circuit, for example, the firing phase angle of the thyristors.

In general, the gate control method of the die Lister has a firing phase angle between the die lister connected to the full-wave bridge plus axis arm and the die lister connected to the minus axis arm, and the phase angle is varied so that the output voltage is zero. Control up to the maximum is well known.

In this method, however, the lower the DC output voltage, the larger the phase delay of the AC current with respect to the AC voltage. Therefore, the invalid messenger increases, the power factor deteriorates, and the ripple component increases, which adversely affects the power supply.

As a method of improving the shortcomings in the region having a small DC output voltage, U.S. Patent No. 4,245,293 has a different firing phase angle between the positive and negative axis die Lister groups in the region having a small output voltage. It is proposed to have a short-circuit mode in both groups. As a result, considerable power factor improvement can be achieved. However, a low frequency current ripple of about 150 Hz is generated in the output current, and further improvement in power factor is desired.

On the other hand, as another way to improve the power factor, IEEE TRANSACTIONS ON INDUSTRY APPLICATION, VOL IA-15 recently titled (A Pulsewidth Controlled AC-to-DC Converter to Improve Power Factor and Waveform of AC Line Current) , NO.6, November / December 1979). In this system, an element having a current interruption function is connected to the positive axis arm of the full-wave bridge circuit. Then, the element is chopped at a predetermined cycle and the chopping pulse width is varied to control the output voltage. According to this method, since the AC voltage and the current are in the same phase, the power factor can be approached to one. In addition, the sixth harmonic component of the power supply is almost not included in the DC output voltage, and the ripple component of the output current can be suppressed small.

However, in order to continuously control the output voltage to a small value by this method, it is necessary to continuously control the chopping pulse width of the device to a very narrow range. However, such control is difficult, and there is a limit to narrowing the chopping pulse width. With the current transistor or GTO (gate turn-off thyristor), the output voltage cannot be controlled to about 1/10 or less of the rated value in relation to the minimum chopping pulse width.

By the way, there are many applications in which control to 1/10 or less of the static voltage is required. For example, in a high speed elevator driven by a DC motor, it is necessary to continuously control up to a speed of 1/1000 or less of the rated speed. In such a case, the above-described method is not applicable, and finally, a method of controlling the firing phase angles of the positive side and the negative side die Lister groups of the propagation thyristor bridge has to be adopted.

A first object of the present invention is to provide an AC-DC power converter having a good power factor and a simple structure capable of optically controlling the output voltage.

It is a second object of the present invention to provide a power exchange apparatus capable of obtaining a stable control function over the entire control range.

A first aspect of the present invention is a power conversion device for converting power between AC direct currents using a control rectification circuit including a phase controllable control rectification means such as a thyristor, comprising: chopping a DC output of the control rectification circuit. And switching means, means for controlling the phase angle of the rectifying circuit in accordance with a desired output voltage, and means for controlling a pulse width for chopping the switching means in accordance with the desired output voltage.

A second feature of the present invention is to control the phase angle of the rectifying means by making the pulse width constant in a range where the desired output voltage is small, and to control the pulse width by making the phase angle constant in a range where the output is large. In one.

Other objects and features will be described in detail in the following embodiments.

First, the operation principle of the present invention will be described with reference to FIGS.

Herein, a case where GTO is employed as a controllable opening and closing means having a current interruption function, taking power exchange between a three-phase alternating current power source and a direct current as an example, is also easily used in the case of using the single-phase alternating current or other switching means described later. I can understand.

1 is an embodiment of the main circuit configuration of the present invention. In the figure, the full-wave bridge is composed of all thyristors U ₁ -W ₂ , and the three-phase AC power supply Ea is connected to the AC terminal of the full-wave bridge, and the DC terminal and the load L are connected to each other. A chopper circuit composed of gate turn-off thyristors GTO ₁ and GTO ₂ is connected therebetween.

A load L including a resistor R _L , an inductance L _L , and a DC voltage E _L by controlling the gates of the thyristors U ₁ -W ₁ , GTO ₁ , and GTO ₂ . Control the voltage applied to

Here, it is assumed that the time constant L _L / R _L of the load L is sufficiently longer than the period of the AC power supply, and the DC voltage E _L is assumed to be slightly lower than the output voltage E _O. . For example, in order to control a DC motor, such conditions are obtained when the armature circuit is set to the load L. FIG.

2 is a waveform diagram illustrating the operation of FIG. 1 and shows waveforms of phase voltages U, V, W, output voltage E _O and U phase current I _U. Here, when the phase angle for controlling the gates of the diesters U ₁ -W ₂ is described as α (U ₁ ), it is represented by the delay angle from the time when the voltage on the UW is inverted from minus to positive, and the following control is performed. Delay angle), (GT O ₁ ) is chopped to the period T, the pulse width P, and (GTO ₂ ) is chopped to the pulse width of (TP).

FIG. 2 (b) shows waveforms when (GTO ₁ ) and (GTO ₂ ) are chopped in the state of control delay angle α = 0 °.

In this case, for example, as shown by the loop ① of FIG. ₁ , when (GTO ₁ ) is fired, current flows through the die Lister U ₁ (V ₂ ), and a load current flows through the power source E _a . Next, when (GTO ₁ ) is interrupted and (GTO ₂ ) is fired, the load current L _L flows through the path indicated by the loop ② and does not pass through the power source E _a . The output voltage E _{0 at} this time becomes zero. Therefore, the voltage waveform of the drawing voltage E _O becomes a waveform of a pulse shape determined by the pulse width P as shown in FIG. Thus, by varying the pulse width P, the output voltage E _O and the U phase current I _U can be controlled.

In this case, the phase delay of the phase current with respect to the output voltage E _O is very small, and the power factor can be approached to one.

However, there is a limit to narrowing the pulse width P, and it is difficult to control the output voltage E _O to about 1/10 or less of the rating.

Therefore, in the present embodiment, the pulse width P at that time is maintained in the controlled state until the pulse width P is minimized, and then the output voltage E _O is zero by controlling the control delay angle α. Control continuously.

FIG. 2C is a waveform in the case where the control delay angle α is approximately 90 °, that is, the output voltage is near zero while the pulse width P is kept to the minimum. That is, the thyristor U ₁ -W _{2 of} FIG. ₁ calls the control delay angle 90 °, and (GTO ₁ ) conducts the thyristor U ₁ -W ₂ determined by the control delay angle 90 °. Chopping at the minimum pulse width (P) during the possible period. Therefore, the reactive power can be made sufficiently small as shown, and the power factor in the low output range can be improved.

The relationship between the phase angle (alpha) and pulse width P with respect to the output voltage mentioned above is shown in FIG. The figure shows the relationship between the pulse puck (S _P) and phase angle _(α S) to the voltage control value (S _V). The voltage command (S _V ) on the horizontal axis represents the ratio to the static voltage, the pulse width (P) and the phase angle (α) on the vertical axis represent the ratio and the control delay angle to the conduction width, respectively. As can be seen from the figure, at 0.1 or more of the rated static pressure, the control delay angle (S _α ) is maintained at 0 ° to vary the pulse width (S5) _N , and at or below 0.1 of the rated voltage, the pulse width (S _P ) is kept to a minimum. By varying the phase angle S _α , it is continuously controlled from zero voltage to rated voltage.

Thus, how to get the pulse width (P) and phase angle (α) according to the voltage command _(V S) are conceivable various, but shows the example in FIG. 4. 4 is constructed using the function generators F _α and F _P having the characteristics shown for the voltage command S _V , and the outputs S _α and S _P are the voltage commands S _{V. V} ) is the characteristic of FIG. 3. Therefore, by setting these (S _P ) and (S _α ) as the pulse width command and the phase angle command, respectively, the gates of the (GTO ₁ ), (GTO ₂ ) and the thyristor (U ₁ -W ₂ ) are controlled. the voltage control according to the command (S _V) is enabled.

로 된다. 이어서, (P)를 0.1에서 1까지 변화시키면 출력전압은 최대

까지 증가한다.That is, if the pulse width P is kept at 0.1 and the control delay angle α is changed from 90 ° to 0 °, the output voltage is set to 0.

It becomes Subsequently, if (P) is changed from 0.1 to 1, the output voltage

To increase.

로 된다. 이어서(P)를 0.1에서 1까지 변화시키면 마이너스 출력전압은 최대

까지 증가한다.When controlling negative output voltage, if α is changed from 90 ° to 180 °, the output voltage is 0

It becomes Then change (P) from 0.1 to 1 and the negative output voltage

To increase.

One specific embodiment of the present invention according to the above-described operation principle will be described with reference to FIGS. 5 and 6 below.

5 is an overall configuration diagram of a power converter according to the present invention, and FIG. 6 is an operation explanatory diagram thereof.

Next, the details will be described. In the drawing, PS is a gate pulse generation circuit of a thyristors, and the three-phase power source E _a has a neutral point between the transformer T _r and the line voltage UW, VU, and ( WV) and input to three-phase phase shifter (PU), (PV), (PW). These phasers PU, PV, and PW are inputted as phase command signals S _α . The phase command signal S _α is a function generator F _α having an output (FIG. 3 characteristic) in which the zero delay time α is zero in a large range of the voltage force signal S _V , that is, a range in which the output voltage is large. Made of) Therefore, the phase shifters PU, PV, and PW generate wide-width pulses GU ₁ -GW _{2 as} shown in FIG. 6 corresponding to the phase command signal S _α , and these purses GU ₁ -GW. ₂ ) is applied to the gate of the thyristors (U ₁ -W ₂ ). Here, the positive half wave of the power supply E _a is controlled from the pin numbers 1 of the phase shifters PU, PV, and PW, and the negative half wave of the power supply E _a is controlled from the pin number 2. Generates a gate pulse.

The pulse width command signal S _P is input to the GTO gate pulse generation circuit PP. The GTO pulse width command signal S _P is a signal in which the voltage command signal S _V is constant in the range where the voltage command signal S _V is small, that is, the output voltage is small, in which the voltage command signal S _V is large. In the range of large voltage, it is obtained by the function generator F _P which generates the signal (the characteristic of FIG. 3) which becomes large pulse width.

Accordingly, in the gate pulse generation circuit PP of the GTO, pulses GT ₁ and GT ₂ , as shown in FIG. 6 (b) corresponding to the pulse width command signal _{SP P} , are generated and (GTO ₁ ). Is applied to the gate of (GTO ₂ ).

These pulses _{(GTO 1), (GTO 2} ) by the period of (P) is the period (GTO ₁₎ of the (GTO ₁₎ by conduction and whole non-a (GTO _2), and (TP) non-trough, (GTO ₂ ) shall be conduction.

7 shows an application example in the case of controlling the speed of a DC motor using the power converter according to the present invention described above.

In the figure, (M) is an armature of a direct current motor, (F) is a field, (TH) is a push-pull die amplifier amplifier for controlling a field current, and (FPS) is an ideal phase.

(PS) is a gate control circuit of the thyristors (U ₁ -W ₂ ) shown in FIG. 5, and (PA) compares the armature current command (S _C ) and the current transformer (CT) for detecting the armature current. Is a comparator.

Armature current has become constant, the armature current control according to the command (S _C). The speed detected by the speed command S _S and the speed generator P _G is compared with the abnormal phase FPS, and is controlled to comply with the speed command by controlling the field current positive and negative.

In such a control system, it is possible to control the armature current in only one direction, thereby enabling the four-phase operation of forward / reverse, reverse and regeneration of the motor.

When the power failure occurs in the regenerative state, the output voltage of the die Lister circuit becomes zero, so that the armature current increases rapidly and the die Lister may be destroyed. For this reason, the DC contactor is normally inserted in the armature circuit. However, in this embodiment, since the chopper circuit consisting of (GTO ₁ ) and (GTO ₂ ) is connected to the direct current terminal of the radio wave bridge and the transmission period, if the armature current exceeds a certain level (GTO ₁ ), (GTO ₂₎ By giving the off-pulse gate signal to turn off), it is possible to break the armature current. Therefore, a direct current contactor becomes unnecessary, and the device can be miniaturized and cost can be reduced.

According to this embodiment of the present invention, the power factor of the three-phase AC power conversion and the reverse conversion is improved, the power supply capacity can be reduced, and the DC contactor is not required.

8 is another embodiment of the present invention, and FIG. 9 is an operation waveform diagram thereof.

In the above embodiment, in order to enable inverse conversion in addition to the conversion from AC to DC, the full-wave bridges are all configured using a thyristor.

However, if such reverse conversion is not performed, as shown in FIG. 8, the thyristors U ₁ , V ₁ and W ₁ on the positive side of the radio bridge and diodes D ₁ and D ₂ on the negative side And (D ₃ ), and one (GTO) is connected between the forward current terminal of the full wave bridge and the load, and a diode (D ₄ ) for refluxing the load current in parallel with the load may be connected.

In this case, the voltage E _O between the DC terminals when controlling the control delay angle α of the thyristors U ₁ , V ₁ , and W ₁ is represented by the solid line in FIG. It becomes the voltage between. Then, a waveform of the output voltage (E _O) when maintaining a constant control delay angle (α), and controls the pulse width (P) for opening and closing (GTO), as in the preceding embodiment in the figure (b) . In this case, the case where the number of chopping times is set to five is shown. In this way, even when only forward conversion is performed, the present invention is effective. By controlling the phase angle α and the pulse width P, a wide range of power conversion is possible with high power factor.

Although only the power conversion between the three-phase alternating current and the direct current has been described above, of course, the present invention can also be applied to the power conversion of other multi-phase alternating current or single-phase alternating current.

FIG. 10 is an embodiment in which the linear force conversion between single-phase alternating current and direct current is performed, and FIG.

As shown in FIG. 10, a chopper circuit composed of (GTO ₁ ) and (GTO ₂ ) is connected between the DC terminal of the full-wave bridge circuit composed of the thyristors R ₁ -S ₂ and the load L.

The output voltage E _{O in} the case of controlling only the phase angle α of the diesters R ₁ -S _{2 is} shown in FIG. 11 (α), and in addition to the phase angle α (GTO ₁ ), The output voltage E _O when (GTO ₂ ) is opened and closed at the pulse width P is shown in this figure (b). This figure shows waveforms when the control delay angle α is set to 90 ° and the output voltage E _O is set to zero. Similarly to the above embodiment, when the pulse width P is minimized, the power factor when the output voltage E ₀ is zero can be remarkably improved. Similarly, if the control delay angle α is controlled from 0 to 180 ° and the pulse width P is controlled from maximum to minimum, it can be controlled over the entire range from forward to inverse transformation.

In the above embodiment, the fundamental power factor can be improved. However, as shown in FIG. 2, an interrupted current flows in the AC power, so that high frequency is generated.

However, the high frequency can be reduced by inserting a filter circuit between the radio wave bridge and the chopper circuit. An embodiment thereof is shown in FIG. 12, and an operation waveform diagram is shown in FIG.

In the drawing, (L _D ) is a reactor, (C _D ) is a condenser, and these constitute a filter circuit, and it is assumed that the current L _LD flowing through the reactor L _D is selected as a constant that is continuous. The output voltage E _O of the full-wave bridge when the control delay angle α of the listers U _1- W ₂ is set to 0 becomes the waveform of FIG. 13 (b).

Then, while maintaining the control delay angle α = 0, (GTO ₁ ) is opened and closed as shown in FIG. 13 (g) in the pulse width (P) and (GTO ₂ ) in the pulse width of (TP) as in the above embodiment. The terminal voltage (E _CD ) of the capacitor (C _C ), the current (L _LD ) flowing through the reactor (L _D ), and the current ( _U ) of the U phase when controlled are as shown in Figs. 13 (c)-(f). .

In this way, not only can the power supply waveform be improved, but also the output current I _LD of the full-wave bridge is continuous, so that the thyristor u ₁ -w ₂ can be narrowed by the gate pulse rudder, so that the gate pulse There is also an effect that the generation circuit is simplified.

In the above embodiment, the case where GTO is used as an opening / closing means having a current blocking function has been described as an example. However, as shown in FIG. 14, the same effect can be obtained even if the transistor is replaced with the known transistor Q ₁ (Q ₂ ). Of course.

As described above, according to the present invention, power conversion can be performed with good power factor over a wide range.

Claims (10)

A power converter for converting power between AC direct currents using a control rectification circuit (U ₁ -W ₂ ) including a control rectification means capable of controlling phase, wherein the DC output of the control rectification circuit (U ₁ -W ₂ ) is controlled. phase of the chopping switching means (GTO _1, GTO or ₂ Q _1, Q ₂₎ and a desired output voltage of the rectifier circuit (U ₁ -W ₂₎ according to _(V S) to means for controlling the angle _(α S) ( F _α, PS) and the desired output voltage (S _V) in accordance means for controlling said switching means (GTO _1, GTO ₂ or Q _1, the pulse width of the chopping the _{Q 2) (S P) (} F α, A power conversion device comprising PP). The pulse width (S _P ) according to claim 1, wherein the phase angle (S _α ) is varied when the desired output voltage (S _V ) is less than or equal to a predetermined value, and when the desired output voltage (S _V ) is greater than or equal to a predetermined value. Power converter configured to vary. The method of claim 2, wherein the phase angle control means (F _α, PS) is to vary the said phase angle (S _α) according to the desired voltage (S _V) when less than the predetermined value, the desired voltage (S _V), It said desired voltage _(V S) is the predetermined value or more when the electric power conversion system configured to hold the phase angle _(α S) to the minimum (the control delay of each case). 3. The pulse width controlling means (F _α , PP) according to claim 2, wherein the pulse width control means (F _α , PP) maintains the pulse width (S _P ) to a minimum when the desired voltage (S _V ) is less than a predetermined value, and the desired voltage (S _V ). The power converter configured to vary the pulse width (S _P ) in accordance with the desired voltage (S _V ) when the value is equal to or greater than the predetermined value. The electric power according to claim 1, wherein a filter circuit (L _D , C _D ) is connected between the control rectifier circuit (U ₁ -W ₂ ) and the switching means (GTO ₁ , GTO ₂ or Q ₁ , Q ₂ ). Inverter. The switching means for chopping the output of the control rectifier circuits U ₁ -W ₂ is an opening and closing means (GTO ₁ or Q ₁ and GTO ₂ or Q ₂ ) having a current blocking function. And a switching device connected between the control rectifier circuit (U ₁ -W ₂ ) and the load (L) and in reverse parallel with the load (L). The method according to claim 6, wherein when the first switching means (GTO ₁ or Q ₁ ) connected between the control rectifier circuit (U ₁ -W ₂ ) and the load (L) is connected in reverse parallel with the load (L) Blocking the _second opening and closing means (GTO ₂ or Q ₂ ), and when blocking the first opening and closing means (GTO ₁ or Q ₁ ), so as to conduct the _second opening and closing means (GTO ₂ or Q ₂ ). Configured power converter. The half-wave rectifying element according to claim 1, wherein a load (L) is connected to an output terminal of the opening / closing means (GTO ₁ , GTO ₂ or Q _1, Q ₂ ), and is antiparallel to the load (L). A power converter formed by connecting (D ₄ ). The power conversion device according to claim 1, wherein the control rectification circuit comprises a full-wave rectification bridge circuit (U ₁ -W ₂ ). The power converter according to claim 1, wherein the desired output voltage (S _V ) is an output voltage command of the power converter.

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