KR20240071282A - Estimating method for rotor time constant of motor using deep slot effect - Google Patents

Abstract

The present invention relates to a method for estimating the rotor time constant of an electric motor, which includes a) detecting the stator current and magnetic flux of an electric motor driven by a speed controller, and b) calculating the rotor time constant using the relationship between the stator current and magnetic flux in an arithmetic device. It may include estimating the electronic time constant, and c) compensating the estimated rotor time constant by applying an error between the actual stator magnetic flux and the detected stator magnetic flux in the computing device.

Description

{Estimating method for rotor time constant of motor using deep slot effect}

The present invention relates to a method for estimating the rotor time constant of an electric motor using the deep sphere effect, and more specifically, to a method for estimating the rotor time constant of an electric motor without a position or speed sensor.

In general, the time constant of the motor rotor is an essential parameter for high-performance control of induction motors, and is used to calculate slip angular velocity in indirect magnetic flux control.

Error in rotor time constant causes slip error and may deteriorate control performance.

Additionally, the rotor time constant is used as a parameter of an adaptive full-order observer. If there is an error in the rotor time constant during sensorless operation, a speed error is caused.

As such, the rotor time constant must be accurately estimated, and in the past, relatively complex devices and methods are used to estimate the rotor time constant of an electric motor.

For example, in Publication Patent No. 10-2004-0084083 (rotor time constant estimation device for induction motor, published on October 6, 2004) of the applicant of the present invention, the synchronous flux voltage command value and torque voltage command value are calculated at the angles in the stationary coordinate. Convert it to a command value, obtain the reference magnetization current and variable magnetization current using the output of the 3-phase/2-phase converter and the parameters of the induction motor, and then estimate the rotor time constant that matches the variable magnetization current to the reference magnetization current. The device is described.

In this way, the conventional method had the disadvantage of using a relatively complex device, requiring a large number of parameters during the calculation process, and the calculation step was also relatively complex.

The problem to be solved by the present invention in consideration of the problems of the prior art as described above is to provide a method of estimating an accurate rotor time constant using a relatively simple calculation using the relationship between stator magnetic flux and stator current. .

The method for estimating the rotor time constant of the electric motor of the present invention to solve the above technical problems includes a) detecting the stator current and magnetic flux of the motor driven by the speed controller, and b) the stator current and magnetic flux in the calculation device. It may include a step of estimating the rotor time constant using the relationship, and c) a step of compensating the estimated rotor time constant by applying the error between the actual stator magnetic flux and the detected stator flux in the calculation device.

In an embodiment of the present invention, the speed of the electric motor may be controlled by current/frequency (I/f) control of the speed controller.

In an embodiment of the present invention, step b) includes b-1) obtaining the relationship between the stator magnetic flux and stator current, and b-2) determining the stator current in a specific axis, assuming that the stator current exists only in one specific axis. It may include a process of finding the size of the magnetic flux, and b-3) a process of estimating the rotor time constant by finding the relationship between the size of the stator magnetic flux and the rotor time constant.

In an embodiment of the present invention, step b-3) determines the rotor time constant in the rotating state from the relationship between the size of the stator magnetic flux and the rotor time constant, and determines the rotor inductance detectable in no-load state operation and the rotating state. The rotor time constant can be estimated using the rotor time constant.

In an embodiment of the present invention, step c) may compensate for the magnitude reduction and phase delay of the actual stator magnetic flux and the detected stator magnetic flux, thereby compensating the estimated rotor time constant.

In an embodiment of the present invention, step c) includes c-1) converting the current command to a stationary coordinate system using a rotating transform, and c-2) phase delay and magnitude of the detected stator magnetic flux. It may include a process of calculating the degree of reduction, c-3) a process of setting the observation frequency, and c-4) a process of compensating the rotor time constant using the stator magnetic flux and stator current compensated at the observation frequency.

The present invention has the effect of estimating the rotor time constant through a simple calculation by estimating the rotor time constant using the relationship between the stator magnetic flux and the stator current.

Additionally, the present invention has the effect of being able to estimate a more accurate rotor time constant by correcting the estimated time constant by considering the relationship between the detected magnetic flux and the actual magnetic flux.

Accurate estimation of the rotor time constant can improve the reliability of motor control using the time constant and improve control performance.

1 is a flowchart of a method for estimating the rotor time constant of an electric motor according to a preferred embodiment of the present invention.
Figure 2 is an example hardware configuration for implementing the method of Figure 1.
Figure 3 is a detailed flowchart of step S30 in Figure 1.
Figure 4 is an example algorithm for calculating the k value.
Figure 5 is a detailed flowchart of step S40 in Figure 1.
6 is an exemplary diagram of a magnetic flux detector.
Figure 7 is a Bode diagram showing the difference between estimated magnetic flux and actual magnetic flux.
Figure 8 is a waveform for explaining the compensation method.

In order to fully understand the configuration and effects of the present invention, preferred embodiments of the present invention will be described with reference to the attached drawings. However, the present invention is not limited to the embodiments disclosed below, but can be implemented in various forms and various changes can be made. However, the description of this embodiment is provided to ensure that the disclosure of the present invention is complete and to fully inform those skilled in the art of the present invention of the scope of the invention. In the attached drawings, components are shown enlarged in size for convenience of explanation, and the proportions of each component may be exaggerated or reduced.

Terms such as 'first' and 'second' may be used to describe various components, but the components should not be limited by the above terms. The above terms may be used only for the purpose of distinguishing one component from another. For example, a 'first component' may be named a 'second component' without departing from the scope of the present invention, and similarly, a 'second component' may also be named a 'first component'. You can. Additionally, singular expressions include plural expressions, unless the context clearly dictates otherwise. Unless otherwise defined, terms used in the embodiments of the present invention may be interpreted as meanings commonly known to those skilled in the art.

Hereinafter, a method for estimating the rotor time constant of an electric motor according to an embodiment of the present invention will be described in detail with reference to the drawings.

Figure 1 is a flow chart of a method for estimating the rotor time constant of an electric motor according to a preferred embodiment of the present invention, and Figure 2 is an example hardware diagram for implementing the method of Figure 1.

Referring to FIGS. 1 and 2, the present invention includes a speed controller 20 that controls the speed of a time constant driving target electric motor 10 by current/frequency (I/f) control, and a speed controller 20 of the electric motor 10. A current detector 30 that detects the stator current, a magnetic flux detector 40 that detects the magnetic flux of the speed controller 20, and a stator current detected by the current detector 30 and a magnetic flux detected by the detector 40. It can be implemented by a calculation device 50 that calculates the stator magnetic flux according to a given calculation equation to estimate the rotor time constant and compensate.

The present invention includes a step of controlling the speed of the electric motor 10 (S10), a step of detecting the stator current and magnetic flux of the electric motor 10 (S20), and estimating the rotor time constant using the relationship between the stator current and magnetic flux. It may include a step (S30) and a step (S40) of correcting the estimated rotor time constant.

Hereinafter, the specific structure and operation of the present invention configured as described above will be described in detail.

First, the electric motor 10 is an induction motor, and since it is difficult to measure the physical quantity of the rotor, the present invention uses a method of estimating the physical quantity of the rotor by detecting the physical quantity of the stator.

In the present invention, the stator magnetic flux is detected and used for calculation.

Self-commissioning of the rotor time constant of the electric motor 10 has been performed in a standstill state or a locked state, but since the magnitude of the voltage is small in the standstill state, nonlinearity of the inverter 21 It is greatly affected, and its application is limited because the locked state requires a separate braking device.

Therefore, the present invention estimates the rotor time constant in the rotation state.

In order to put the electric motor 10 in a rotating state, the speed controller 20 that controls the speed of the electric motor 10 through I/f control is used, as described above.

First, the speed of the electric motor 10 is controlled as in step S10, and the electric motor 10 is driven.

Next, the stator current and magnetic flux of the electric motor 10 are detected as in step S20.

The reason for detecting stator current and magnetic flux is that it is difficult to measure physical quantities such as rotor current.

Next, as in step S30, the rotor time constant is estimated using the relationship between stator current and magnetic flux.

Figure 3 is a detailed flowchart of step S30.

Referring to FIG. 3, the relationship between the stator voltage and the stator magnetic flux is obtained using the stator voltage and stator magnetic flux equation as in step S31.

The stator voltage equation and magnetic flux equation of an induction motor in an arbitrary synchronous reference frame ω are expressed by Equations 1 and 2 below, respectively.

In the above equations, v represents the stator voltage, R represents the stator resistance, and λ represents the magnetic flux.

In the case of subscripts, s refers to the stator, r refers to the rotor, and d and q represent the d-axis and q-axis, respectively. The superscript ω is an arbitrary synchronous coordinate system.

Additionally, the stator flux is expressed by Equation 2, which includes stator current and rotor flux.

The stator current (i) is commanded as a DC biased sinusoidal current during I/f control.

Here, the leakage coefficient (σ) is 1-(L ² m/LsLr), so Equation 1 above can be solved as Equation 3 below.

The rotor current equation is as Equation 4 below.

In the case of a squirrel cage induction motor, v ^ω _dqr is 0, so by organizing Equation 3 and Equation 4, Equation 5 below can be obtained.

Equation 6 can be expressed from Equation 5.

If Equation 6 above is simplified and converted to an equation for stator magnetic flux, it can be expressed as Equation 7 below.

In Equation 7 above, A and B are defined by the following Equation 8 and Equation 9, respectively.

In this way, the magnetic flux of the stator can be expressed as stator current. At this time, the stator current is a value detected through the current detector 30.

Next, estimate the parameters as in step S32.

When controlling the I/f speed using the speed controller 20, the current component corresponding to the magnetizing flux and the current component resolving the torque cannot be separated, and therefore, in the present invention, all stator currents exist in the d-axis. Assume.

Assuming that all stator currents exist in the d-axis, the stator current reference coordinate is applied.

At this time, the stator current reference coordinates can be expressed as i ^m _ds =i _s and i ^m _ds =0.

The superscript “m” indicates the current reference coordinate of the stator.

As a result, if the stator magnetic flux component is separated from Equation 7 above, λ ^m _ds =(A)i _s , It can be expressed as λ ^m _qs =(-B)i _s .

At this time, assuming that the occurrence of slip is very small in the no-load state, ω-ωr is 0. Therefore, the stator magnetic flux considering only the d-axis current can be expressed as Equation 10 below.

Equation 10 can be converted to Equation 11 below.

It can be seen that the left term in Equation 11 above shows the response to the stator current in the form of a first-order low-pass filter (LPF), and the cutoff frequency is the reciprocal of the rotor time constant.

Additionally, the stator current is sinusoidal, as previously mentioned, and is specifically commanded in the form of a DC biased sinusoidal wave. The stator current command is expressed in Equation 12.

Additionally, the rotor time constant can be obtained from the detected stator flux.

The stator magnetic flux can be detected using the magnetic flux detector 40, which is a frequency adaptive magnetic flux observer.

Figure 4 is an example diagram of an algorithm for calculating the k value.

At AC frequency, the magnitude of magnetic flux (magnitude of magnetic flux response) k corresponds to the magnitude of the Bode diagram, and because the test frequency is very low, it can be estimated more quickly using an averaging filter (AF) than using a low-pass filter. there is.

k is defined as Equation 13 below.

Next, the rotor time constant is finally estimated as in step S33. That is, by obtaining the relationship between the detected k value (detected value of the magnetic flux detector 40) and the rotor time constant, Equation 14 and Equation 15 below can be derived.

Equation 14 above is the rotor time constant calculated from the dynamic inductance of the electric motor 10. That is, the subscript dyn represents dynamic inductance.

Since the rotor time constant must be calculated from the apparent inductance, it can be obtained as Equation 15.

Therefore, in the present invention, the rotor time constant is first estimated from the relationship between the stator current and magnetic flux, and the final rotor time constant is estimated using the estimated rotor time constant and dynamic inductance.

Next, the obtained rotor time constant is corrected as in step S40.

The rotor time constant estimated by the computing device 50 through step S30 is different from the rotor time constant of the actual electric motor.

In the present invention, the error of the estimated rotor time constant with respect to the actual rotor time constant is compensated.

Figure 5 is a detailed flowchart of step S40.

First, in step S41, the stationary coordinate system stator current command is analyzed.

The current command expressed in Equation 12 above is expressed in a stationary coordinate system using a rotating transform.

In Equation 16 above, the frequency component ω ₊ is the sum of ω _IF and ω _test , and ω _- is the difference between ω _IF and ω _test .

Next, as in step S42, the phase delay and magnitude reduction of the detected stator flux are obtained.

Figure 6 is an exemplary configuration diagram of the magnetic flux detector 40.

Referring to FIG. 6, the magnetic flux detector 40 includes a first adder (A1), a first integrator (I1), a second adder (A2), a gain (N1), a third adder (A3), a gain (G1), and a first adder (A1). It may include a second integrator (I2), a third integrator (I3), a gain (G2), a third adder (A3), a gain (N2), and a gain (G3).

The magnetic flux detector 40 uses a frequency adaptive magnetic flux observer, and the input is the stator voltage. In particular, it is assumed to be a stator voltage that eliminates the influence of the stator resistance component. Disturbance fed back from the stator voltage through the first adder (A1) ( ) and input it into the first integrator (I1).

In the above configuration, the stator voltage is input and integrated to obtain the estimated stator magnetic flux ( ) was obtained and the final estimated ω frequency component of the stator magnetic flux ( ) is output.

Magnetic flux error ( ), the frequency component of the stator flux is fed back and the difference with the estimated stator flux value is obtained using the first adder (A1).

Equation 17 and Equation 18 below are the input-output relational expressions of the magnetic flux detector 40.

Equation 17 represents the transfer function of the stator magnetic flux, which is the output, to the stator voltage, which is the input.

Equation 18 refers to the disturbance to the stator voltage.

In Equations 17 and 18 above, the frequency (ω) is the frequency assuming that the I/f driving frequency and observation frequency are the same.

The magnitude and phase of the ω _IF +ω _test transfer function are defined by Equation 19 below.

In the frequency transfer function, the reduction in phase magnitude (mag ₊ ) and phase delay (phase ₊ ) can be defined as Equation 19 below.

Here, α is 2ζ·ω _IF ·ω ₊ , and β is 2ζ·ω _test ·ω ₊ .

In the same way, the magnitude and phase of the ω _IF -ω _test transfer function are expressed in Equation 21 below.

At this time, the magnitude (mag _- ) and phase (phase _- ) of the ω _IF -ω _test transfer function are defined by Equation 22 below.

Here, γ is 2ζ·ω _IF ·ω _- , and δ is 2ζ·ω _test ·ω _- .

As such, the two transfer functions expressed in Equation 19 and Equation 21 have differences in the size of the phase and degree of delay, and assuming that the current commands for the two frequency components are the same, the size of the actual magnetic flux is expressed in Equation 23 below. can be expressed.

Assuming that there is a current only in the d-axis as previously assumed in Equation 23 above, the d-axis magnetic flux estimated in the stator current coordinate system is expressed as Equation 24 below.

Here, when estimating the rotor time constant, only the ω _test component of the estimated magnetic flux is extracted using a band-pass filter. It can be seen that the alternating current (AC) component extracted in this way is reduced in size and delayed in phase.

Equation 25 below is the band-pass filtering result of Equation 24 above.

In Equation 25 above, the variables M and θ can be defined as Equation 26 and Equation 27 below, respectively.

Next, set the observation frequency as in step S43.

Assuming that the magnetic flux response at the two frequencies is the same size and has opposite phases, the magnetic flux size and phase delay of the alternating current component are simply defined by Equation 28 below.

Figure 7 is a Bode diagram showing the difference between estimated magnetic flux and actual magnetic flux.

Referring to FIG. 7, it is easy to obtain an observation frequency that satisfies the conditions that the phases of the two frequency responses are the same and the signs are opposite.

The observation frequency is selected as shown in Equation 29.

The d-axis magnetic flux estimated from the stator current reference coordinate system is shown in Equation 31 below, and only the alternating current component can be extracted using a band-pass filter. The extracted ω _test component is expressed in Equation 32.

Next, the phase and magnitude are compensated as in step S44.

By compensating for the decrease in the size of the estimated magnetic flux as shown in Equation 33, the accurate rotor time constant can be estimated, and thus control performance can be improved.

This can be expressed as Equation 33 below.

When compensating for phase, a signal with the same phase as the actual magnetic flux must be generated.

Figure 8 is a waveform diagram for explaining the compensation method.

Referring to FIG. 8, the rotor time constant can be estimated using the relationship between actual magnetic flux and current.

In Figure 8, the solid lines represent the actual stator flux and stator current, and the dotted lines represent the compensated magnetic flux and current.

Under the assumption that the compensated magnetic flux and the compensated current are accurately compensated, the relationship between the actual magnetic flux and the current is the same as the relationship between the compensated magnetic flux and the compensated current.

Therefore, if the rotor time constant is estimated using the compensated magnetic flux and current, it becomes the same value as the time constant estimated using the actual magnetic flux and current.

The delay signal described above is generated through an all-pass filter (APF). If you check this with the mathematical equation, it is as shown in Equation 34, Equation 35, and Equation 36 below.

Equation 35 is the phase at the test frequency (ω _test ).

Equation 36 represents the cut off frequency of the all-pass filter.

Through this process, in the present invention, the rotor time constant can be easily estimated by using a method of estimating the rotor time constant using the relationship between the stator magnetic flux and current.

Additionally, a more accurate rotor time constant can be estimated by checking the error between the actual magnetic flux and the measured magnetic flux and compensating by applying the error to the estimated rotor time constant.

Although embodiments according to the present invention have been described above, they are merely illustrative, and those skilled in the art will understand that various modifications and equivalent scope of embodiments are possible therefrom. Therefore, the true technical protection scope of the present invention should be determined by the following claims.

10: electric motor 20: speed controller
30: Current detector 40: Magnetic flux detector
50: computing device

Claims (6)

a) detecting the stator current and magnetic flux of the electric motor driven by the speed controller;
b) estimating the rotor time constant using the relationship between stator current and magnetic flux in an arithmetic device; and
c) A method of estimating the rotor time constant of an electric motor including the step of compensating the estimated rotor time constant by applying the error between the actual stator magnetic flux and the detected stator flux in the calculation device. According to paragraph 1,
A method of estimating a rotor time constant of an electric motor, characterized in that the speed of the electric motor is controlled by current/frequency (I/f) control of the speed controller. According to claim 1 or 2,
In step b),
b-1) Process of calculating the relationship between stator magnetic flux and stator current;
b-2) Process of finding the magnitude of stator magnetic flux in a specific axis, assuming that the stator current exists only in one specific axis; and
b-3) A method of estimating the rotor time constant of an electric motor that includes the process of estimating the rotor time constant by finding the relationship between the size of the stator magnetic flux and the rotor time constant. According to paragraph 3,
In step b-3),
The rotor time constant in the rotating state is obtained from the relationship between the size of the stator magnetic flux and the rotor time constant, and the rotor time constant is estimated using the rotor inductance that can be detected in no-load operation and the rotor time constant in the rotating state. A method for estimating the rotor time constant of a characterized electric motor. According to paragraph 3,
In step c),
A method of estimating the rotor time constant of an electric motor, characterized in that the estimated rotor time constant is compensated by compensating for the reduction in size and phase delay of the actual stator flux and the detected stator flux. According to clause 5,
In step c),
c-1) Process of converting the current command into a stationary coordinate system using rotating transform;
c-2) Process of determining the degree of phase delay and size reduction of the detected stator magnetic flux;
c-3) Process of setting observation frequency;
c-4) A method of estimating the rotor time constant of an electric motor including the process of compensating the rotor time constant using the stator magnetic flux and stator current compensated at the observed frequency.

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