KR101530543B1 - Induction motor and controlling apparatus for induction motor - Google Patents

Abstract

Disclosed is a control device for controlling the operation of an induction motor. The control device for controlling the operation of the induction motor according to one embodiment of the present invention includes a dq coordinate converting unit which converts a current and a voltage applied to the induction motor into dq coordinates, a synchronous dq coordinate entire state/speed measuring unit which calculates estimated rotor magnetic flux and an estimated angular velocity by using the current and the voltage converted into the dq coordinates, a slip angular velocity calculating unit which calculates the slip angular velocity by using the estimated rotor magnetic flux which is calculated, and a phase angle calculating unit which calculates a phase angle by integrating the error of the estimated angular velocity and the slip angular speed which is calculated. The present invention provides the induction motor and the control device for controlling the operation of the induction motor using an indirect vector control method without a velocity sensor.

Description

TECHNICAL FIELD [0001] The present invention relates to a control device for controlling the operation of an induction motor and an induction motor,

The present invention relates to an induction motor, and more particularly, to a control apparatus and an induction motor for controlling the operation of an induction motor using a vector controller.

A technique of vector-controlling an induction motor using an inverter is a technique in which a torque component current that is orthogonal to a secondary magnetic flux in an induction motor is manipulated by separately operating an inverter output voltage, (Instantaneous) at high speed.

Since the rotational angular velocity of the motor is not known, the direct vector controller is applied by using the fixed dq coordinate system in the conventional variable speed drive of the induction motor without the speed sensor.

The conventional method for variable speed drive of induction motor without velocity sensor is as follows: 1) MRAS (Model Reference Adaptive System) method, 2) speed calculation method by estimating stator flux, 3) Luenberger type all state observer as variable model And a technique for estimating the rotor flux and the speed.

The above methods apply direct vector controller using fixed dq coordinate system instead of synchronous dq coordinate system. However, since the torque and speed control of a general motor with a speed sensor is based on an indirect vector control, it is difficult to apply a direct vector control without a speed sensor. In addition, when the fixed dq coordinate system observer is used, there is a problem that the observation error is large in the high-speed region as the variables to be observed have the AC value, and the performance of the torque and speed control is greatly degraded.

It is an object of the present invention to provide a control device and an induction motor for controlling the operation of an induction motor using an indirect vector control method without a speed sensor. Specifically, we design the Luenberger type flux observer in the dq coordinate system rotating at the synchronous speed and provide the basic rule for selecting the gain of the speed adaptation rule.

) 및 추정 각속도(

)를 계산하는 동기 dq 좌표계 전상태/속도 관측부와, 상기 계산된 추정 회전자 자속(

)을 이용하여 슬립 각속도(ω_sl )를 계산하는 슬립각속도 계산부와, 상기 계산된 슬립각속도(ω _sl ) 및 상기 추정 각속도(

)의 오차를 적분하여 위상각(θ_e )을 구하는 위상각 계산부를 포함한다. According to another aspect of the present invention, there is provided a control apparatus for controlling an induction motor, the control apparatus comprising: A dq coordinate converter for converting the estimated rotor flux into a dq coordinate system using the current and voltage converted into the dq coordinate system,

) And the estimated angular velocity (

A synchronous dq coordinate system total state / velocity observing section for calculating the estimated rotor flux

A slip angular velocity calculator for calculating a slip angular velocity ome _sl by using the calculated slip angular velocity omega _sl and the estimated angular velocity ome _sl ,

) To obtain a phase angle ( [ _epsilon ] _e ).

The control device further includes an abc coordinate converter for converting the command voltage into the abc coordinate system using the phase angle ? _E , and a PWM inverter for converting the command voltage converted into the abc coordinate system into ac .

)의 오차를 이용하여 d 축 지령 전류(i_ds ^* )를 생성하는 자속 제어기와, 지령 각속도(ω_r ^*)와 상기 추정 각속도(

)의 오차를 이용하여 q 축 지령 전류(i_qs ^* )를 생성하는 속도 제어기를 더 포함할 수 있다.Further, the control device may be configured to calculate the difference between the command rotor flux ? _Dr ^* and the estimated rotor flux?

), A magnetic flux controller for generating a d-axis command current ( i _ds ^* ) by using the error of the command angular velocity ( _r ^* ) and the estimated angular velocity

) To generate the q-axis command current ( i _qs ^* ) using the error of the q-axis command current ( i _qs ^* ).

) 및 상기 q축 지령 전류(i_qs ^* )를 이용하여 상기 슬립 각속도(ω_sl )를 계산할 수 있다.At this time, the slip angular velocity calculator calculates the estimated rotor magnetic flux (

) And the q-axis command current ( i _qs ^* ) can be used to calculate the slip angular velocity ? _Sl .

The controller further includes a current controller for generating a d-axis command voltage ( v _ds ^* ) by using an error between the d-axis command current ( i _ds ^* ) and the d-axis current converted into the dq coordinate system, And a q-axis command voltage ( _{q q s} ^* ) using an error of the q-axis current converted into the dq coordinate system and the axis command current ( i _{q s} ^* ).

The control device may be driven by an indirect vector control method.

In addition, the induction motor may not include a speed sensing sensor.

) 및 추정 각속도(

)를 계산하는 동기 dq 좌표계 전상태/속도 관측부와, 상기 계산된 추정 회전자 자속(

)을 이용하여 슬립 각속도(ω_sl )를 계산하는 슬립각속도 계산부와, 상기 계산된 슬립각속도(ω _sl ) 및 상기 추정 각속도(

)의 오차를 적분하여 위상각(θ_e )을 구하는 위상각 계산부를 포함한다.According to another aspect of the present invention, there is provided an induction motor including an induction motor and a control device for controlling the induction motor, A dq coordinate converter for converting a current and a voltage applied to the dq coordinate system into a dq coordinate system,

) And the estimated angular velocity (

A synchronous dq coordinate system total state / velocity observing section for calculating the estimated rotor flux

A slip angular velocity calculator for calculating a slip angular velocity ome _sl by using the calculated slip angular velocity omega _sl and the estimated angular velocity ome _sl ,

) To obtain a phase angle ( [ _epsilon ] _e ).

The control device further includes an abc coordinate converter for converting the command voltage into the abc coordinate system using the phase angle ? _E , and a PWM inverter for converting the command voltage converted into the abc coordinate system into ac .

)의 오차를 이용하여 d 축 지령 전류(i_ds ^* )를 생성하는 자속 제어기와, 지령 각속도(ω_r ^*)와 상기 추정 각속도(

)의 오차를 이용하여 q 축 지령 전류(i_qs ^* )를 생성하는 속도 제어기를 더 포함할 수 있다.Further, the control device may be configured to calculate the difference between the command rotor flux ? _Dr ^* and the estimated rotor flux?

), A magnetic flux controller for generating a d-axis command current ( i _ds ^* ) by using the error of the command angular velocity ( _r ^* ) and the estimated angular velocity

) To generate the q-axis command current ( i _qs ^* ) using the error of the q-axis command current ( i _qs ^* ).

The control device may be driven by an indirect vector control method.

In addition, the induction motor may not include a speed sensing sensor.

In addition, the induction motor may not include a speed sensing sensor.

) 및 상기 q축 지령 전류(i_qs ^* )를 이용하여 상기 슬립 각속도(ω_sl )를 계산할 수 있다.At this time, the slip angular velocity calculator calculates the estimated rotor magnetic flux (

) And the q-axis command current ( i _qs ^* ) can be used to calculate the slip angular velocity ? _Sl .

The controller further includes a current controller for generating a d-axis command voltage ( v _ds ^* ) by using an error between the d-axis command current ( i _ds ^* ) and the d-axis current converted into the dq coordinate system, And a q-axis command voltage ( _{q q s} ^* ) using an error of the q-axis current converted into the dq coordinate system and the axis command current ( i _{q s} ^* ).

According to various embodiments of the present invention as described above, the present invention provides a control apparatus and an induction motor for controlling an operation of an induction motor using an indirect vector control method without a speed sensor. The present invention is capable of estimating a velocity with a small observer gain through a DC current and a rotor flux, as well as a low error as well as a high speed region.

1 is a block diagram showing the configuration of an induction motor control apparatus according to an embodiment of the present invention;
2 shows the response characteristics of the velocity estimator of the synchronous dq coordinate system when the electric vehicle is running at a low speed and there is no parameter change;
3 shows the response characteristics of the velocity estimator of the synchronous dq coordinate system when the electric vehicle is running at a low speed and there is a 20% stator resistance change,
4 shows the response characteristics of the velocity estimator of the synchronous dq coordinate system when the electric vehicle is running at a low speed and there is a rotor resistance change of 20%
5 is a graph showing the response characteristics of the velocity estimator of the synchronous dq coordinate system when the electric vehicle is operated at a high speed and there is no parameter change;
6 is a graph showing the response characteristics of the velocity estimator of the synchronous dq coordinate system when the electric vehicle is operated at a high speed and there is a stator resistance change of 20%
7 shows the response characteristics of the velocity estimator of the synchronous dq coordinate system when the electric vehicle is running at a high speed and there is a rotor resistance change of 20%
8 is a graph showing a comparison between the synchronous dq coordinate system (left graph) and the fixed dq coordinate system (right graph) with respect to the response characteristic of the velocity estimator of the dq coordinate system according to each parameter change when the electric vehicle runs at a low speed, and ,
9 is a diagram showing a comparison between the synchronous dq coordinate system (left graph) and the fixed dq coordinate system (right graph) with respect to the response characteristic of the velocity estimator of the dq coordinate system according to each parameter change when the electric vehicle is operated at a high speed.

Various embodiments of the present invention will now be described with reference to the accompanying drawings.

1 is a block diagram showing the configuration of an induction motor control apparatus according to an embodiment of the present invention.

Referring to FIG. 1, an induction motor control apparatus according to an embodiment of the present invention includes a dq coordinate transformation unit 110, a dq coordinate system total state / velocity observation unit 120, a slip angular velocity calculation unit 130, An abc coordinate transformation unit 190, a PWM inverter 195, a magnetic flux controller 150, a velocity controller 160, current controllers 170 and 180, and an induction motor 200. [

The induction motor 200 according to the present invention does not include a speed sensor.

Referring to FIG. 1, the operation of the induction motor control apparatus will be described. First, the dq coordinate transformation unit 110 converts the current and voltage applied to the induction motor 200 into a dq coordinate system. As a result, the d-axis current i _ds , the d-axis voltage v _ds , the q-axis current i _{q s} , and the q-axis voltage ( i _ds ) vq _s ) is applied.

) 및 추정 각속도(

)를 계산한다. 구체적인 계산 방법에 대해서는 후술한다. 추정 회전자 자속(

)은 슬립각속도 계산부(130)로 인가되며, 추정 각속도(

)는 슬립 각속도(ω_sl )와의 오차를 계산하는데 사용된다.dq coordinate system state before / speed observation unit 120, the dq the d axis current converted into a coordinate system (i _ds), the d-axis voltage (v _ds), q-axis current (i _{q s)} and q-axis voltage (v _{q s} ) is used to estimate the estimated rotor flux (

) And the estimated angular velocity (

). A concrete calculation method will be described later. The estimated rotor flux (

Is applied to the slip angular velocity calculation unit 130, and the estimated angular velocity (

) Is used to calculate the error with the slip angular velocity [ omega] _sl .

)과 속도 제어기(160)에서 산출된 q 축 지령 전류(i_qs ^* )를 이용하여 슬립 각속도(ω_sl )를 계산한다. The slip angular velocity calculator 130 calculates the slip angular velocity

) And the q-axis command current ( i _qs ^* ) calculated by the speed controller 160 are used to calculate the slip angular speed ? _Sl .

)의 오차를 적분하여 위상각(θ_e )을 구한다. 구해진 위상각(θ_e )은 abc 좌표 변환부(190)와 dq 좌표 변환부(110)에 인가된다. The phase angle calculator 140 calculates the phase angle? _{L based} on the calculated slip angular velocity? _Sl and the estimated angular velocity?

) Integrating an error of the phase is obtained for each of (θ _e). The obtained phase angle _e is applied to the abc coordinate transformation unit 190 and the dq coordinate transformation unit 110.

The present invention uses an indirect vector control method for calculating the phase angle ? _E by the above-described method. Conventionally, an induction motor using a direct vector control method using a fixed dq coordinate system was generally used. This method has a problem that the performance of torque and speed control is deteriorated due to a large observation error in a high speed region as the variables to be observed have ac values. However, since the present invention basically uses direct current, as will be described later, observation errors do not occur in a high speed region.

Continuing explanation of the operation of the invention, the command rotor flux ? _Rd ^* and the command angular speed ? _R ^* are inputted to the control device of the induction motor 200.

)의 오차를 이용하여 d 축 지령 전류(i_ds ^* )를 생성한다. The magnetic flux controller 150 calculates the estimated rotor magnetic flux ( ? _Dr ^* ) output from the dq coordinate system pre-state / velocity observer 120

) Is used to generate the d-axis command current ( i _ds ^* ).

The current controller 170 generates the d-axis command voltage v _ds ^* using the error between the d-axis command current i _ds ^* and the d-axis current i _ds converted into the dq coordinate system described above. The generated d-axis command voltage v _ds ^* is applied to the abc coordinate converter 190.

)의 오차를 이용하여 q 축 지령 전류(i_qs ^* )를 생성한다. The speed controller 160 receives the command angular speed ? _R ^* and the estimated angular velocity ( ? _R ^* ) output from the dq coordinate system total state / velocity observer 120

) Is used to generate the q-axis command current ( i _qs ^* ).

The current controller 180 is connected to the speed controller 160 is the q-axis instruction current (i _{q s} ^*) and the converted into dq coordinate system, a q-axis current (i _{q s),} q-axis command voltage (v using the error _{q s} ^* ). The q-axis command voltage ( v _{q s} ^* ) is applied to the abc coordinate converter 190.

The abc coordinate transforming unit 190 transforms the d-axis command voltage v _ds ^* and the q-axis command voltage v _{q s} ^* into the abc coordinate system using the phase angle θ _e . The abc coordinate conversion unit 190 outputs the a-axis command voltage v _{a s} ^* , the b-axis command voltage v _{b s} ^* , and the c-axis command voltage v _{c s} ^* .

The PWM inverter 195 converts the command voltage converted to the abc coordinate system, that is, the a-axis command voltage v _{a s} ^* , the b-axis command voltage v _{b s} ^* , and the c-axis command voltage v _{c s} ^* Converted into an alternating current and input to the induction motor 200.

Hereinafter, a technique for variable speed driving of the induction motor for the induction motor control apparatus will be described in more detail.

The technique for the variable speed drive of the induction motor without the velocity sensor of the present invention includes a full-state flux observer (corresponding to the above-described dq coordinate system pre-state / velocity observer 120), a speed adaptation rule and an adaptive rule gain selection method do.

The state equation of the induction motor expressed in the dq coordinate system rotating at the synchronous speed is given by the following equation.

(1)

(One)

here,

: 고정자 전류,

: 회전자 자속

: Stator current,

: Rotor flux

: 고정자 전압,

: Stator voltage,

,

,

,

,

,

,

: 고정자 및 회전자 저항, [Ω]

: Stator and rotor resistance, [Ω]

: 고정자 및 회전자 자기인덕턴스, [H]

: Stator and rotor self-inductance, [H]

: 자화 인덕턴스, [H]

: Magnetizing inductance, [H]

: 회전자 및 동기 각속도 [rad/sec]

: Rotor and synchronous angular velocity [rad / sec]

: 슬립 각속도 (=

-

), [rad/sec]

: Slip angular velocity (=

-

), [rad / sec]

here,

The measurable variable is the stator current, and all electrical signals have a DC value.

When the slip condition for the indirect vector control is obtained from the magnetic flux equation of the state equation (1)

(2)

(2)

When this condition is satisfied, the q-axis rotor flux is always zero, the generated torque is proportional to the q-axis stator current, and the d-axis rotor flux is proportional to the d-axis stator current. Therefore, the coupling phenomenon is eliminated between the phases, and the torque and the magnetic flux can be directly controlled through the stator current. Since the slip condition in Eq. (2) is applied in the form of forward compensation, a closed-loop observer is required because accurate information about the d-axis rotor flux, λ _dr , is required. If the slip is generated by using an inaccurate ? _Dr , overshoot or pulsation occurs in the torque, coupling between phases occurs, and the maximum torque of the motor can not be used. Therefore, we first describe a parameter-insensitive all-state observer for rotor flux estimation and velocity estimation.

The total state observer for estimating the stator current and rotor flux using the stator current as a measurement variable is as follows in the synchronous dq coordinate system rotating at the synchronous speed.

(3)

(3)

Where G is the observer gain and the suffix "^" represents the estimate and the parameter value of the motor used in the observer. The observer gain is obtained by the same equation as the gain of the fixed dq coordinate system, given that the pole of the observer corresponds to k times the pole of the induction motor.

(4)

(4)

,

,

,

,

Since the speed of the motor can change rapidly, the following proportional-integral adaptive law is used.

(5)

(5)

이 작으면 속도추정의 전달함수는 다음의 2차함수로 생각할 수 있다.Whether the gain k of the pre-state observer is small,

The transfer function of the velocity estimation can be regarded as the following quadratic function.

(6)

(6)

을 자연 주파수,

를 제동비로 표시하면 비례-적분 이득은 다음과 같다.Therefore, the proportional-integral gain is obtained based on the standard form of the quadratic transfer function.

The natural frequency,

Is expressed by the braking ratio, the proportional-integral gain is as follows.

(7)

(7)

(6)

(6)

,

이므로 비례-적분 이득은 다음 식으로 나타낼 수 있다.In the steady state,

,

The proportional-integral gain can be expressed by the following equation.

(8)

(8)

(9)

(9)

이

의 크기보다 작게 되어 비례이득

는 음수가 되어 속도 적응칙이 불안정하게 된다. 이를 방지하기 위해서 속도 적응칙의 자연 주파수

을 크게 선정하면 고정자전류의 맥동과 잡음에 의해 추정속도에 큰 잡음이 나타나므로, 전상태 관측이득을 작게 하고,

을 비교적 크게 선정하는 것이 바람직하다.From the above equation, if the total state observer gain k is selected to be large

this

And the proportional gain

Becomes negative and the speed adaptation rule becomes unstable. To prevent this, the natural frequency of the speed adaptation law

A large noise is generated at the estimated speed due to pulsation and noise of the stator current, so that the total state observation gain is reduced,

Is selected to be relatively large.

FIGS. 2 to 8 are graphs showing simulation results of a response characteristic of a speed estimator when the control apparatus for an induction motor according to various embodiments of the present invention is applied to an electric vehicle.

2 is a graph showing the response characteristics of the velocity estimator of the synchronous dq coordinate system when the electric vehicle runs at a low speed and there is no parameter change.

In FIG. 2A, the actual speed is indicated by a dotted line, the estimated speed is indicated by a thick solid line, and the error between the two is indicated by a thin solid line.

As shown in FIG. 2 (B), it can be seen that the electric vehicle travels at a low speed and there is almost no error when there is no parameter change.

3 is a graph showing the response characteristics of the velocity estimator of the synchronous dq coordinate system when the electric vehicle is running at a low speed and there is a change in stator resistance of 20%.

In FIG. 3 (A), the actual speed is indicated by a dotted line, the estimated speed is indicated by a thick solid line, and the error between the two is indicated by a thin solid line.

As shown in FIG. 3 (B), it can be seen that even when the electric vehicle is operated at a low speed and the stator resistance is changed by 20%, the error hardly occurs.

4 is a graph showing the response characteristics of the velocity estimator of the synchronous dq coordinate system when the electric vehicle is running at a low speed and there is a rotor resistance change of 20%.

In FIG. 4 (A), the actual speed is indicated by a dotted line, the estimated speed is indicated by a thick solid line, and the error between the two is indicated by a thin solid line.

As shown in FIG. 4 (B), it can be seen that even when the electric vehicle runs at a low speed and there is a rotor resistance change of 20%, the error hardly occurs.

5 is a graph showing the response characteristics of the velocity estimator of the synchronous dq coordinate system when the electric vehicle is operated at a high speed and there is no parameter change.

In FIG. 5 (A), the actual speed is indicated by a dotted line, the estimated speed is indicated by a thick solid line, and the error between the two is indicated by a thin solid line.

As shown in FIG. 5 (B), it can be seen that the electric vehicle runs at a high speed, and there is no significant error when there is no parameter change.

6 is a graph showing the response characteristics of the velocity estimator of the synchronous dq coordinate system when the electric vehicle is operated at a high speed and there is a change in stator resistance of 20%.

In FIG. 6 (A), the actual speed is indicated by a dotted line, the estimated speed is indicated by a thick solid line, and the error between the two is indicated by a thin solid line.

As shown in FIG. 6 (B), it can be seen that even when the electric vehicle is operated at a high speed and the stator resistance is changed by 20%, no significant error occurs. As described above, since the present invention uses direct current, observation errors do not occur largely even in a high speed region.

7 is a graph showing the response characteristics of the velocity estimator of the synchronous dq coordinate system when the electric vehicle is operated at a high speed and there is a rotor resistance change of 20%.

In FIG. 7 (A), the actual speed is indicated by a dotted line, the estimated speed is indicated by a thick solid line, and the error between the two is indicated by a thin solid line.

As shown in FIG. 7 (B), it can be seen that even when the electric vehicle runs at a high speed and the rotor resistance changes by 20%, the error does not occur to a large extent. As described above, since the present invention uses direct current, observation errors do not occur largely even in a high speed region.

FIG. 8 is a diagram showing a comparison between the synchronous dq coordinate system (left graph) and the fixed dq coordinate system (right graph) with respect to the response characteristic of the velocity estimator of the dq coordinate system according to each parameter change when the electric vehicle runs at low speed.

In each graph, the actual speed is indicated by a dotted line, the estimated speed is indicated by a thick solid line, and the error between the two is indicated by a thin solid line.

Fig. 8 (a) shows the results when the stator resistance changes by 20%, and Fig. 8 (c) shows the results when the rotor resistance changes by 20%.

8, the response characteristics of the velocity estimator of the dq coordinate system according to each parameter change are significantly different between the synchronous dq coordinate system (left graph) and the fixed dq coordinate system (right graph) when the electric vehicle travels at a low speed .

9 is a diagram showing a comparison between the synchronous dq coordinate system (left graph) and the fixed dq coordinate system (right graph) with respect to the response characteristic of the velocity estimator of the dq coordinate system according to each parameter change when the electric vehicle is operated at a high speed.

In each graph, the actual speed is indicated by a dotted line, the estimated speed is indicated by a thick solid line, and the error between the two is indicated by a thin solid line.

Fig. 9 (a) shows the result when the stator resistance changes by 20%, and Fig. 9 (c) shows the result when the rotor resistance changes by 20%.

In the result shown in Fig. 9, the response characteristics of the velocity estimator of the dq coordinate system according to the change of each parameter in the case of running at a high speed, unlike the case where the electric vehicle is operated at a low speed, (Right graph).

As described above, in the induction motor for driving an electric vehicle according to various embodiments of the present invention, a technique for applying an indirect vector controller without a speed sensor uses a full-state observer and a speed adaptation rule that are robust against parameter variations. The above-mentioned technique shows that the state observer and the velocity adaptation rule designed in the dq coordinate system rotating at the synchronous speed can estimate the velocity more accurately in the high-speed operation region than in the conventional fixed dq coordinate system, . In addition, it has an advantage that it can be applied to an indirect vector control method which is mostly used in variable speed operation of an induction motor.

While the present invention has been particularly shown and described with reference to exemplary embodiments thereof, it is clearly understood that the same is by way of illustration and example only and is not to be construed as limiting the scope of the invention as defined by the appended claims. It will be understood by those skilled in the art that various changes in form and details may be made therein without departing from the spirit and scope of the present invention.

110: dq coordinate conversion unit 120: dq coordinate system total state / velocity observation unit
130: slip angular velocity calculation unit 140: phase angle calculation unit
150: magnetic flux controller 160: speed controller
170, 180: current controller
190: abc coordinate conversion unit 195: PWM inverter
200: Induction motor

Claims (17)

A control device for controlling an operation of an induction motor,
A dq coordinate converter for converting a current and a voltage applied to the induction motor into a dq coordinate system;
The current and voltage converted into the dq coordinate system are used to estimate the estimated rotor flux

) And the estimated angular velocity (

) Dq coordinate system total state / velocity observer;
The calculated estimated rotor flux (

A slip angular velocity calculation unit for calculating a slip angular velocity omega _sl using the angular velocity ? And
The calculated slip angular velocity? _Sl and the estimated angular velocity?

) To obtain a phase angle ( ? _E ). The method according to claim 1,
An abc coordinate converter for converting the command voltage into an abc coordinate system using the phase angle ? _E ; And
And a PWM inverter for converting the command voltage converted into the abc coordinate system into AC. The method according to claim 1,
The command rotor flux ? _Rd ^* and the estimated rotor flux?

A magnetic flux controller for generating a d-axis command current i _ds ^* using an error of the d-axis command current i _ds ^* ; And
The command angular velocity ? _R ^* and the estimated angular velocity?

) To generate a q-axis command current ( i _qs ^* ) using the error of the q-axis command current ( i _qs ^* ). The method of claim 3,
Wherein the slip angular velocity calculator comprises:
The calculated estimated rotor flux (

) And the q-axis command current ( i _qs ^* ) to calculate the slip angular velocity ( ? _Sl ). The method of claim 3,
A current controller for generating a d-axis command voltage ( v _ds ^* ) using an error between the d-axis command current ( i _ds ^* ) and the d-axis current converted into the dq coordinate system; And
A current controller for generating a q-axis command voltage ( _{q q s} ^* ) using an error between the q-axis command current ( i _{q s} ^* ) and the q-axis current converted into the dq coordinate system; Further comprising: The method according to claim 1,
Wherein the control device is driven by an indirect type vector control method. The method according to claim 1,
The induction motor includes:
And a speed sensor is not provided. The method according to claim 1,
The estimated angular velocity (

),
Wherein the calculation is performed by the following equation.

Here, in the synchronous dq coordinate system for estimating the stator current and rotor flux using the stator current as a measurement variable,

Where G is the observer gain that causes the pole of the observer to be k times the pole of the induction motor as follows,

,

,

And,

: Proportional gain,

: Integral gain

: Stator current,

: Rotor flux

: Stator voltage,

,

,

,

: Stator and rotor resistance, [Ω]

: Stator and rotor self-inductance, [H]

: Magnetizing inductance, [H]

: Rotor and synchronous angular velocity [rad / sec]

: Slip angular velocity (=

-

), [rad / sec]
i _qs : q-axis current converted to dq coordinate system
λ _dr : d-axis rotor flux in the dq coordinate system In the induction motor,
Induction motor; And
And a control device for controlling the induction motor,
The control device includes:
A dq coordinate converter for converting a current and a voltage applied to the induction motor into a dq coordinate system; and an estimated rotor flux meter using the current and voltage converted into the dq coordinate system

) And the estimated angular velocity (

A synchronous dq coordinate system total state / velocity observing section for calculating the estimated rotor flux

A slip angular velocity calculator for calculating a slip angular velocity ome _sl by using the calculated slip angular velocity omega _sl and the estimated angular velocity ome _sl ,

) To obtain a phase angle ( ? _E ). 10. The method of claim 9,
The control device includes:
An abc coordinate converter for converting the command voltage into an abc coordinate system using the phase angle ? _E ; And
And a PWM inverter for converting the command voltage converted into the abc coordinate system into an AC voltage. 10. The method of claim 9,
The control device includes:
The command rotor flux ? _Rd ^* and the estimated rotor flux?

A magnetic flux controller for generating a d-axis command current i _ds ^* using an error of the d-axis command current i _ds ^* ; And
The command angular velocity ? _R ^* and the estimated angular velocity?

Further comprising a velocity controller for generating a q-axis command current ( i _qs ^* ) using the error of the q-axis command current ( i _qs ^* ). 10. The method of claim 9,
The control device includes:
And is driven by an indirect type vector control method. 10. The method of claim 9,
The induction motor includes:
Wherein the speed sensor is not provided. delete delete 12. The method of claim 11,
Wherein the slip angular velocity calculator comprises:
The calculated estimated rotor flux (

) And the q-axis command current ( i _qs ^* ) to calculate the slip angular speed ( ? _Sl ). 12. The method of claim 11,
The control device includes:
A current controller for generating a d-axis command voltage ( v _ds ^* ) using an error between the d-axis command current ( i _ds ^* ) and the d-axis current converted into the dq coordinate system; And
A current controller for generating a q-axis command voltage ( _{q q s} ^* ) using an error between the q-axis command current ( i _{q s} ^* ) and the q-axis current converted into the dq coordinate system; Further comprising: an induction motor.

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