KR20150129704A - Excitation and use of guided surface wave modes on lossy media - Google Patents

Abstract

Various embodiments are disclosed for transmitting and / or receiving energy carried in the form of an induced surface waveguide mode along the surface of a lossy conduction medium by exciting a polyphase waveguide probe.

Description

[0001] EXCITATION AND USE OF GUIDED SURFACE WAVE MODES ON LOSSY MEDIA [0002]

Cross reference of related application

This patent application is a continuation-in-part of US patent application Ser. No. 13 / 789,525 entitled "EXCITATION AND USE OF GUIDED SURFACE WAVE MODES ON LOSSY MEDIA" filed on Mar. 7, 2013, No. 13 / 789,538 entitled " EXCITATION AND USE OF GUIDED SURFACE WAVE MODES ON LOSSY MEDIA ", both of which are incorporated herein by reference in their entirety.

For more than a century, signals transmitted by radio waves have been accompanied by launched fields using conventional antenna structures. Unlike radio science, power distribution systems of the last century were accompanied by the transfer of energy induced along conductors. This understanding of the difference between radio frequency (RF) and power delivery has emerged since the early 1900s.

Many aspects of the present disclosure can be better understood with reference to the following drawings. The components in the figures are not necessarily drawn to scale, emphasis instead being placed upon illustrating the principles of the present disclosure. Moreover, in the drawings, like reference numbers all refer to corresponding elements throughout the several views.
Figure 1 is a chart showing field intensities as a function of distance to an induced electromagnetic field and a radiated electromagnetic field.
2 is a diagram showing a radio interface having two regions used for transmission of surface acoustic waves in accordance with various embodiments of the present disclosure;
Figure 3 is a diagram showing a polyphase waveguide probe disposed for the propagation interface of Figure 2, in accordance with one embodiment of the present disclosure;
4 is an illustration of one example of the phase shift of the ground current facilitating the launch of the induced surface-waveguide mode on the lossy conduction medium in the propagation interface of Fig. 3, in accordance with one embodiment of the present disclosure.
Figure 5 is a diagram showing complex insertion angles of an electric field synthesized by polyphase waveguide probes, in accordance with various embodiments of the present disclosure.
6 is a schematic diagram of a polyphase waveguide probe according to one embodiment of the present disclosure;
Figures 7A-J are schematic diagrams of specific examples of the polyphase waveguide probe of Figure 6, in accordance with various embodiments of the present disclosure.
8A-C are graphs illustrating field intensities of surface acoustic waves at selected transmission frequencies generated by various embodiments of polyphase waveguide probes, in accordance with various embodiments of the present disclosure.
9 shows an example of a graph of experimental measurements of the field strength of a surface acoustic wave at 59 MHz as a function of distance generated by a polyphase waveguide probe, according to one embodiment of the present disclosure.
Figure 10 shows a graph of experimental measurements of phase as a function of distance of the surface acoustic wave of Figure 9, according to one embodiment of the present disclosure.
11 shows another example of a graph of experimental measurements of field intensities as a function of distance of surface acoustic waves generated by a polyphase waveguide probe at 1.85 MHz, according to one embodiment of the present disclosure.
12A-B illustrate examples of receivers that can be used to receive energy transmitted in the form of surface-acoustic waves launched by a polyphase waveguide probe, in accordance with various embodiments of the present disclosure.
13 shows an example of an additional receiver that may be used to receive energy transmitted in the form of surface-acoustic waves launched by a polyphase waveguide probe, in accordance with various embodiments of the present disclosure.
Figure 14A shows a schematic diagram illustrating the Thevenin-equivalent of the receivers shown in Figures 12A-B, in accordance with one embodiment of the present disclosure.
Figure 14B shows a schematic diagram illustrating the Norton equivalent of the receiver shown in Figure 13, in accordance with one embodiment of the present disclosure.

First, with reference to FIG. 1, a predetermined terminology will be set to provide clarity in the description of the following concepts. First, there is a formal difference between the radiated electromagnetic field and the induced electromagnetic field as contemplated herein.

As contemplated herein, a radiated electromagnetic field includes electromagnetic energy emitted from a source structure in the form of waves that do not direct to the waveguide. For example, a radiated electromagnetic field is a field that typically originates from an electrical structure, such as an antenna, propagates through the atmosphere or other medium, and does not lead to any waveguide structure. When radiated electromagnetic waves originate from an electrical structure such as an antenna, they continue to propagate in the propagation medium (such as air) regardless of their source until they disappear, regardless of whether their source continues to operate. When electromagnetic waves are emitted, they can not be restored unless they are intercepted, and if not intercepted, the energy inherent in the radiated electromagnetic waves is lost forever. Electrical structures such as antennas are designed to radiate electromagnetic fields by maximizing the ratio of the radiation resistance to the structural loss resistance. The radiant energy diffuses into the space and is lost regardless of the presence of the receiver. The energy density of radiation fields is a function of distance due to geometric spreading. Thus, the term "radiation" is used herein to refer to this type of electromagnetic propagation in all its forms as used herein.

An induced electromagnetic field is a radio-frequency electromagnetic wave whose energy is concentrated in or near the boundaries between media having different electromagnetic characteristics. In this sense, the induced electromagnetic field is an electromagnetic field directed to the waveguide and can be characterized as being carried by the current flowing in the waveguide. If there is no load to receive and / or dissipate the energy carried in the inductive electromagnetic wave, no energy is lost other than being dissipated by conduction of the induction medium. That is, when there is no load on the induced electromagnetic wave, no energy is consumed. Thus, a generator or other source generating an inductive electromagnetic field will not deliver actual power in the absence of a resistive load. Because of this, such a generator or other source is essentially idle until a load is provided. This is analogous to driving a generator to generate 60 Hertz electromagnetic waves transmitted through power lines where no electrical load is present. It should be noted that the induction electromagnetic field or electromagnetic wave is equivalent to what is called "transmission line mode ". This is in contrast to the radiated electromagnetic waves, which are always supplied with the actual power to generate the radiation waves. Unlike radiating electromagnetic waves, the induced electromagnetic energy does not propagate along the finite-length waveguide after the energy source is turned off. Thus, the term "induction " refers to this mode of electromagnetic propagation in all its forms as used herein.

To further illustrate the difference between the radiated and induced electromagnetic fields, FIG. 1, which shows a graph 100 of field strength in decibels (dB) above a certain reference value in volts per meter as a function of distance in kilometer units on a log- . The graph 100 of FIG. 1 shows an induced field strength curve 103 that shows the field strength of the inductive field as a function of distance. This derivative field strength curve 103 is essentially the same as the transmission line mode. The graph 100 of Figure 1 also shows a field strength curve 106 that shows the field strength of the radiated electromagnetic field as a function of distance.

의 특성 지수 감쇠를 가지며, 특이한 무릎(109)을 갖는다. 따라서, 도시된 바와 같이, 유도 전자기장의 장 강도는

의 레이트로 떨어지는 반면, 방사 전자기장의 장 강도는

의 레이트로 떨어지며, 여기서 d는 거리이다. 유도 장 강도 곡선(103)이 지수적으로 떨어진다는 사실로 인해, 유도 장 강도 곡선(103)은 전술한 바와 같은 무릎(109)을 특징으로 한다. 유도 장 강도 곡선(103) 및 방사 장 강도 곡선(106)은 교차 거리에서 발생하는 교차점(113)에서 교차한다. 교차 거리보다 작은 거리들에서, 유도 전자기장의 장 강도는 대부분의 위치들에서 방사 전자기장의 장 강도보다 훨씬 더 높다. 교차 거리보다 큰 거리들에서는 그 반대이다. 따라서, 유도 및 방사 장 강도 곡선들(103, 106)은 유도 및 방사 전자기장들 사이의 기본적인 전파 차이를 더 설명한다. 유도 및 방사 전자기장들 사이의 차이에 대한 비공식적인 설명에 대해, Milligan, T., Modern Antenna Design, McGraw-Hill, 1^st Edition, 1985, pp.8-9를 참조하며, 이 논문 전체는 본 명세서에 참고로 포함된다.The shapes of the curves 103/106 for radiation and guided wave propagation are important. The field strength curve 106 is geometrically (1 / d, where d is the distance) and is a straight line on the log-log scale. On the other hand, the induced field strength curve 103

And has a unique knee 109. The knee 109 has a characteristic index attenuation of? Thus, as shown, the field strength of the inductive field is

, While the field strength of the radiated electromagnetic field decreases

, Where d is the distance. Due to the fact that the induced field strength curve 103 falls exponentially, the field strength curve 103 is characterized by the knee 109 as described above. The induced field strength curve 103 and the field strength curve 106 intersect at an intersection point 113 that occurs at an intersection distance. At distances less than the crossover distance, the field strength of the induced electromagnetic field is much higher than the field strength of the radiated electromagnetic field at most locations. At distances greater than the intersection distance, the opposite is true. Thus, the induced and radiated field intensity curves 103 and 106 further illustrate the fundamental propagation differences between the induced and radiated electromagnetic fields. See Milligan, T., Modern Antenna Design , McGraw-Hill, 1 ^st Edition, 1985, pp. 8-9 for an informal description of the differences between inductive and radiated fields, ≪ / RTI >

The difference between the above-mentioned radiation and induced electromagnetic waves is easily expressed formally and is strictly evaluated. It is analytically deduced from the boundary conditions imposed on the problem that two such different solutions can come from one and the same linear partial differential equation, the wave equation. The Green function itself for the wave equation includes the difference between the characteristics of the radiation and the induced waves.

In an empty space, the wave equation is a differential operator whose eigenfunctions possess a continuous spectrum of eigenvalues on the complex frequency plane. These transverse electromagnetic (TEM) fields are called fields of radiation, and such fields are called "Hertz waves". However, in the presence of a conduction boundary, the wave equations and boundary conditions mathematically derive a spectral representation of the waves constituting the sum of continuous spectra and individual spectra. Because of this, Sommerfeld, A., "Uber die Ausbreitung der Wellen in der Drahtlosen Telegraphie," Annalen der Physik, Vol. 28, 1909, pp. 665-736. Also, Partial Differential Equations in Physics - Lectures on Theoretical Physics: Volume VI , Academic Press, 1949, pp. Sommerfeld, A., "Problems of Radio," as published in Chapter 6 at 236-289, 295-296; Collin, RE, "Hertzian Dipole Radiating Over a Lossy Earth or Sea: Some Early and Late 20 th Century Controversies," IEEE Antennas and Propagation Magazine, Vol. 46, No. 2, April 2004, pp. 64-79; And Reich, HJ, Ordnung, PF, Krauss, HL, and Skalnik, JG, Microwave Theory and Techniques , Van Nostrand, 1953, pp. 291-293, each of which is incorporated herein by reference in its entirety.

To summarize, first, a continuous part of the wave-like eigenvalue spectrum corresponding to branch-cut integra- tions produces an emissive field, and second, the individual spectra from the poles surrounded by the path of integration The corresponding residual sum causes non-TEM mobile surface waves to be exponentially damped in the direction across the wave. Such surface waves are inductive transmission modes. For further explanation, Friedman, B., Principles and Techniques of Applied Mathematics , Wiley, 1956, pp. pp. 214, 283-286, 290, 298-300.

및

를 갖는 밖으로 전파하는 RF 에너지는 영원히 손실된다. 한편, 도파관 프로브들은 개별 고유값들을 여기하며, 이는 송신선 전파를 유발한다. Collin, R. E., Field Theory of Guided Waves, McGraw-Hill, 1960, pp. 453, 474-477을 참조한다. 그러한 이론적 분석들은 손실 균질 매체의 평면 또는 구면 위에 개방 표면 유도파들을 런칭할 가설적 가능성을 보였지만, 일 세기 이상 동안, 임의의 실용적인 효율로 이것을 달성하기 위한 어떠한 알려진 구조도 엔지니어링 분야에 존재하지 않았다. 불행하게도, 전술한 이론적 분석은 1900년대 초에 출현한 이후로, 본질적으로 이론으로 남았으며, 손실 균질 매체의 평면 또는 구면 위의 개방 표면 유도파들의 런칭을 실용적으로 달성하기 위한 어떠한 알려진 구조도 존재하지 않았다.In free space, the antennas excite consecutive eigenvalues of the wave equation, which is the radiation field,

And

The outgoing RF energy with out is lost forever. On the other hand, the waveguide probes excite individual eigenvalues, which cause propagation of the transmission current. Collin, RE, Field Theory of Guided Waves , McGraw-Hill, 1960, pp. 453, 474-477. While such theoretical analyzes have shown the hypothetical possibility of launching open surface induction waves on a planar or spherical surface of a lossy homogeneous medium, there has been no known structure in the engineering field for more than a century to achieve this with any practical efficiency. Unfortunately, the above-mentioned theoretical analysis has remained essentially theorem since its appearance in the early 1900's, and there is no known structure for practically achieving the launch of open surface induction waves on a planar or spherical surface of a lossy homogeneous medium Did not do it.

In accordance with various embodiments of the present disclosure, various polyphase waveguide probes are described that are configured to excite radial surface currents having resulting chan- nels that synthesize the shape of surface waveguide modes along the surface of the lossy conduction medium. Such induced electromagnetic fields are substantially mode matched to the induced surface wave mode on the surface of the lossy conduction medium in magnitude and phase. Such a surface acoustic wave mode can also be referred to as a Zenneck surface wave mode. Due to the fact that the resulting fields excited by the polyphase waveguide probes described herein are substantially mode-matched to the first-order surface-acoustic-wave mode on the surface of the loss-conducting medium, the induced electromagnetic field in the form of a first- Launch. According to one embodiment, the lossy conduction medium comprises a ground medium such as earth.

Referring to FIG. 2, Jonathan Zenek's paper Zenneck, J., "On the Propagation of Plane Electromagnetic Waves Along a Flat Conducting Surface and Their Relation to Wireless Telegraphy, Annalen der Physik, Serial 4, Vol. 23, September 20, 1907, pp. There is shown a propagation interface which provides an examination of boundary value solutions for Maxwell equations derived in 1907 by Jonathan Zenack, Fig. 2 shows the cylindrical coordinates for the waves propagating radially along the interface between the lossy conductive medium designated as region 1 and the insulator designated as region 2. Fig. Region 1 may comprise, for example, any lossy conduction medium. In one example, such a loss conduction medium may comprise a ground medium, such as earth or other medium. Region 2 is a second medium that shares a boundary interface with region 1 and has region 1 and other configuration parameters. Region 2 may comprise, for example, any insulator, such as an atmosphere or other medium. The reflection coefficient for such a boundary interface is zero only for incidence at a complex Brewster angle. Stratton, JA, Electromagnetic Theory , McGraw-Hill, 1941, p. 516.

According to various embodiments, the present disclosure describes various polyphase waveguide probes that generate electromagnetic fields that are substantially mode-matched to the first-order surface-acoustic-wave mode on the surface of the lossy conduction medium comprising region 1. According to various embodiments, such electromagnetic fields substantially combine with a wavefront incident at a complex Brewster angle of the lossy conduction medium causing zero reflection.

장 변화가 가정되고,

이고,

(z는 영역 1의 표면에 수직인 수직 좌표이고, ρ는 원통 좌표들에서의 방사상 차원임)인 영역 2에서, 인터페이스를 따르는 경계 조건들을 맥스웰 방정식들의 제넥의 폐쇄 형태의 정확한 해는 아래의 전기장 및 자기장 성분들에 의해 표현된다.More specifically,

Chang changes are assumed,

ego,

(z is the vertical coordinate perpendicular to the surface of the region 1, and rho is the radial dimension in the cylindrical coordinates), the exact solution of the closed form of the first order of the Maxwell equations is given by the following electric field And magnetic field components.

장 변화가 가정되고,

이고,

인 영역 1에서, 인터페이스를 따르는 경계 조건들을 맥스웰 방정식들의 제넥의 폐쇄 형태의 정확한 해는 아래의 전기장 및 자기장 성분들에 의해 표현된다.

Chang changes are assumed,

ego,

In region 1, the exact solution of the closed form of the boundary conditions along with the interface of the Maxwell equations is expressed by the following electric and magnetic field components.

는 제2 유형 및 차수

의 복소 인수 핸켈(Hankel) 함수이고,

은 영역 1에서의 양의 수직 방향에서의 전파 상수이고,

는 영역 2에서의 수직 방향에서의 전파 상수이고,

은 영역 1의 전도율이고,

는 2πf와 동일하고, f는 여기 주파수이고,

는 자유 공간의 유전율이고,

은 영역 1의 유전율이고, A는 소스에 의해 부과되는 소스 상수이고, z는 영역 1의 표면에 수직인 수직 좌표이고,

는 표면파 방사상 전파 상수이고, ρ는 방사상 좌표이다.In these expressions,

The second type and order

Hankel < / RTI > function of < RTI ID =

Is the propagation constant in the positive vertical direction in the region 1,

Is the propagation constant in the vertical direction in the region 2,

Is the conductivity of region 1,

Is equal to 2? F, f is the excitation frequency,

Is the permittivity of the free space,

Is the dielectric constant of region 1, A is the source constant imposed by the source, z is the vertical coordinate perpendicular to the surface of Region 1,

Is the surface wave radial wave constant, and r is the radial coordinate.

Propagation constants in the ± z directions are determined by separating the wave equation above and below the interface between region 1 and region 2 and imposing boundary conditions.

This challenge is addressed in Region 2

, And in region 1

Lt; / RTI >

는 복소 식인Radial wave constant

Is a complex expression

Lt; / RTI > In all of the above expressions,

는 자유 공간의 투자율을 포함하고,

은 영역 1의 상대 유전율을 의미한다. 따라서, 생성되는 표면파는 인터페이스에 평행하게 전파하며, 그에 수직으로 지수적으로 감쇠한다. 이것은 소실로서 알려져 있다., Where

Includes the permeability of free space,

Is the relative permittivity of region 1. Thus, the generated surface wave propagates parallel to the interface, and exponentially decays perpendicularly thereto. This is known as loss.

Thus, Equations 1-3 can be considered to be a cylindrical symmetric, radial waveguide mode. Barlow, HM, and Brown, J., Radio Surface Waves , Oxford University Press, 1962, pp. 10-12, 29-33. The present disclosure details the structures that excite this "open boundary" waveguide mode. Specifically, in accordance with various embodiments, there is provided a system and method for receiving voltages and / or currents to excite the relative phases of fields of a surface waveguide mode disposed against each other and launched along a boundary interface between region 2 and region 1 A multi-phase waveguide probe is provided having charge terminals of appropriate size.

Further, the Leontovich impedance boundary condition between region 1 and region 2 is determined as follows.

은 양의 수직(+z) 방향에서의 단위 법선이고,

는 위의 식 1에 의해 표현되는 영역 2에서의 자기장 강도이다. 식 12는 식 1-3에서 지정된 장들이 경계 인터페이스를 따라 방사상 표면 전류 밀도를 구동함으로써 획득될 수 있다는 것을 의미하며, 그러한 방사상 표면 전류 밀도는 아래 식에 의해 지정된다.here,

Is the unit normal in the positive vertical (+ z) direction,

Is the magnetic field strength in region 2 expressed by Equation 1 above. Equation 12 means that the chapters designated in equations 1-3 can be obtained by driving a radial surface current density along a boundary interface, and such radial surface current density is specified by the following equation.

Here, A is a constant that has not yet been determined. In addition, near the polyphase waveguide probe (for ρ << λ), the above equation 13 has the following behavior.

에 대한 장 매칭에 의해, 식 1-6 및 13에서 다음을 발견한다.You may want to pay attention to negative signs. This means that when the source current flows vertically upward, the necessary "close" ground current flows radially inward. "near"

By the field matching on Equation 1-6 and 13, we find the following.

Thus, Equation 13 can be rewritten as follows.

3, there is shown an example of a polyphase waveguide probe 200 including a charge terminal T ₁ and a charge terminal T ₂ arranged along a vertical axis z. The multi-phase waveguide probe 200 is disposed on the lossy conduction medium 203 according to one embodiment of the present disclosure. The loss conduction medium 203 constitutes region 1 (FIG. 2) in accordance with one embodiment. In addition, the second medium 206 shares a boundary interface with the lossy conduction medium 203 and constitutes a region 2 (Fig. 2). The multiphase waveguide probe 200 includes a probe coupling circuit 209 that couples the excitation source 213 to the charge terminals T ₁ and T ₂ as will be described in more detail with reference to subsequent figures.

)의 점근선들을 가까운(close-in)

및 먼(far-out)

인 것으로 결정할 수 있으며, Charge terminals (T ₁ , T ₂ ) are disposed on the lossy conduction medium (203). The charge terminal T ₁ may be regarded as a capacitor and the charge terminal T ₂ may comprise a corresponding or lower capacitor as described herein. According to one embodiment, the charge terminal T ₁ is located at a height H ₁ , the charge terminal T ₂ is located at a height H ₂ , just below T ₁ along a vertical axis, H ₂ is H ₁ Lt; / RTI &gt; The height h of the transmission structure provided by the polyphase waveguide probe 200 is h = H ₁ - H ₂ . Given the above description, the radial neck surface current on the surface of the lossy conductive medium (

Lt; RTI ID = 0.0 &gt; close-in &

And far-out

, &Lt; / RTI &gt;

은 제1 전하 단자(T₁) 상에 전하(

)를 공급하는 전도 전류이고, 는 제2 전하 단자(T₂) 상에 전하(

)를 공급하는 전도 전류이다. 상부 전하 단자(T₁) 상의 전하(

)는

에 의해 결정되며, 여기서

은 전하 단자(T₁)의 격리된 용량이다. 전술한

에 대한 제3 성분이

로 주어지며, 이는 레온토비크 경계 조건으로부터 추정되고, 제1 전하 단자 상의 상승된 발진 전하(

)의 의사 정적 장에 의해 펌핑되는 손실 전도 매체(203) 내의 방사상 전류 기여라는 점에 유의한다. 양

은 손실 전도 매체의 방사상 임피던스이며, 여기서

이다.here,

Lt; _{RTI ID} = 0.0 &_gt; (T1) &_lt; / RTI &

), &Lt; / RTI &gt; On the second charge terminal T ₂

). &Lt; / RTI &gt; The charge on the upper charge terminal T ₁ (

)

Lt; RTI ID = 0.0 &gt;

Is the isolated capacitance of the charge terminal T ₁ . Described above

Lt; RTI ID = 0.0 &gt;

, Which is estimated from the Leon Tobik boundary condition, and the elevated oscillating charge on the first charge terminal

Lt; RTI ID = 0.0 &gt; 203 &lt; / RTI &gt; amount

Is the radial impedance of the lossy conduction medium, where

to be.

)는 크기 및 위상에 있어서 전류 점근선들에 가능한 한 밀접하게 매칭되도록 합성된다. 즉, 가까운

은

에 접해야 하며, 먼

은

에 접해야 한다. 또한, 다양한 실시예들에 따르면,

의 위상은 가까운

의 위상으로부터 먼

의 위상으로 전이해야 한다.The asymptotic lines representing the near and distant radial currents as described by equations 17 and 18 are complex quantities. According to various embodiments, the physical surface current (&lt; RTI ID = 0.0 &gt;

) Is synthesized to match the current asymptotes in size and phase as closely as possible. That is,

silver

And distant

silver

. Also, according to various embodiments,

The phase of

Far from the phase of

To the phase of &lt; / RTI &gt;

)은

인

의 위상으로 인해

에 대응하는 전파 위상 + 고정 "위상 부스트"에 비례한다는 점에 유의해야 한다.According to one embodiment, if any of the various embodiments of the multiphase waveguide probes described herein are suitably tuned, this arrangement provides at least approximate magnitude and phase matches for the neck, will be. Far phase

)silver

sign

Due to the phase of

Is proportional to the propagation phase + fixed "phase &lt; / RTI &gt; boost corresponding to &quot;

는 위의 식 9에서 표현되며, 손실 전도 매체 상의 송신 위치에서의

및

에 대한 값들 및 2개의 복소 루트를 갖는 동작 주파수(f)(

)에 따라, 통상적으로 약 45도 또는 225도 정도이다. 즉, 제넥 표면파를 런칭하도록 송신 위치에서 제넥 표면파 모드에 매칭시키기 위해, 먼 표면 전류(

)의 위상은

에 대응하는 전파 위상 + 약 45도 또는 225도의 상수만큼 가까운 표면 전류(

)의 위상과 달라야 한다. 이것은

에 대한 2개의 루트, 즉 하나의 가까운

및 하나의 가까운

가 존재하기 때문이다. 적절히 조정된 합성 방사상 표면 전류는 다음과 같다.here,

Lt; / RTI &gt; is expressed in Equation (9)

And

(F) &lt; / RTI &gt; having two complex roots &lt; RTI ID = 0.0 &gt;

), Typically about 45 degrees or about 225 degrees. That is, in order to match the generator surface wave mode at the transmitting position to launch the generator surface wave,

) Is in phase

/ RTI &gt; of about &lt; RTI ID = 0.0 &gt; 45 &lt; / RTI &

). this is

Lt; RTI ID = 0.0 &gt;

And one near

. The properly adjusted synthetic radial surface current is:

표면 전류는 아래 식에 따르는 장들을 자동으로 생성한다.By Maxwell's equations, such

The surface currents automatically generate fields according to the equation below.

)와 가까운 표면 전류(

) 간의 위상차는 전술한 식 20-23의 핸켈 함수들의 고유 특성들에 기인한다. 식 1-6 및 20에 의해 표현되는 장들은 그라운드 파 전파와 관련된 바와 같은 방사 장들이 아니라, 손실 인터페이스와 관련된 송신선 모드의 특성을 갖는다는 것을 인식하는 것이 중요하다. Barlow, H. M. and Brown, J., Radio Surface Waves, Oxford University Press, 1962, pp. 1-5를 참고한다. 이러한 장들은 0의 반사에 대한 복소 브루스터 각도 요구를 자동으로 충족시키며, 이는 아래에 제공되는 실험 결과들에서 확인되고 지지되는 바와 같이 방사가 무시되는 반면에 표면 유도파 전파가 극적으로 향상된다는 것을 의미한다.Thus, the distant surface current (&lt; RTI ID = 0.0 &gt;

) And near surface current (

) Is due to the intrinsic characteristics of the Hankel functions of the above-mentioned formulas 20-23. It is important to note that the chapters represented by equations 1-6 and 20 have characteristics of the transmission line mode associated with the lost interface, not the radiation fields associated with ground wave propagation. Barlow, HM and Brown, J., Radio Surface Waves , Oxford University Press, 1962, pp. Refer to 1-5. These chapters automatically meet the complex Brewster angle requirement for zero reflection, which means that surface induced wave propagation is dramatically improved while radiation is ignored, as confirmed and supported in the experimental results provided below do.

At this point, a review of the properties of the Hankel functions used in equations 20-23 is provided with an emphasis on the peculiarities of these solutions of the wave equation. The first and second types and the Hankel functions of degree n are defined as complex combinations of the first and second types of standard Bessel functions.

와 유사하다. 예를 들어, Harrington, R.F., Time-Harmonic Fields, McGraw-Hill, 1961, pp. 460-463을 참고한다.These functions represent cylindrical waves propagating radially inward (superscript (1)) and outward (superscript (2)). This definition

. For example, Harrington, RF, Time-Harmonic Fields , McGraw-Hill, 1961, pp. 460-463.

가 나가는 파라는 것은

및

의 시리즈 정의들로부터 직접 획득되는 그의 큰 인수 점근선 거동으로부터 쉽게 인식된다.

The para that goes out

And

Is easily recognized from its large argument asymptotic behavior obtained directly from the series definitions of &lt; RTI ID = 0.0 &gt;

과 곱해질 때

공간 변화를 갖는

형태의 밖으로 전파하는 원통파이다. 지수 성분의 위상은

이다. 아래 식도 명백하며,This expression

When multiplied by

With spatial variation

It is a cylindrical wave that propagates out of form. The phase of the exponential component is

to be. The following esophageal is obvious,

The more useful properties of the Hankel functions are expressed as follows,

These equations are described in Jahnke, E., and F. Emde, Tables of Functions , Dover, 1945, p. 145 &lt; / RTI &gt;

In addition, the small arguments and large argument asymptotes of the handler function propagating out are as follows.

만큼 위상이 다르다.Note that these asymptotic representations are complex quantities. Also, unlike normal sinusoidal functions, the behavior of complex Hanckel functions is different at near and far from origin. When x is the actual positive, equations 29 and 30 are 45 degrees of additional phase advance or "phase boost" or equivalent corresponding to lambda / 8

The phase is different.

(도 3)과

(도 3) 간의 위상 전이를 더 설명하기 위해, 다상 도파관 프로브(200)(도 3)의 위치에 대한 가까운 표면 전류(

) 및 먼 표면 전류(

)의 위상들의 예가 도시된다. 도 4에 도시된 바와 같이, 3개의 상이한 관측점(

,

,

)이 존재한다. 전이 영역은 관측점(

)과 관측점(

) 사이에 위치한다. 관측점(

)은 다상 도파관 프로브(200)의 위치에 위치한다. 관측점(

)은 전이 영역(216)과 관측점(

) 사이에 관측점(

)을 배치하는 관측점(

)으로부터의 거리(

)에 "가깝게" 위치한다. 관측점(

)은 도시된 바와 같이 전이 영역 216을 넘는 관측점(

)으로부터의 거리(

)에 "멀리" 위치한다.Referring to Figure 4,

(Fig. 3) and

Phase waveguide probe 200 (FIG. 3) to further describe the phase transition between the multi-phase waveguide probe 200 (FIG. 3)

) And distant surface current (

) Are shown. As shown in Fig. 4, three different viewpoints (

,

,

). The transition area is the observation point (

) And the observation point

. Viewpoints (

Is located at the position of the polyphase waveguide probe 200. Viewpoints (

) &Lt; / RTI &gt; includes a transition region 216 and a viewpoint (

) Between the observation points (

) Of the observation point

) &Lt; / RTI &gt;

). &Lt; / RTI &gt; Viewpoints (

) &Lt; / RTI &gt; passes through the observation point (&lt; RTI ID = 0.0 &gt;

) &Lt; / RTI &gt;

Quot; far ").

)에서, 방사상 전류(J)의 크기 및 위상은

으로 표현된다. 관측점(

)에서, 방사상 전류(J)의 크기 및 위상은

로 표현되며, 여기서

의 위상 시프트는 관측점들(

,

) 사이의 거리(

)에 기인할 수 있다. 관측점(

)에서, 방사상 전류(J)의 크기 및 위상은

로 표현되며, 여기서

의 위상 시프트는 관측점들(

,

) 사이의 거리(

) 및 전이 영역(216)에서 발생하는 추가 위상 시프트에 기인할 수 있다. 추가 위상 시프트(

)는 전술한 바와 같은 핸켈 함수의 특성으로서 발생한다.Viewpoints (

), The magnitude and phase of the radial current (J)

. Viewpoints (

), The magnitude and phase of the radial current (J)

Lt; / RTI &gt;

The phase shifts of the observation points (

,

)

). &Lt; / RTI &gt; Viewpoints (

), The magnitude and phase of the radial current (J)

Lt; / RTI &gt;

The phase shifts of the observation points (

,

)

And an additional phase shift that occurs in the transition region 216. [ Additional phase shift (

) Occurs as a characteristic of the Hankel function as described above.

)를 생성한 후에 먼

전류로 전이한다는 사실을 반영한다. 전이 영역(216)에서, 제넥 표면 도파관 모드의 위상은 약 45도 또는

만큼 전이한다. 이러한 전이 또는 위상 시프트는 "위상 부스트"로서 간주될 수 있는데, 그 이유는 제넥 표면 도파관 모드의 위상이 전이 영역(216)에서 45도만큼 진행하는 것으로 나타나기 때문이다. 전이 영역(216)은 동작 주파수의 파장의 1/10 미만의 어느 곳에선가 발생하는 것으로 나타난다.The above description indicates that the polyphase waveguide probe 200 is close to the surface current

RTI ID = 0.0 &gt;

Reflects the fact that the current transitions to the current. In the transition region 216, the phase of the generator surface waveguide mode is about 45 degrees or

. This transition or phase shift can be viewed as a "phase boost" because the phase of the generator surface waveguide mode appears to advance by 45 degrees in the transition region 216. [ Transition region 216 appears to occur anywhere less than one tenth of the wavelength of the operating frequency.

Referring again to FIG. 3, in accordance with one embodiment, a polyphase waveguide probe can be created that launches an appropriate radial surface current distribution. According to one embodiment, a necked waveguide mode is created in the radial direction. If J (r) given by Eq. 20 can be generated, this will automatically launch the generator surface waves.

In addition, regarding the charge images (Q ₁ ', Q ₂ ') of the charges (Q ₁ , Q ₂ ) on the charge terminals (T ₁ , T ₂ ) of one exemplary polyphase waveguide probe shown in FIG. Additional explanations are provided. The analysis is derived valid image charges below the multi-phase waveguide probe matching the charge of the (Q _1, Q ₂₎ on the electric charges storage, as described herein (T _1, T ₂₎ on loss of conductive media (Q ₁ ', Q ₂ '). These image charges (Q ₁ ', Q ₂ ') should also be considered in the analysis. These image charges Q ₁ ', Q ₂ ' are only 180 degrees out of phase with respect to the main source charges Q ₁ , Q ₂ on the charge reservoirs T ₁ , T ₂ as in the case of a perfect conductor It is not different. For example, lossy conductive media such as terrestrial media provide phase shifted images. That is, the image charges Q ₁ ', Q ₂ ' are at complex depths. For a description of complex images, see Wait, JR, "Complex Image Theory-Revised," IEEE Antennas and Propagation Magazine , Vol. 33, No. 4, August 1991, pp. 27-29, the entirety of which is incorporated herein by reference.

)에 있는 대신에, 전도 미러(215)가 깊이 z = -d/2에 배치되고, 이미지 자체는

에 의해 주어지는 "복소 거리"(즉, "거리"는 크기 및 위상 양자를 가짐)에 나타나며, 여기서 n = 1, 2이고, 수직 분극 소스들에 대해,The image charges Q ₁ 'and Q ₂ ' are at the same depth as the heights of the charges Q ₁ and Q ₂ (that is,

, The conducting mirror 215 is arranged at the depth z = -d / 2, and the image itself

(I. E., The "distance" has both magnitude and phase) given by n = 1, 2, and for vertically polarized sources,

ego,

ego,

to be.

에 배치되며, 여기서 n = 1, 2이다. 이어서, (z = +h에서의) 물리 전하와 (z' = -D에서의) 그의 이미지의 중첩을 이용하여 그라운드(z≥0) 위의 장들을 계산할 수 있다. 복소 깊이들에서의 전하 이미지들(Q₁', Q₂')은 위의 식 20 및 21에서 지정된 원하는 전류 위상들의 획득을 실제로 돕는다.In addition, the complex spacing of image charges (Q ₁ ', Q ₂ ') implies that the outer fields will experience additional phase shifts that are not experienced when the interface is a lossless dielectric or a perfect conductor. Loss Dielectric Imagery - The essence of the theory is to replace the finite conduction earth (or lossy dielectric) with a perfect conductor at a complex depth z = -d / 2. Subsequently, the source image has a complex depth

, Where n = 1, 2. Then, the superposition of the image (at z = + h) and its image (at z '= -D) can be used to calculate the fields on ground (z≥0). The charge images (Q ₁ ', Q ₂ ') at the complex depths actually help in obtaining the desired current phases specified in equations 20 and 21 above.

에 대한

의 비율은 아래 식에 의해 주어진다는 점에 유의한다.From the above equations 2 and 3,

For

Is given by the following formula.

Also, asymptotically,

.

As a result, from equations 2 and 3,

.

는 복소 브루스터 각도이다. 소스 분포들을 조정하고, 손실 전도 매체(203)의 표면에서 복소 브루스터 각도 조명을 합성함으로써, 제넥 표면파들이 여기될 수 있다.here

Is the complex Brewster angle. By adjusting the source distributions and synthesizing the complex Brewster angle illumination at the surface of the lossy conduction medium 203, the generator surface waves can be excited.

Referring to Fig. 5, an incident field E polarized parallel to the plane of incidence is shown. The electric field vector E should be synthesized as an incoming nonuniform plane wave polarized parallel to the plane of incidence. The electric field vector E can be generated from independent horizontal and vertical components as follows.

Geometrically, the drawing of FIG. 5 implies the following.

&Lt; EMI ID =

Equation (38b)

This means that the length ratio is as follows.

However, recalling from Eq. 36,

인 것을 필요로 하며, 이는 다음을 유발한다.Thus, for a generic surface wave,

, Which leads to the following.

,

) 간의 상대 위상을 제어하는 경우에, 합성된 E 장 벡터가 복소 브루스터 각도로 유효하게 입사될 것이라는 것을 의미한다. 그러한 상황은 영역 1과 영역 2 사이의 인터페이스 위에 제넥 표면파를 합성적으로 여기할 것이다.These equations show the magnitude of the complex field ratio and the incident vertical and horizontal components in a plane parallel to the plane of incidence

,

), It means that the synthesized E-field vector will be effectively incident at a complex Brewster angle. Such a situation would synthetically excite the top surface wave on the interface between region 1 and region 2.

Referring now to Fig. 6, there is shown another view of a polyphase waveguide probe 200 disposed on a lossy conduction medium 203, in accordance with one embodiment of the present disclosure. The loss conduction medium 203 constitutes region 1 (FIG. 2) in accordance with one embodiment. In addition, the second medium 206 shares a boundary interface with the lossy conduction medium 203 and constitutes a region 2 (Fig. 2).

According to one embodiment, the lossy conduction medium 203 comprises a ground medium, such as a planet earth. For this reason, such surface media include all structures or formations contained thereon whether natural or artificial. For example, such surface media can include natural elements such as rocks, dirt, sand, fresh water, seawater, trees, plants, and all other natural elements that make up our planet. In addition, such surface media may include artificial elements such as concrete, asphalt, building materials and other artificial materials. In other embodiments, lossy conduction medium 203 may comprise any other medium than earth, regardless of whether it is spontaneous or artificial. In other embodiments, the lossy conduction medium 203 may include other media, such as artificial surfaces and structures such as automobiles, airplanes, artificial materials (such as plywood, plastic sheeting or other materials) or other media .

If the lossy conduction medium 203 comprises a ground medium or earth, the second medium 206 may comprise an atmosphere above ground. Thus, an atmosphere can be referred to as an "atmospheric medium" that includes the air and other elements that make up the Earth's atmosphere. In addition, it is possible that the second medium 206 may include other media relative to the lossy conduction medium 203.

The polyphase waveguide probe 200 includes a pair of charge terminals T ₁ and T ₂ . Although two charge terminals T ₁ and T _{2 are} shown, it is understood that there may be more charge terminals than two charge terminals T ₁ and T ₂ . According to one embodiment, the charge terminals T ₁ , T ₂ are disposed on the lossy conduction medium 203 along a vertical axis z perpendicular to the plane provided by the lossy conduction medium 203. In this regard, the charge terminal T ₁ is placed directly over the charge terminal T ₂ , but it is possible that some other arrangement of two or more charge terminals T _N can be used. According to various embodiments, charges (Q ₁ , Q ₂ ) may be imposed on each charge terminal (T ₁ , T ₂ ).

The charge terminals T ₁ and / or T ₂ ) may comprise any conducting mass capable of holding charge. The charge terminal T ₁ has a magnetic capacity C ₁ and the charge terminal T ₂ has a magnetic capacity C ₂ . The charge terminals T ₁ and / or T ₂ ) may comprise any shape, such as a sphere, disc, cylinder, cone, toroid, randomized formation or any other shape. It should also be noted that the charge terminals T ₁ , T ₂ do not have to be the same, can each have individual sizes and shapes, and can include different conductive materials. According to one embodiment, the shape of the charge terminal T ₁ is specified to hold substantially as much charge as possible. Ultimately, the field intensity of the generatrix surface wave launched by the polyphase waveguide probe 200 is directly proportional to the amount of charge on the terminal T ₁ .

이며, 여기서 r은 미터 단위의 구의 반경을 의미한다. 격리된 원반의 자기 용량은

이며, 여기서 r은 미터 단위의 원반의 반경을 의미한다.The charge terminals T ₁ and / or T ₂ ) are spheres or discs, the respective magnetic capacities (C ₁ , C ₂ ) can be calculated. For example, the isolated conductor volume has a magnetic capacity of

, Where r is the radius of the sphere in meters. The magnetic capacity of the isolated disc is

, Where r is the radius of the disk in meters.

로서 계산될 수 있다.Therefore, the charge terminal to the (T ₁₎ when the charge (Q ₁₎ is stored on a given voltage (V) applied to the self-capacitance (C ₁₎ and a charge terminal (T ₁₎ of charge storage (T ₁₎

Lt; / RTI &gt;

6, according to one embodiment, the polyphase waveguide probe 200 includes a probe coupling circuit 209 coupled to charge terminals T ₁ , T ₂ . Probe coupling circuit 209 is a charge terminals and facilitate the coupling of the excitation source (213) for (T _1, T _2), the charge terminals for a given operating frequency (T _1, T ₂₎ each of the voltage level on the And the generation of phase. When more than two charge terminals T _N are used, the probe coupling circuit 209 will be configured to facilitate the generation of various voltage magnitudes and phases on each charge terminal T _{N to} each other. In the embodiment of the polyphase waveguide probe 200, the probe coupling circuit 209 includes various circuit configurations as described.

In one embodiment, the probe coupling circuit 209 is designated to cause the polyphase waveguide probe 200 to be electrically half-resonant. This places a -V on the first terminal (T ₁ or T ₂₎ the second terminal of the voltage + V on and the charge terminal (T ₁ or T ₂₎ of the terminal at any given time. In such a case, the voltages on the respective charge terminals T ₁ , T ₂ are 180 degrees out of phase, as can be appreciated. When the voltages on the respective charge terminals T ₁ and T ₂ are different in phase by 180 degrees, they experience the largest voltage magnitude difference on the charge terminals T ₁ and T ₂ . Alternatively, the probe coupling circuit 209 may be configured such that the phase difference between the charge terminals T ₁ and T ₂ is 180 degrees. For this purpose, the probe coupling circuit 209 can be adjusted to change voltage magnitudes and phases during adjustment of the polyphase waveguide probe 200.

A charge terminal (T ₂₎ directly due to the arrangement of the terminals of the charges (T ₁₎ above, the mutual capacitance (C _M) between the charge terminals (T _1, T ₂₎ is generated. Further, as described above, the charge terminal T ₁ has the magnetic capacity C ₁ , and the charge terminal T ₂ has the magnetic capacity C ₂ . Among the charge terminal according to the respective height of (T _1, T _2), the charge terminals (T ₁₎ and the bond between the loss-conducting medium 203, the capacity and the charge terminal (T ₂₎ and loss-conducting medium 203, Binding capacity may also be present. The mutual capacitance C _M depends on the distance between the charge terminals T ₁ and T ₂ .

이므로 전하 단자(T₁)와 관련된 자기 용량(C₁)에 비례하며, 여기서 V는 전하 단자(T₁) 상에 부과되는 전압이다.Ultimately, the field intensities produced by the polyphase waveguide probes 200 will be directly proportional to the magnitude of the charge Q ₁ imposed on the top terminal T ₁ . The charge (Q ₁ )

, Which is proportional to the capacitance C ₁ associated with the charge terminal T ₁ , where V is the voltage imposed on the charge terminal T ₁ .

According to one embodiment, an excitation source 213 is coupled to the probe coupling circuit 209 to apply a signal to the polyphase waveguide probe 200. The source 213 may be any suitable electrical source such as a voltage or current source capable of generating a voltage or current at an operating frequency applied to the multiphase waveguide probe 200. For this purpose, the excitation source 213 may comprise, for example, a generator, a functional generator, a transmitter or other electrical source.

In one embodiment, the excitation source 213 may be coupled to the polyphase waveguide probe 200 via magnetic coupling, capacitive coupling, or conduction coupling (direct tap) as described. In some embodiments, the probe coupling circuit 209 may be coupled to the lossy conduction medium 203. Also, in various embodiments, excitation source 213 may be coupled to lossy conduction medium 203 as described.

)이 매우 작거나 심지어는 무시 가능하다는 특성을 갖는다는 점에 유의해야 한다. 방사 저항(

)은 안테나로부터 궁극적으로 방사되는 것과 동일한 전력량을 방산하는 등가 저항이라는 것을 상기해야 한다. 다양한 실시예들에 따르면, 다상 도파관 프로브(200)는 유도 전자기파인 제넥 표면파를 런칭한다. 다양한 실시예들에 따르면, 본 명세서에서 설명되는 다상 도파관 프로브들은 방사 저항(

)을 거의 갖지 않는데, 그 이유는 그러한 다상 도파관 프로브들의 높이가 통상적으로 그들의 동작 파장들에 비해 작기 때문이다. 즉, 일 실시예에 따르면, 본 명세서에서 설명되는 다상 도파관 프로브들은 "전기적으로 작다". 본 명세서에서 고려되는 바와 같이, "전기적으로 작다"는 표현은 λ/2π와 동일한 반경을 갖는 구에 의해 구 형태로 둘러싸일 수 있는 본 명세서에서 설명되는 다상 도파관 프로브들의 다양한 실시예들과 같은 구조로서 정의되며, 여기서 λ는 자유 공간 파장이다. Fujimoto, K., A. Henderson, K. Hirasawa, and J.R. James, Small Antennas, Wiley, 1987, p. 4를 참고한다.In addition, according to one embodiment, the polyphase waveguide probe 200 described herein has its radiation resistance

) Is very small or even negligible. Radiation resistance

) Is an equivalent resistor that dissipates the same amount of power as the one that ultimately emits from the antenna. According to various embodiments, the polyphase waveguide probe 200 launches a generic surface wave that is an inductive electromagnetic wave. According to various embodiments, the polyphase waveguide probes described herein may be fabricated using radiation resistance

), Because the height of such polyphase waveguide probes is typically small relative to their operating wavelengths. That is, according to one embodiment, the polyphase waveguide probes described herein are "electrically small &quot;. As contemplated herein, the expression "electrically small" means that the same structure as the various embodiments of the polyphase waveguide probes described herein, which may be surrounded by a sphere having a radius equal to lambda / , Where lambda is the free space wavelength. Fujimoto, K., A. Henderson, K. Hirasawa, and JR James, Small Antennas, Wiley , 1987, p. 4.

)은 아래 식에 의해 표현될 수 있다.More specifically, the radiation resistance for a short unipolar antenna (

) Can be expressed by the following equation.

는 동작 주파수에서의 파장이다. Stutzman, W.L. et al., "Antenna Theory and Design," Wiley & Sons, 1981, p. 93을 참조한다.Here, if a short unipolar antenna has a height h with a uniform current distribution,

Is the wavelength at the operating frequency. Stutzman, WL et al., " Antenna Theory and Design ," Wiley & Sons, 1981, p. 93.

)의 값이

의 함수로서 결정되는 경우, 구조의 높이(h)가 동작 주파수에서의 동작 신호의 파장에 비해 작은 경우, 방사 저항(

)도 작을 것으로 추정된다. 일례로서, 송신 구조의 높이(h)가 동작 주파수에서의 동작 신호의 파장의 10%인 경우,

의 결과적인 값은 (.1)² = .01일 것이다. 따라서, 방사 저항(

)도 작을 것으로 추정될 것이다.Radiation resistance

) Is the value of

, When the height h of the structure is smaller than the wavelength of the operation signal at the operating frequency,

) Is also expected to be small. As an example, when the height h of the transmission structure is 10% of the wavelength of the operation signal at the operating frequency,

The resulting value of (1) ² = .01. Therefore, the radiation resistance

) Is expected to be small.

이하인 경우(

는 동작 주파수에서의 파장), 방사 저항(

)은 비교적 작을 것이다. 후술하는 다상 도파관 프로브들(200)의 다양한 실시예들에 대해, 송신 구조의 높이(h)는

로서 계산될 수 있으며, 여기서

은 전하 단자(T₁)의 높이이고,

는 전하 단자(T₂)의 높이이다. 본 명세서에서 설명되는 다상 도파관 프로브들(200)의 각각의 실시예에 대한 송신 구조의 높이(h)는 유사한 방식으로 결정될 수 있다는 것을 알아야 한다.Thus, according to various embodiments, the effective height h of the transmission structure is

Or less

The wavelength at the operating frequency), the radiation resistance (

) Will be relatively small. For various embodiments of the multi-phase waveguide probes 200 described below, the height h of the transmission structure is

, Where &lt; RTI ID = 0.0 &gt;

Is the height of the charge terminal T ₁ ,

Is the height of the charge terminal T ₂ . It should be noted that the height h of the transmission structure for each embodiment of the multiphase waveguide probes 200 described herein can be determined in a similar manner.

가 하나의 벤치마크로서 제공되지만, 동작 주파수에서의 동작 신호의 파장에 대한 송신 구조의 높이(h)의 비율은 임의 값일 수 있다는 것을 이해한다. 그러나, 주어진 동작 주파수에서, 주어진 송신 구조의 높이가 증가할 때, 그에 따라 방사 저항(

)도 증가할 것이라는 것을 이해한다.

Is provided as a benchmark, it is understood that the ratio of the height h of the transmission structure to the wavelength of the operating signal at the operating frequency can be any value. However, at a given operating frequency, as the height of a given transmission structure increases, the radiation resistance

) Will also increase.

)은 제넥 표면파의 런칭과 더불어 소정량의 방사가 발생할 수 있게 하는 값을 가질 수 있는 것이 가능하다. 이 때문에, 다상 도파관 프로브(200)는 방사의 형태로 에너지가 거의 또는 전혀 손실되지 않는 것을 보증하기 위해 동작 주파수에서의 파장에 비해 짧은 높이(h)를 갖도록 구성될 수 있다.Depending on the actual values for the wavelength of the operating signal at the height and operating frequency, the radiation resistance

) Can have a value that allows a predetermined amount of radiation to occur along with the launch of the top surface wave. For this reason, the polyphase waveguide probe 200 may be configured to have a short height h relative to the wavelength at the operating frequency to ensure that little or no energy is lost in the form of radiation.

In addition, the arrangement of the charge reservoirs (T ₁ , T ₂ ) along the vertical axis z is the same as that in the _first embodiment of the present invention, Provides symmetry. The multiphase waveguide probe 200 is shown as having two charge storages T ₁ and T ₂ along a vertical axis z perpendicular to the plane constituting the surface of the lossy conduction medium 203 but also provides the desired symmetry &Lt; / RTI &gt; For example, additional charge tanks T _N may be arranged along the vertical axis z, or any other arrangement may be used. In some embodiments, the symmetry of the transmission may not be necessary. In such a case, the charge tanks T _N may be arranged in a different configuration than along the vertical axis z to provide an alternative transmission distribution pattern.

When properly adjusted to operate at a predefined operating frequency, the polyphase waveguide probe 200 generates a surface-acoustic-wave along the surface of the lossy conduction medium 203. To this end, an excitation source 213 may be used to generate a predefined frequency of electrical energy applied to the polyphase waveguide probe 200 to excite the structure. The energy from the excitation source 213 is coupled to the lossy conduction medium 203 or transmitted by the polyphase waveguide probe 200 to one or more receivers located within the effective transmission range of the polyphase waveguide probe 200 in the form of surface- do. Thus, the energy is conveyed in the form of a surface waveguide mode or a generic surface wave which is an induced electromagnetic field. With regard to modern power grids that use high voltage lines, the surface wave of the first order includes the transmission line mode.

Therefore, the generator surface wave generated by the polyphase waveguide probe 200 is an induction wave, not a radiation wave, depending on the meaning of the terms described above. Surface wave is launched due to the fact that the multiphase waveguide probe 200 creates electromagnetic fields that are substantially mode-matched to the first-order surface-acoustic-wave mode on the surface of the lossy conduction medium 203. When the electromagnetic fields generated by the polyphase waveguide probe 200 are substantially mode-matched such that the electromagnetic fields substantially combine the wavefront incident at a complex Brewster angle of the lossy conduction medium 203 that causes little or no reflection do. Note that if the polyphase waveguide probe 200 is not substantially mode matched to the generator surface wave mode, then the complex Brewster angle of the lossy conduction medium 203 would not have been achieved and the generator surface wave would not be launched.

) 및 전도율(

)에 의존할 것이다. 따라서, 위의 식 20-23의 핸켈 함수들의 위상은 런칭 장소에서의 이러한 구성 파라미터들에 그리고 동작 주파수에 의존한다.When the lossy conduction medium 203 comprises a ground medium such as earth, the first surface acoustic wave mode may have a dielectric constant at a location where the polyphase waveguide probe 200 is located, as indicated in equations 1-11 above

) And conductivity

). Thus, the phase of the Hankel functions of equations 20-23 above depends on these configuration parameters at the launch location and on the operating frequency.

To excite fields associated with the generator surface wave mode, according to one embodiment, the multiphase waveguide probe 200 substantially combines the radial surface current density on the lossy conduction medium of the generator surface wave as represented by Equation 20 above . When this occurs, the electromagnetic fields are substantially or substantially mode matched to the first surface-acoustic-wave mode on the surface of the lossy conduction medium 203. Because of this, the match must be as closely as practicable. According to one embodiment, such a generator surface wave mode in which the electromagnetic fields are substantially matched is expressed in Equations 21-23 described above.

In order to synthesize the radial surface current density in the lossy conduction medium in the surface acoustic wave mode of the generator, the electrical properties of the polyphase waveguide probe 200 are determined for the given operating frequency and for the charge terminals (T ₁ , T ₂ ) to apply appropriate voltage magnitudes and phases. If more than two charge terminals T _N are used, then appropriate voltage magnitudes and phases are required to be imposed for each charge terminal T _N , where N is a very high It may be a large number.

및 영역 1의 유전율

에 대한 값들에 기초하여 위의 식 1-11을 이용하여 유도 장 강도 곡선(103)(도 1)이 생성될 수 있다. 그러한 유도 장 강도 곡선(103)은 최적 송신이 달성되었는지를 결정하기 위해 측정된 장 강도들이 유도 장 강도 곡선(103)에 의해 지시되는 크기들과 비교될 수 있게 하는 동작에 대한 벤치마크를 제공할 것이다.An iterative approach may be used to obtain appropriate voltage magnitudes and phases for a given design of the polyphase waveguide probe 200 at a given location. Them in detail, the terminals to determine the generated radial surface current density of the supplied current for the (T _1, T _2), the charge terminals (T _1, T ₂₎ charge and the loss-conducting medium 203, on the An analysis of a given excitation and configuration of the polyphase waveguide probe 200 can be performed considering the images. This process can be repeatedly performed until an optimal configuration and excitation for a given polyphase waveguide probe 200 is determined based on desired parameters. To help determine whether a given polyphase waveguide probe 200 is operating at an optimal level, the conductivity of region 1 at the location of the polyphase waveguide probe 200

And the dielectric constant of region 1

The derivative field strength curve 103 (Fig. 1) can be generated using the above Equations 1-11 based on the values for the derivative field strength curve 103 (Fig. Such an indicator field strength curve 103 provides a benchmark for the operation so that the field intensities measured can be compared to the magnitudes indicated by the indicator field strength curve 103 to determine whether optimal transmission has been achieved will be.

Various parameters related to the polyphase waveguide probe 200 can be adjusted to reach the optimized polyphase waveguide probe 200. [ That is, various parameters associated with the polyphase waveguide probe 200 may be modified to adjust the polyphase waveguide probe 200 to the desired operating configuration.

One parameter that can be changed to adjust the polyphase waveguide probe 200 is the charge terminals T ₁ and / or T ₂ to the surface of the lossy conduction medium 203 T ₂ ), or both. In addition, the distance or spacing between the charge terminals (T ₁ , T ₂ ) can also be adjusted. The mutual capacitance C _M or any coupling capacitances between the charge terminals T ₁ , T ₂ and the lossy conduction medium 203 can be minimized or otherwise altered as can be appreciated.

Alternatively, other parameters that can be adjusted are the respective charge terminals T ₁ and / or T ₂ ). As can be appreciated, the charge terminals T ₁ and / or T ₂ ), by changing the magnitude of each of the magnetic capacities C ₁ and / or C ₂ and mutual capacity C _M will change. In addition, any coupling capacities present between the charge terminals (T ₁ , T ₂ ) and the lossy conduction medium 203 will change. Thus, the voltage magnitudes and phases for the charge terminals T ₁ , T ₂ are changed.

Another parameter that can be adjusted is the probe coupling circuit 209 associated with the polyphase waveguide probe 200. This can be achieved by adjusting the size of the inductive and / or capacitive reactances that make up the probe coupling circuit 209. For example, if such inductive reactances include coils, the number of windings on such coils can be adjusted. Ultimately, adjustment to the probe coupling circuit 209 can be done to alter the electrical length of the probe coupling circuit 209 to affect the voltage magnitudes and phases for the charge terminals T ₁ , T ₂ have.

The frequency of the excitation source 213 applied to the polyphase waveguide probe 200 may be adjusted to optimize the transmission of the surface acoustic wave. However, if it is desired to transmit at a given frequency, it may be necessary to adjust other parameters to optimize the transmission.

It should be noted that the repetition of transmission performed by performing various adjustments may be implemented using computer models or by adjusting physical structures as can be appreciated. In one approach, a long meter tuned to the transmit frequency can be placed at an appropriate distance from the polyphase waveguide probe 200, and until a maximum or any other desired field strength of the resulting generatrix surface wave is detected, as described above, Adjustments can be made. Thus, the field strength can be compared to the induced field strength curve 103 (FIG. 1) generated at the desired operating frequency and voltages for the terminals T ₁ , T ₂ . According to one approach, a suitable distance for the placement of such an instrument may be specified to be greater than the transition region 216 (FIG. 4) within the aforementioned "far " region where the surface current J ₂ predominates.

)을 근사화하는 대응하는 "가까운" 표면 전류(J₁) 및 "먼" 표면 전류(J₂)를 생성할 수 있다. 따라서, 결과적인 전자기장들은 손실 전도 매체(203)의 표면 상의 제넥 표면파 모드에 실질적으로 또는 대략 모드 매칭될 것이다.By making the above adjustments, the same currents of the first-order surface-acoustic-wave mode specified in Equations 17 and 18

Quot; near "surface current J ₁ and a" far "surface current J ₂ that approximate the surface current J ₂ . Thus, the resulting electromagnetic fields will be substantially or approximately mode matched to the first surface-acoustic-wave mode on the surface of the lossy conduction medium 203.

Referring now to Figures 7A-7J, additional examples of polyphase waveguide probes 200 are shown, here represented by polyphase waveguide probes 200a-j, in accordance with various embodiments of the present disclosure. Each of the polyphase waveguide probes 200a-j includes a different probe coupling circuit 209, here represented as probe coupling circuits 209a-j, in accordance with various embodiments. Although several examples of probe coupling circuits 209a-j are described, these embodiments are illustrative only and are not intended to limit the scope of the present invention to those embodiments in which charge terminals T ₁ and T ₂ are used in accordance with the principles described herein to facilitate launching of surface- It will be appreciated that there may be many other probe coupling circuits 209 that are not described herein that can be used to provide the desired voltage magnitudes and phases for the probe.

In addition, each of the probe coupling circuits 209a-j may utilize inductive impedances including coils, but are not limited thereto. Coils are used, but it is understood that other circuit elements that are not only concentrated but also may be used as reactances. In addition, other circuit elements than those described herein may be included within the probe coupling circuits 209a-j. In addition, it should be noted that various polyphase waveguide probes 200a-j with their respective probe coupling circuits 209a-j are only described herein to provide examples. For this reason, there may be a variety of different probe interconnecting circuits 209 that may be used to launch the surface wave generations in accordance with the various principles described herein and many other polyphase waveguide probes 200 that utilize other circuitry.

Referring now to FIG. 7A, a first example of a polyphase waveguide probe 200 (FIG. 6) is depicted herein as a polyphase waveguide probe 200a, according to one embodiment. The multiphase waveguide probe 200a includes charge terminals T ₁ and T ₂ disposed along a vertical axis z substantially perpendicular to the plane provided by the lossy conduction medium 203. The second medium 206 is located above the lossy conduction medium 203. The charge terminal T ₁ has a magnetic capacity C ₁ and the charge terminal T ₂ has a magnetic capacity C ₂ . During operation, charges Q ₁ , Q ₂ are imposed on each of the charge terminals T ₁ , T ₂ in accordance with the voltages applied to the charge terminals T ₁ , T ₂ at any given moment . Between the charge terminals T ₁ and T ₂ there may be a mutual capacitance C _M depending on the distance therebetween. In addition, coupling capacitances exist between the respective charge terminals (T ₁ , T ₂ ) and the lossy conduction medium 203 depending on the heights of the respective charge terminals (T ₁ , T ₂ ) for the lossy conduction medium 203 .

The polyphase waveguide probe 200a includes a probe coupling circuit 209a including an inductive impedance including a coil L _1a having a pair of leads coupled to each of the charge terminals T ₁ and T ₂ . In one embodiment, the coil L _1a is designated to have an electrical length that is half (1/2) of the wavelength at the operating frequency of the polyphase waveguide probe 200a.

Although the electrical length of the coil L _1a is specified as about half (1/2) of the wavelength at the operating frequency, the coil L _1a can be specified to have an electrical length of different values. According to one embodiment, the fact that the coil L _1a has an electrical length of about half the wavelength at the operating frequency offers the advantage in that a maximum voltage difference is created on the charge terminals T ₁ , T ₂ . However, the length or diameter of the coil L _1a may increase or decrease when adjusting the polyphase waveguide probe 200a to obtain an optimal excitation of the first surface acoustic wave mode. Alternatively, the inductive impedance may be specified to have an electrical length much smaller or larger than one-half the wavelength at the operating frequency of the polyphase waveguide probe 200a.

According to one embodiment, the excitation source 213 is coupled to the probe coupling circuit 209 via magnetic coupling. Specifically, the excitation source 213 is coupled to a coil L _P that is inductively coupled to the coil L _1a . This can be done by link coupling, tapped coil, variable reactance or other coupling approach as can be appreciated. For this reason, as can be appreciated, the coil L _P acts as a primary coil and the coil L _la acts as a secondary coil.

The heights of the respective charge terminals T ₁ and T ₂ can be changed with respect to the lossy conduction medium 203 and with respect to each other in order to adjust the polyphase waveguide probe 200a for transmission of the desired genex surface wave. Also, the sizes of the charge terminals (T ₁ , T ₂ ) can be changed. In addition, the size of the coil (L _1a) may be varied by changing the dimensions of the predetermined other by addition or removal of winding or coil (L _1a).

Based on experiments on the multiphase waveguide probe 200a it appears to be easiest to adjust and operate the multiphase waveguide probes 200a-j to achieve the desired frequency.

Referring now to FIG. 7B, an example of a polyphase waveguide probe 200 (FIG. 6) is depicted herein as a polyphase waveguide probe 200b, according to one embodiment. The multiphase waveguide probe 200b includes charge terminals T ₁ and T ₂ disposed along a vertical axis z that is substantially perpendicular to the plane provided by the lossy conduction medium 203. The second medium 206 is located above the lossy conduction medium 203. The charge terminals T ₁ , T ₂ are disposed along the vertical axis z to provide cylindrical symmetry at the resulting surface-to-surface wave as described above. The charge terminal T ₁ has a magnetic capacity C ₁ and the charge terminal T ₂ has a magnetic capacity C ₂ . During operation, charges Q ₁ , Q ₂ are imposed on each of the charge terminals T ₁ , T ₂ in accordance with the voltages applied to the charge terminals T ₁ , T ₂ at any given moment . Between the charge terminals T ₁ and T ₂ there may be a mutual capacitance C _M depending on the distance therebetween. In addition, coupling capacitances exist between the respective charge terminals (T ₁ , T ₂ ) and the lossy conduction medium 203 depending on the heights of the respective charge terminals (T ₁ , T ₂ ) for the lossy conduction medium 203 .

The polyphase waveguide probe 200b also includes a probe coupling circuit 209b including a first coil L _1b and a second coil L _2b . The first coil L _1b is coupled to each of the charge terminals T ₁ and T ₂ as shown. The second coil L _2b is coupled to the charge conduction medium 203 and to the charge terminal T ₂ .

The excitation source 213 is magnetically coupled to the probe coupling circuit 209b in a manner similar to that described in connection with the polyphase waveguide probe 200a (FIG. 7A) described above. For this reason, the excitation source 213 is coupled to a coil L _P acting as a primary coil, and the coil L _1b acts as a secondary coil. Alternatively, the coil L _2b may act as a secondary coil.

The heights of the respective charge terminals T ₁ and T ₂ can be changed with respect to the lossy conduction medium 203 and with respect to each other to adjust the polyphase waveguide probe 200b for transmission of the desired top surface wave. Also, the sizes of the charge terminals (T ₁ , T ₂ ) can be changed. In addition, the coils (L _1b, L _2b) have respective size can be changed by adding or removing the winding or by changing some other dimension of each coil (L _1b, L _2b).

Referring now to FIG. 7C, another example of a polyphase waveguide probe 200 (FIG. 6) is depicted herein as a polyphase waveguide probe 200c, according to one embodiment. The multiphase waveguide probe 200c includes charge terminals T ₁ and T ₂ disposed along a vertical axis z substantially perpendicular to the plane provided by the lossy conduction medium 203. The second medium 206 is located above the lossy conduction medium 203. The charge terminal T ₁ has a magnetic capacity C ₁ and the charge terminal T ₂ has a magnetic capacity C ₂ . During operation, charges Q ₁ , Q ₂ are imposed on each of the charge terminals T ₁ , T ₂ in accordance with the voltages applied to the charge terminals T ₁ , T ₂ at any given moment . Between the charge terminals T ₁ and T ₂ there may be a mutual capacitance C _M depending on the distance therebetween. In addition, coupling capacitances exist between the respective charge terminals (T ₁ , T ₂ ) and the lossy conduction medium 203 depending on the heights of the respective charge terminals (T ₁ , T ₂ ) for the lossy conduction medium 203 .

The polyphase waveguide probe 200c also includes a probe coupling circuit 209c including a coil L _1c . One end of the coil L _1c is coupled to the charge terminal T ₁ as shown. The second end of the coil L _1c is coupled to the lossy conduction medium 203. A tab coupled to the charge terminal T ₂ is disposed along the coil L _1c .

The excitation source 213 is magnetically coupled to the probe coupling circuit 209c in a manner similar to that described in connection with the polyphase waveguide probe 200a (FIG. 7A) described above. For this reason, the excitation source 213 is coupled to a coil L _P acting as a primary coil, and the coil L _1c acts as a secondary coil. The coil L _P may be disposed at an arbitrary position along the coil L _1c .

The height of each of the charge terminals T ₁ and T ₂ can be changed with respect to the lossy conduction medium 203 and with respect to each other to adjust the polyphase waveguide probe 200c for excitation and transmission of the desired genex surface wave . Also, the sizes of the charge terminals (T ₁ , T ₂ ) can be changed. In addition, the size of the coil (L _1c) can be changed by changing the dimension of some other of the addition or removal of winding or coil (L _1c). In addition, the inductance provided by the portions of the coil _L1c above and below the tabs can be adjusted by moving the position of the tabs.

Referring now to FIG. 7D, another example of a polyphase waveguide probe 200 (FIG. 6) is depicted herein as a polyphase waveguide probe 200d, according to one embodiment. The multiphase waveguide probe 200d includes charge terminals T ₁ and T ₂ disposed along a vertical axis z substantially perpendicular to the plane provided by the lossy conduction medium 203. The second medium 206 is located above the lossy conduction medium 203. The charge terminal T ₁ has a magnetic capacity C ₁ and the charge terminal T ₂ has a magnetic capacity C ₂ . During operation, charges Q ₁ , Q ₂ are imposed on each of the charge terminals T ₁ , T ₂ in accordance with the voltages applied to the charge terminals T ₁ , T ₂ at any given moment . Between the charge terminals T ₁ and T ₂ there may be a mutual capacitance C _M depending on the distance therebetween. In addition, coupling capacitances exist between the respective charge terminals (T ₁ , T ₂ ) and the lossy conduction medium 203 depending on the heights of the respective charge terminals (T ₁ , T ₂ ) for the lossy conduction medium 203 .

The polyphase waveguide probe 200d also includes a probe coupling circuit 209d including a first coil L _1d and a second coil L _2d . The first lead is coupled to a charge terminal (T ₁₎ of the first coil (L _1d), the second lead of the first coil (L _1d) is coupled to a loss-conducting medium 203. The second coil and the first lead is coupled to a charge terminal (T ₂₎ of the (L _2d), the second lead of the second coil (L _2d) is coupled to a loss-conducting medium 203.

The excitation source 213 is magnetically coupled to the probe coupling circuit 209d in a manner similar to that described with respect to the polyphase waveguide probe 200a (Fig. 7A) described above. For this reason, the excitation source 213 is coupled to the coil L _P acting as a primary coil, and the coil L _2d acts as a secondary coil. Alternatively, the coil L _1d may act as a secondary coil.

The height of each of the charge terminals T ₁ and T ₂ may be varied with respect to the lossy conduction medium 203 and with respect to each other to adjust the polyphase waveguide probe 200d for excitation and transmission of the desired genex surface wave . Also, the sizes of the charge terminals (T ₁ , T ₂ ) can be changed. In addition, the coils (L _1d, L _2d) has a respective size can be changed by adding or removing the winding or by changing some other dimension of each coil (L _1d, L _2d).

Referring now to FIG. 7E, another example of a polyphase waveguide probe 200 (FIG. 6) is depicted herein as a polyphase waveguide probe 200e, according to one embodiment. The polyphase waveguide probe 200e includes charge terminals T ₁ and T ₂ disposed along a vertical axis z substantially perpendicular to the plane provided by the lossy conduction medium 203. The second medium 206 is located above the lossy conduction medium 203. The charge terminals T ₁ , T ₂ are disposed along the vertical axis z to provide cylindrical symmetry at the resulting surface-to-surface wave as described above. The charge terminal T ₁ has a magnetic capacity C ₁ and the charge terminal T ₂ has a magnetic capacity C ₂ . During operation, charges Q ₁ , Q ₂ are imposed on each of the charge terminals T ₁ , T ₂ in accordance with the voltages applied to the charge terminals T ₁ , T ₂ at any given moment . Between the charge terminals T ₁ and T ₂ there may be a mutual capacitance C _M depending on the distance therebetween. In addition, coupling capacitances exist between the respective charge terminals (T ₁ , T ₂ ) and the lossy conduction medium 203 depending on the heights of the respective charge terminals (T ₁ , T ₂ ) for the lossy conduction medium 203 .

The polyphase waveguide probe 200e also includes a probe coupling circuit 209e including a first coil L _1e and a resistor R ₂ . The first lead is coupled to a charge terminal (T ₁₎ of the first coil (L _1e), the second lead of the first coil (L _1e) is coupled to a loss-conducting medium 203. The first lead of the resistor R ₂ is coupled to the charge terminal T ₂ and the second lead of the resistor R ₂ is coupled to the lossy conduction medium 203.

The excitation source 213 is magnetically coupled to the probe coupling circuit 209e in a manner similar to that described in connection with the polyphase waveguide probe 200a (FIG. 7A) described above. For this reason, the excitation source 213 is coupled to a coil L _P acting as a primary coil, and the coil L _1e acts as a secondary coil.

The height of each of the charge terminals T ₁ and T ₂ can be changed with respect to the lossy conduction medium 203 and with respect to each other, in order to adjust the polyphase waveguide probe 200e for transmission of the desired top surface wave. Also, the sizes of the charge terminals (T ₁ , T ₂ ) can be changed. In addition, the size of the coil (L _1e) can be changed by adding or removing the winding or by changing some other dimension of each coil (L _1e). In addition, the magnitude of the resistor R ₂ can also be adjusted.

7f, a further example of a polyphase waveguide probe 200 (FIG. 6) is depicted herein as a multiphase waveguide probe 200f, according to one embodiment. The polyphase waveguide probe 200f includes a charge terminal T ₁ and a ground screen G serving as a second charge terminal. The charge terminal T ₁ and the ground screen G are disposed along a vertical axis z that is substantially perpendicular to the plane provided by the lossy conduction medium 203. The second medium 206 is located above the lossy conduction medium 203. The height H ₂ of the ground screen G is subtracted from the height H ₁ of the charge terminal T ₁ to calculate the height h of the transmission structure.

The charge terminal T ₁ has a magnetic capacity C ₁ and the ground screen G has a magnetic capacity C ₂ . During operation, the electric charge at any given moment the terminal (T ₁₎ and the charge on each charge terminal (T ₁₎ and the ground screen (G) depending on the voltages applied to the ground screen (G) (Q _1, Q ₂ ) Is imposed. A mutual capacitance C _M may exist between the charge terminal T ₁ and the ground screen G depending on the distance therebetween. In addition, between the charge terminal (T ₁ ) and / or the ground screen (G) and the lossy conduction medium (203), depending on the height of the charge terminal (T ₁ ) and the ground screen Coupling capacities can exist. Generally, there will be a coupling capacitance between the ground screen G and the lossy conduction medium 203 due to its proximity to the lossy conduction medium 203.

The polyphase waveguide probe 200f includes a probe coupling circuit 209f constituted by an inductive impedance including a coil L _1f having a pair of leads coupled to the charge terminal T ₁ and the ground screen G . In one embodiment, the coil L _1f is specified to have an electrical length that is half (1/2) of the wavelength at the operating frequency of the polyphase waveguide probe 200f.

The electrical length of the coil (L _1f), but is specified as about half (1/2) of the wavelength at the operating frequency, the coils (L _1f) will be understood that may be specified so as to have an electrical length of a different value. According to one embodiment, the fact that the coil L _1f has an electrical length of about half the wavelength at the operating frequency has the advantage that in that a maximum voltage difference is created on the charge terminal T ₁ and the ground screen G . However, the length or diameter of the coil L _1f may increase or decrease when adjusting the polyphase waveguide probe 200f to obtain an optimum transmission of the surface-wave-wise surface. Alternatively, the inductive impedance may be specified to have an electrical length much smaller or larger than one-half the wavelength at the operating frequency of the polyphase waveguide probe 200f.

According to one embodiment, the excitation source 213 is coupled to the probe coupling circuit 209f via magnetic coupling. Specifically, the excitation source 213 is coupled to a coil L _P that is inductively coupled to the coil L _1f . This can be done by link combining, uplink / join networks or other approaches as can be appreciated. For this reason, as can be appreciated, the coil L _P acts as a primary coil and the coil L _1f acts as a secondary coil.

The height of each of the charge terminals T ₁ and T ₂ may be changed with respect to the lossy conduction medium 203 and with respect to each other to adjust the polyphase waveguide probe 200a for launching and transmitting the desired genex surface wave . Also, the sizes of the charge terminals (T ₁ , T ₂ ) can be changed. In addition, the size of the coil (L _1f) can be changed by changing the dimension of some other of the addition or removal of winding or coil (L _1f).

Referring now to FIG. 7G, another example of a polyphase waveguide probe 200 (FIG. 6) is depicted herein as a polyphase waveguide probe 200g, according to one embodiment. The multiphase waveguide probe 200g includes charge terminals T ₁ and T ₂ disposed along a vertical axis z substantially perpendicular to the plane provided by the lossy conduction medium 203. The second medium 206 is located above the lossy conduction medium 203. The charge terminals T ₁ , T ₂ are arranged along the vertical axis z to provide cylindrical symmetry at the resulting surface-to-surface wave, as described above. The charge terminal T ₁ has a magnetic capacity C ₁ and the charge terminal T ₂ has a magnetic capacity C ₂ . During operation, charges Q ₁ , Q ₂ are imposed on each of the charge terminals T ₁ , T ₂ in accordance with the voltages applied to the charge terminals T ₁ , T ₂ at any given moment . Between the charge terminals T ₁ and T ₂ there may be a mutual capacitance C _M depending on the distance therebetween. In addition, coupling capacitances exist between the respective charge terminals (T ₁ , T ₂ ) and the lossy conduction medium 203 depending on the heights of the respective charge terminals (T ₁ , T ₂ ) for the lossy conduction medium 203 .

The polyphase waveguide probe 200g also includes a probe coupling circuit 209g including a first coil L _1g , a second coil L _2g and a variable capacitor C _V. The first coil L _1g is coupled to each of the charge terminals T ₁ and T ₂ as shown. The second coil L _2g has a first lead coupled to the variable capacitor C _V and a second lead coupled to the lossy conduction medium 203. The variable capacitor C _V is also coupled to the charge terminal T ₂ and the first coil L _1g .

The excitation source 213 is magnetically coupled to the probe coupling circuit 209g in a manner similar to that described in connection with the polyphase waveguide probe 200a (FIG. 7A) described above. For this reason, the excitation source 213 is coupled to a coil L _p acting as a primary coil, and the coil L _1g or the coil L _2g can act as a secondary coil.

The height of each charge terminal T ₁ , T ₂ may be varied with respect to the lossy conduction medium 203 and with respect to each other, to adjust the polyphase waveguide probe 200g for launching and transmitting the desired genex surface wave . Also, the sizes of the charge terminals (T ₁ , T ₂ ) can be changed. In addition, the respective sizes of the coils (L _{1g, 2g} L) can be changed by adding or removing the winding or by changing some other dimension of each coil (L _{1g, 2g} L). In addition, the variable capacitance C _V can be adjusted.

Referring now to FIG. 7H, another example of a polyphase waveguide probe 200 (FIG. 6) is depicted herein as a polyphase waveguide probe 200h, according to one embodiment. The multiphase waveguide probe 200h includes charge terminals T ₁ and T ₂ disposed along a vertical axis z substantially perpendicular to the plane provided by the lossy conduction medium 203. The second medium 206 is located above the lossy conduction medium 203. The charge terminal T ₁ has a magnetic capacity C ₁ and the charge terminal T ₂ has a magnetic capacity C ₂ . During operation, charges Q ₁ , Q ₂ are imposed on each of the charge terminals T ₁ , T ₂ in accordance with the voltages applied to the charge terminals T ₁ , T ₂ at any given moment . Between the charge terminals T ₁ and T ₂ there may be a mutual capacitance C _M depending on the distance therebetween. In addition, coupling capacitances exist between the respective charge terminals (T ₁ , T ₂ ) and the lossy conduction medium 203 depending on the heights of the respective charge terminals (T ₁ , T ₂ ) for the lossy conduction medium 203 .

The polyphase waveguide probe 200h also includes a probe coupling circuit 209h including a first coil L _1h and a second coil L _2h . The first lead is coupled to a charge terminal (T ₁₎ of the first coil (L _1h), a second lead of the first coil (L _1h) is coupled to a second charge terminal (T _2). The first lead is coupled to a terminal (T _T) of the second coil (L _2h), the second lead of the second coil (L _2h) is coupled to a loss-conducting medium 203. Terminal (T _T) is disposed on the charge terminal (T ₂₎ and the terminal (T _T) a charge terminal (T ₂₎ such that the presence coupling capacitance (C _C) in between.

The excitation source 213 is magnetically coupled to the probe coupling circuit 209h in a manner similar to that described in connection with the polyphase waveguide probe 200a (FIG. 7A) described above. For this reason, the excitation source 213 is coupled to the coil L _P acting as a primary coil, and the coil L _2h acts as a secondary coil. Alternatively, the coil L _1h may act as a secondary coil.

The height of each of the charge terminals T ₁ and T ₂ can be changed with respect to the lossy conduction medium 203 and with respect to each other to adjust the polyphase waveguide probe 200h for launching and transmitting the desired genex surface wave . Also, the sizes of the charge terminals (T ₁ , T ₂ ) can be changed. In addition, the respective sizes of the coils (L _1h, L _2h) can be changed by adding or removing the winding or by changing some other dimension of each coil (L _1h, L _2h). In addition, the distance between the charge terminal T ₂ and the terminal T _T can be changed, and therefore the coupling capacitance C _C can be changed as can be appreciated.

Referring now to FIG. 7i, another example of a polyphase waveguide probe 200 (FIG. 6) is depicted herein as a polyphase waveguide probe 200i, according to one embodiment. The polyphase waveguide probe 200i is very similar to the polyphase waveguide probe 200h (Fig. 7H) except that the excitation source 213 is coupled in series with the probe coupling circuit 209i as described.

To this end, the polyphase waveguide probe 200i comprises charge terminals T ₁ , T ₂ disposed along a vertical axis z substantially perpendicular to the plane provided by the lossy conduction medium 203. The second medium 206 is located above the lossy conduction medium 203. The charge terminal T ₁ has a magnetic capacity C ₁ and the charge terminal T ₂ has a magnetic capacity C ₂ . During operation, charges Q ₁ , Q ₂ are imposed on each of the charge terminals T ₁ , T ₂ in accordance with the voltages applied to the charge terminals T ₁ , T ₂ at any given moment . Between the charge terminals T ₁ and T ₂ there may be a mutual capacitance C _M depending on the distance therebetween. In addition, coupling capacitances exist between the respective charge terminals (T ₁ , T ₂ ) and the lossy conduction medium 203 depending on the heights of the respective charge terminals (T ₁ , T ₂ ) for the lossy conduction medium 203 .

The polyphase waveguide probe 200i also includes a probe coupling circuit 209i including a first coil L _1i and a second coil L _2i . The first lead is coupled to a charge terminal (T ₁₎ of the first coil (L _1i), a second lead of the first coil (L _1i) is coupled to a second charge terminal (T _2). The first lead is coupled to a terminal (T _T) of the second coil (L _2i), the second lead of the second coil (L _2i) is coupled to the output of an excitation source (213). Also, the ground lead of the excitation source 213 is coupled to the lossy conduction medium 203. Terminal (T _T) is disposed on the charge terminal (T ₂₎ and the terminal (T _T) a charge terminal (T ₂₎ such that the presence coupling capacitance (C _C) in between.

The polyphase waveguide probe 200i provides an example in which the excitation source 213 is coupled in series to the probe coupling circuit 209i as described above. Specifically, the excitation source 213 is coupled between the coil L _2i and the lossy conduction medium 203.

The height of each of the charge terminals T ₁ and T ₂ may be varied with respect to the lossy conduction medium 203 and with respect to each other to adjust the polyphase waveguide probe 200i for launching and transmitting the desired genex surface wave . Also, the sizes of the charge terminals (T ₁ , T ₂ ) can be changed. In addition, the respective sizes of the coils (L _{1i, 2i} L) can be changed by adding or removing the winding or by changing some other dimension of each coil (L _1i, L _2i). In addition, the distance between the charge terminal T ₂ and the terminal T _T can be changed, and therefore the coupling capacitance C _C can be changed as can be appreciated.

Referring now to FIG. 7J, an example of a polyphase waveguide probe 200 (FIG. 6) is depicted herein as a polyphase waveguide probe 200j, in accordance with one embodiment. The multiphase waveguide probe 200j includes charge terminals T ₁ and T ₂ disposed along a vertical axis z substantially perpendicular to the plane provided by the lossy conduction medium 203. The second medium 206 is located above the lossy conduction medium 203. In this embodiment, the charge terminal T ₁ comprises a sphere and the charge terminal T ₂ comprises a disk. In this regard, the multiphase waveguide probe 200j provides an example that the charge terminals T _N may include any shape.

The charge terminal T ₁ has a magnetic capacity C ₁ and the charge terminal T ₂ has a magnetic capacity C ₂ . During operation, charges Q ₁ , Q ₂ are imposed on each of the charge terminals T ₁ , T ₂ in accordance with the voltages applied to the charge terminals T ₁ , T ₂ at any given moment . Between the charge terminals T ₁ and T ₂ there may be a mutual capacitance C _M depending on the distance therebetween. In addition, coupling capacitances exist between the respective charge terminals (T ₁ , T ₂ ) and the lossy conduction medium 203 depending on the heights of the respective charge terminals (T ₁ , T ₂ ) for the lossy conduction medium 203 .

The polyphase waveguide probe 200j includes a probe coupling circuit 209j including an inductive impedance including a coil L _1j having a pair of leads coupled to respective charge terminals T ₁ and T ₂ . In one embodiment, the coil L _1j is specified to have an electrical length that is half (1/2) of the wavelength at the operating frequency of the polyphase waveguide probe 200j. Although the electrical length of the coil L _1j is designated as about half (1/2) of the wavelength at the operating frequency, the coil L _1j is the same as that described with respect to the polyphase waveguide probe 200a (FIG. 7A) And can be specified to have electrical lengths of different values as well. In addition, the probe coupling circuit 209j includes a tab 223 on the coil L _1j coupled to the lossy conduction medium 203.

The excitation source 213 is magnetically coupled to the probe coupling circuit 209j in a manner similar to that described in connection with the above-described polyphase waveguide probe 200a (Fig. 7A). For this reason, the excitation source 213 is coupled to the coil L _P acting as a primary coil, and the coil L _1j acts as a secondary coil. The coil L _P may be disposed at an arbitrary position along the coil L _1j . Further, the coil L _P may be disposed above or below the tab 223.

The height of each of the charge terminals T ₁ and T ₂ can be changed with respect to the lossy conduction medium 203 and with respect to each other to adjust the polyphase waveguide probe 200j for launching and transmitting the desired genex surface wave . Also, the sizes of the charge terminals (T ₁ , T ₂ ) can be changed. In addition, the size of the coil (L _1j) can be changed by changing the dimension of some other of the addition or removal of winding or coil (L _1j). Further, the position of the tab 223 on the coil L _1j can be adjusted.

Referring to various embodiments of the multiphase waveguide probes 200a-j of FIGS. 7a-j, each of the multiphase waveguide probes 200a-j may be configured as a guided wave or waveguide mode along the surface of the lossy conduction medium 203 Lt; RTI ID = 0.0 &gt; energy. In order to facilitate such transfer, the multi-phase waveguide probes (200a-j) and each one desired voltage level for each of the charge terminal (T _1, T ₂₎ when the respective multi-phase waveguide probe (200a-j) being excited &Lt; / RTI &gt; and phases. Such excitation can occur by applying the energy from the excitation source 213 to each of the polyphase waveguide probes 200a-j as described above.

), 전도율(

) 및 잠재적으로 다른 파라미터들이 주어질 경우에 송신 위치에서 손실 전도 매체(203)의 유도 또는 제넥 표면 도파관 모드에 실질적으로 모드 매칭되는 장들을 실질적으로 합성하도록 조정될 수 있다. 표면 유도파의 도파관 모드는 전술한 식 21, 22 및 23에서 표현된다. 이러한 표면 도파관 모드는 식 20에서 미터당 암페어 단위로 표현되는 방사상 표면 전류 밀도를 갖는다.The voltage magnitudes and phases imposed on the charge terminals (T ₁ , T ₂ ) are dependent on the local permittivity of the lossy conduction medium 203

), Conductivity (

) And substantially match the mode-matched fields to the induction of the lossy conduction medium 203 at the transmission location or to the generator surface waveguide mode when potentially other parameters are given. The waveguide mode of the surface guided wave is expressed in Equations 21, 22 and 23 described above. This surface waveguide mode has a radial surface current density expressed in units of amperes per meter in Equation 20.

It is understood that it may be difficult to synthesize fields that exactly match the surface waveguide modes represented in equations 21, 22 and 23 described above. However, a surface acoustic wave can be launched when such sheets are at least similar to the surface waveguide mode. According to various embodiments, the fields are synthesized to match the surface waveguide mode within an acceptable engineering tolerance to launch a surface acoustic wave.

It may also be difficult to synthesize a radial surface current density that exactly matches the radial surface current density of the generator surface waveguide mode, and the synthesized radial surface current density is generated from the synthesized fields described above. According to various embodiments, the multiphase waveguide probes 200 can be adjusted to match the radial surface current density of the induced surface waveguide mode within an acceptable engineering tolerance to launch the first surface mode. By generating specific charge distributions and their images at the complex distances, the various polyphase waveguide probes 200a-j described above excite the surface currents - these chunks are designed to approximately match the propagating generator surface wave mode - Surface waves are launched. By virtue of this complex imaging technique inherent in the various polyphase waveguide probes 200a-j described above, the inductive interface can be substantially mode-matched to the surface waveguide modes desired to support in the transmit position. The inductive interface is the interface between region 1 (FIG. 2) and region 2 (FIG. 2) as described above. According to one embodiment, the guidance interface is the interface between the lossy conduction medium 203 and the standby medium provided by the earth as described above.

The voltage magnitudes and phases imposed on the charge terminals T ₁ and T ₂ and those at their complex depths and their effective images are transmitted to the neck surface of the lossy conduction medium 203 at the transmission location by the Leon- When adjusted to excite complex surface currents having fields that compose fields substantially matched with waveguide mode, such fields are automatically synthesized to substantially combine the wavefront incident at the complex Brewster angle of the lossy conduction medium 203 to produce a zero reflection Lt; / RTI &gt; This is a condition for wave matching at the boundary.

8A, 8B and 8C, examples of graphs 300a, 300b, and 300c illustrating field intensities in units of volts per meter as a function of distance in kilometers for purposes of comparison between generic surface waves and traditional radiation fields Are shown. In addition, the various graphs 300a, 300b and 300c show how the transmission distance of the surface acoustic wave varies with the transmission frequency.

및

의 구성 파라미터들이 계산의 목적을 위해 가정되었다. 아래의 표는 유도 장 강도 곡선(303a, 303b, 303c) 각각의 생성을 위해 가정된 다상 도파관 프로브 동작 파라미터들을 제공한다.Each of the graphs 300a, 300b and 300c shows corresponding induced field strength curves 303a, 303b and 303c and corresponding radiation field intensity curves 306a, 306b and 306c. Inductive field strength curves 303a, 303b, and 303c were generated assuming various parameters. Specifically, graphs 300a, 300b and 300c illustrate a constant charge Q ₁ (FIG. 3) applied to the top terminal T ₁ (FIG. 3) at frequencies of 10 MHz, 1 MHz and 0.1 MHz, respectively . From the R-3 map for Central Ohio published by the Federal Communications Commission (FCC)

And

Are assumed for the purpose of calculation. The table below provides assumed multiphase waveguide probe operating parameters for generation of each of the derivatized field strength curves 303a, 303b, and 303c.

로 지정되었지만, 10 MHz에서는 전류 분포를 균일하게 유지하기 위해 0.8 미터로 단축되었다. 또한, 단자(T₁)의 자기 용량(C₁)은 f = 0.1 MHz 및 1.0 MHz에서의 동작에 대해 100 pF로 설정되었다. 이 용량은 10 MHz에서의 사용에 대해서는 불합리하게 크며, 따라서 자기 용량(C₁)은 그 경우에 대해 감소되었다. 그러나, 장 강도에 대한 제어 파라미터인 결과적인 단자 전하(

)는 모든 3개의 유도 장 강도 곡선(303a, 303b, 303c)에 대해 동일한 값으로 유지되었다.In order to have a physically feasible operation, the height of the terminal T ₁ , for f = 0.1 MHz and 1.0 MHz,

, But at 10 MHz it was shortened to 0.8 meters to keep the current distribution uniform. Also, the capacitance C ₁ of the terminal T ₁ was set at 100 pF for operation at f = 0.1 MHz and 1.0 MHz. This capacity was unreasonably large for use at 10 MHz, and therefore the magnetic capacity (C ₁ ) was reduced for that case. However, the resulting terminal charge (the control parameter for the field strength

) Were maintained at the same values for all the three derivation field strength curves 303a, 303b and 303c.

From the graphs, it can be seen that the lower the frequency, the lower the attenuation and the longer the fields reach a greater distance. However, with energy conservation, the energy density decreases with distance. In other words, the higher the frequency, the smaller the area where the energy spreads, and thus the larger the energy density. Therefore, as the frequency increases, the range of the "knee" Alternatively, the lower the frequency, the lower the signal attenuation and the greater the field strength of the generator surface wave at very distant distances from the position of the transmission using the polyphase waveguide probe 200 (FIG. 6).

The surface acoustic waves for each case are identified asinduced field strength curves 303a, 303b and 303c, respectively. The Norton ground wave intensities in units of volts per meter for a short vertical monopole antenna having the same height as the respective polyphase waveguide probes 200 with the assumed ground loss of 10 ohms are respectively the radiation field intensity curves 306a, 306b, 306c ). This asserts a fairly realistic assumption for monopole antenna structures operating at these frequencies. Importantly, the appropriately mode-matched polyphase waveguide probes dramatically exceed the radiation field of any single pole at distances immediately preceding the "knee " in the induction field curves 303a-303c of each generator surface wave .

의 레이트로 떨어지는 반면, 방사 전자기장의 장 강도는 기하학적으로, 1/d에 비례하여 떨어지며, 여기서 d는 킬로미터 단위의 거리이다. 따라서, 유도 장 강도 곡선들(303a, 303b, 303c) 각각은 전술한 바와 같이 무릎을 특징으로 한다. 본 명세서에서 설명되는 다상 도파관 프로브의 송신 주파수가 감소함에 따라, 대응하는 유도 장 강도 곡선들(303a, 303b, 303c)의 무릎은 그래프에서 우측으로 밀릴 것이다.Given the above, according to one embodiment, the propagation distance of the surface acoustic wave changes as a function of the transmission frequency. Specifically, the lower the transmission frequency, the smaller the exponential decay of the surface acoustic wave, and therefore the surface acoustic wave will propagate farther. As described above, the field intensity of the induced surface wave is

While the field strength of the radiated electromagnetic field decreases geometrically in proportion to 1 / d, where d is the distance in kilometer units. Thus, each of the curve length curves 303a, 303b, and 303c is characterized by a knee as described above. As the transmit frequency of the polyphase waveguide probe described herein decreases, the knee of the corresponding derived field strength curves 303a, 303b, 303c will be pushed to the right in the graph.

8A shows an induced field intensity curve 303a and a field intensity curve 306a generated at a frequency of 10 MHz. As shown, the surface acoustic wave falls below 10 km. In Fig. 8B, the derivatized field strength curve 303b and the field strength curve 306b are generated at a frequency of 1 MHz. The induced field strength curve 303b falls at about 100 km. Finally, in Fig. 8C, the derivatized field strength curve 303c and the field strength curve 306c are generated at a frequency of 100 kHz (i.e., 1 Mhz). The induced field strength curve 303c falls between 4000-7000 km.

If the frequency is sufficiently low, it may be possible to transmit the surface acoustic wave across the earth. Such frequencies may be less than about 20-25 kHz. At such low frequencies, the lossy conduction medium 203 (Figure 6) ceases to be planar and becomes spherical. Thus, when the lossy conduction medium 203 comprises a terrestrial medium, the calculation of the guided field intensity curves will be altered to account for the sphere shape at low frequencies where the propagation distances approach the size of the terrestrial medium.

Given the above, some general derivation is provided in constructing the polyphase waveguide probe 200 (Fig. 6) using the terrestrial media as the lossy conduction medium 203 according to various embodiments. As a practical approach, it is possible to specify the operating frequency and to identify the desired field strength of the surface acoustic wave at a distance of interest from each polyphase waveguide probe 200 to be constructed.

) 및 전도율(

)을 획득하는 것이 필요할 것이다. 이러한 값들은 측정에 의해 또는 예를 들어 연방 통신 협회 또는 CCIR(Committee Consultif International Radio)에 의해 발행된 전도율 차트들을 참조하여 획득될 수 있다. 지정된 거리에서의 유전율(

), 전도율(

) 및 원하는 장 강도가 알려질 때, 위의 식 21-23에서 설명된 제넥의 정확한 표현들로부터의 장 강도의 직접 계산에 의해 필요한 전하(Q₁)가 결정될 수 있다.Given these parameters, then it is possible to determine the charge Q ₁ (FIG. 6) imposed on the top charge terminal T ₁ (FIG. 6) to produce the desired field strength at the specified distance. In order to determine the required charge (Q ₁ )

) And conductivity

). &Lt; / RTI &gt; These values may be obtained by measurement or by reference to conductivity charts issued by, for example, the Federal Communications Commission or Committee Consulting International Radio (CCIR). Permittivity at a specified distance (

), Conductivity (

) And the desired field intensities are known, the required charge (Q ₁ ) can be determined by direct calculation of field intensities from the precise representations of the neck neck described in Eqs. 21-23 above.

Then, when the required charge Q ₁ is determined, it is determined which magnitude C ₁ of the charge terminal T ₁ at a certain voltage V generates the required charge Q ₁ on the charge terminal T ₁ . The charge Q on any charge terminal T is calculated as Q = CV. In one approach, after choosing what is considered to be an allowable voltage (V) that may be placed on the charge terminal T ₁ , the charge (C ₁ ) required to achieve the required charge (Q ₁ ) It is possible to configure the terminal T ₁ . In an alternative, another approach can achieve what is by the specific configuration of the charge terminal (T ₁₎ self-capacitance (C ₁₎ After determining whether the resulting charge terminal necessary to increase the voltage (V) required for (T ₁₎ The charge (Q ₁ ) can be achieved.

In addition, there is a self-capacitance (C ₁₎ and a charge terminal of the operational bandwidth issues which must be considered when determining the voltage (V) to be imposed on the phase (T ₁₎ required for the charge terminal (T _1). Specifically, the bandwidth of the polyphase waveguide probes 200 described herein is relatively large. This provides a considerable degree of flexibility in specifying the capacitance C ₁ or voltage V as described above. However, as the magnetic capacitance C ₁ decreases and the voltage V increases, the bandwidth of the resulting polyphase waveguide probe 200 will decrease.

) 또는 전도율(

)의 작은 변화에 더 민감하게 할 수 있다. 그러한 유전율(

) 또는 전도율(

)의 변화는 계절 변화에 따른 기후의 변화로 인해 또는 비, 가뭄 및/또는 다른 국지적 날씨 변화의 시작과 같은 국지적 날씨 조건들의 변화로 인해 발생할 수 있다. 결과적으로, 일 실시예에 따르면, 전하 단자(T₁)는 실행 가능할 때 비교적 큰 자기 용량(C₁)을 갖도록 지정될 수 있다.Experimentally, the smaller magnetic capacity C ₁ allows a given polyphase waveguide probe 200 to have a dielectric constant of the earth at or near the transmitting location

) Or conductivity

Lt; RTI ID = 0.0 &gt; change. &Lt; / RTI &gt; Such permittivity (

) Or conductivity

) Can occur due to changes in weather due to seasonal changes or changes in local weather conditions such as rain, drought and / or the beginning of other local weather changes. As a result, according to one embodiment, the charge terminal T ₁ can be specified to have a relatively large capacitance C ₁ when practicable.

Subsequently, when the magnetic capacity C ₁ of the charge terminal T ₁ and the voltage to be imposed thereon are determined, the magnetic capacity C ₂ and the physical position of the second charge terminal T ₂ have to be determined. In reality, it has been found to self-capacitance (C ₂₎ of the charge terminal (T ₂₎ that is most easy to specify in the same manner as in the magnetic capacitor (C ₁₎ of the charge terminal (T _1). This may be accomplished by the size and shape of the charge terminal (T ₂₎ by the same manner as the size and shape of the charge terminal (T _1). This will ensure that the symmetry is maintained and will prevent the possibility of a specific phase shift between the two charge terminals (T ₁ , T ₂ ) which can adversely affect the match with the complex Brewster angle as described above. The fact that the same for self-capacitance (C _1, C ₂₎ the amount the charge terminals (T _1, T ₂₎ will result in the same voltage amplitude on the charge terminals (T _1, T _2). However, it is understood that the magnetic capacitances C ₁ , C ₂ may be different and the shape and size of the charge terminals T ₁ , T ₂ may be different.

To facilitate symmetry, the charge terminal T ₂ may be disposed directly below the charge terminal T ₁ along the vertical axis z (Fig. 6) as described above. Alternatively, it may be possible to arrange the charge terminal T ₂ at some other location so as to have a predetermined resultant effect.

The distance between the charge terminals T ₁ and T ₂ should be specified to provide the best match between the fields produced by the polyphase waveguide probe 200 and the induced surface waveguide mode at the transmit position. As a proposed starting point, this distance can be set so that the mutual capacitance _CM (FIG. 6) between the charge terminals T ₁ , T ₂ is less than or equal to the isolated capacitance C ₁ on the charge terminal T ₁ . Ultimately, the distance between the charge terminals T ₁ and T ₂ must be specified such that the mutual capacitance C _M is as small as possible. The mutual capacitance C _M can be determined by measurement and the charge terminals T ₁ , T ₂ can be arranged accordingly.

(도 7a-j)가 결정된다. 여기서, 소위 "이미지 복소 깊이" 현상이 발생한다. 이것은 전하들(Q₁, Q₂)을 갖는 전하 저장소들(T₁, T₂)로부터 그리고 높이(h)가 변할 때의 전하들(Q₁, Q₂)의 하위 표면 이미지들로부터 지구의 표면 상의 중첩 장들의 고려를 필요로 할 것이다. 주어진 다상 도파관 프로브(200)가 송신 위치에서 지구의 유도 표면 도파관 모드와 모드 매칭되는 것을 보증하기 위해 고려해야 할 상당한 수의 변수로 인해, 실질적인 시작점은 전하 단자들(T₁, T₂)과 관련된 용량이 각각 본질적으로 그들의 격리된 자기 용량(C₁, C₂)이 되도록 그라운드에 대한 전하 저장소들(T₁, T₂) 각각의 결합 용량이 무시될 수 있는 높이(h)이다.Then, the appropriate height of the polyphase waveguide probe 200

(Figs. 7A-J) are determined. Here, the so-called "image complex depth" phenomenon occurs. This is on the surface of the earth from the sub-surface images of the charge of (Q _1, Q ₂₎ of electric charges when the charge storage of (T _1, T ₂₎ and the height (h) from the change with (Q _1, Q ₂₎ You will need to consider overlapping chapters. Due to the considerable number of variables that must be considered to ensure that a given multiphase waveguide probe 200 is mode-matched to the earth's guided surface waveguide mode at the transmitting location, the actual starting point is the capacity associated with the charge terminals (T ₁ , T ₂ ) Is the height (h) at which the coupling capacitances of each of the charge reservoirs (T ₁ , T ₂ ) to ground can be ignored so that they are essentially their isolated capacitances (C ₁ , C ₂ ).

Another matter to consider when determining the height (h) associated with the polyphase waveguide probe 200 is whether radiation should be prevented. Specifically, as the height h of the polyphase waveguide probe 200 approaches a significant portion of the wavelength at the operating frequency, the radiation resistance R _r will increase secondarily with respect to height h, It will start to dominate the generation of the surface acoustic wave as described above. One of the benchmarks described above that ensures that the top surface wave dominates any radiation is to ensure that the height h is less than 10% of the wavelength at the operating frequency, but other benchmarks can be specified. In some instances, it may be desirable to allow a certain amount of radiation to occur in addition to launching the surface acoustic wave, so that the height h can be specified accordingly.

The probe coupling circuit 209 (FIG. 6) is then designated to provide a voltage phase between the charge terminals T ₁ and T ₂ . The voltage phase is shown to have a significant impact on generating mode-matched fields in the inductive surface waveguide mode at the transmit position. Assuming that the arrangement of the charge terminals T ₁ and T ₂ follows the vertical axis z to promote symmetry, the probe coupling circuit 209 has a voltage phase difference of 180 degrees on the charge terminals T ₁ and T ₂ . That is, the probe coupling circuit 209 is specified such that the voltage V on the charge terminal T ₁ is 180 degrees out of phase with the voltage on the charge terminal T ₂ .

As described above, one exemplary approach is to place a coil L _1a (FIG. 7A) between the charge terminals T ₁ and T ₂ as described above with respect to the multiphase waveguide probe 200a, Is to adjust the coil L _1a until the system is half-wave resonant electrically. This charge terminals (T _1, T ₂₎ the charge terminals so that the maximum voltage are arranged in 180-degree phase difference between the phase (T ₁₎ voltage to the voltage (V) on the charge and on the terminal (T ₂₎ (-V) .

The excitation source 213 (FIG. 6) may then be coupled to the probe coupling circuit 209 and the output voltage may be adjusted to achieve the voltage V necessary to provide the required charge Q ₁ , as described above . The excitation source 213 may be coupled to the probe coupling circuit 209 through magnetic coupling, capacitive coupling, or (direct) conduction coupling. It should be noted that the output of the source 213 here may be raised using a transformer if necessary or through some other approach. The position of the coil L _1a may be any position such as below the ground by the excitation source 213. Alternatively, in accordance with best RF practice, the coil L _1a may be placed directly between the charge reservoirs T ₁ and T ₂ . The principles of impedance matching can be applied when coupling the excitation source 213 to the probe coupling circuit 209.

Note that the phase difference does not necessarily have to be 180 degrees. For this purpose, the charge terminals T ₁ and / or T ₂₎ to one or both up or lowering of the charge terminal (T ₁ and / or T ₂ ) or by adjusting the probe coupling circuit 209 to adjust the voltage magnitudes and phases to generate fields that most closely match the induced surface waveguide mode to generate guided surface waves Respectively.

Experimental results

)은 0.0002 mhos/m이고, 테스트 위치에서의 그라운드의 유전율(

)은 5였다. 이러한 값들은 사용중인 주파수에서 인시투(in situ) 측정되었다.The above disclosures are supported by experimental results and data. Referring to FIG. 9, there is shown a graph depicting the measured field intensities of electromagnetic fields transmitted by one embodiment of an experimental polyphase waveguide probe measured on October 14, 2012 in Plymouth, New Hampshire. The transmission frequency was 59 MHz and a voltage of 60 mV was applied on the charge terminal (T ₁ ) of the experimental polyphase waveguide probe. The capacitance (C ₁ ) of the experimental polyphase waveguide probe was 8.5 pF. Conductivity of ground at test position (

) Is 0.0002 mhos / m, and the dielectric constant of the ground at the test position (

) Was 5. These values were measured in-situ at the frequency in use.

The graph includes an Inductance field intensity curve 400 indicated by a "Genek" curve at 80% efficiency and a Field strength curve 403 indicated by a "Norton" To this end, the field strength curve 403 represents the radiated electromagnetic fields produced by the quarter-wave monopole antenna operating at a frequency of 59 MHz. Circles 406 on the graph represent the measured strengths of the steel produced by the experimental polyphase waveguide probes. Field strength measurements were performed using a NIST-traceable Potomac Instruments FIM-71 commercial VHF field strength meter. As can be seen, the measured field intensities fall along the theoretical induction field intensity curve 400. These measured field intensities are consistent with the propagation of inductive or generic surface waves.

Referring now to FIG. 10, there is shown a graph illustrating the measured phase of electromagnetic waves transmitted from an experimental polyphase waveguide probe. Curve J (r) represents the phase of the field in accordance with the current (J _1, J _2), with the transition between the current as shown (J _1, J _2). Curve 503 indicates an asymptote representing the phase of current J ₁ and curve 506 indicates an asymptote indicating the phase of current J ₂ . Between the phases of the respective currents (J _1, J ₂₎ it is generally present from about 45 degrees difference. Circles 509 indicate measurements of the phase of current J (r) produced by an experimental polyphase waveguide probe operating at 59 MHz as in FIG. As shown, the circles 509 fall along the curve J (r), indicating that there is a transition of the phase of the current J (r) from the curve 503 to the curve 506. This indicates that the phase of the current J (r) produced by the experimental polyphase waveguide probe transitions from the phase produced by the near current (J ₁ ) to the current far from the phase (J ₂ ). Thus, these phase measurements coincide with the phase in the presence of induction or generic surface waves.

)은 15 정도였다. 이들은 사용중인 주파수에서 결정되었다.Referring to FIG. 11, there is shown a measurement of the field strength of an electromagnetic field transmitted by a second embodiment of an experimental polyphase waveguide probe, measured November 1, 2003, across the area of Lake Winnipesky near New Hampshire, A graph of the second set of data is shown. The transmission frequency was 1850 kHz and a voltage of 1250 V was applied on the charge terminal (T ₁ ) of the experimental polyphase waveguide probe. The experimental polyphase waveguide probe has a physical height of H ₁ = 2 meters. The capacitance (C ₁ ) of the experimental polyphase waveguide probe in this experiment, which is a flat conductive disc of 1 meter radius, was measured to be 70 pF. The polyphase waveguide probes were arranged as shown in Fig. 7J, with an interval h = 1 meter, and the height of the charge terminal T ₂ on the ground (lossy conduction medium 203) was H ₂ = 1 meter. The average conductivity (σ) of the ground near the experiment was 0.006 mhos / m and the relative dielectric constant of the ground

) Was about 15. These were determined at the frequency in use.

The graph is drawn by an experimental polyphase waveguide probe and shows an inductive field strength curve 600 indicated by the "third-order" curve at 85% efficiency, and a ground screen 600 consisting of 20 radial wires equally spaced and each having a length of 200 feet Quot; Norton "curve as emitted from a resonance monopolar of the same height H ₂ = 2 meters above it. For this reason, the field strength curve 603 represents a conventional Norton ground wave radiated from a conventional stub monopole antenna operating at a frequency of 1850 kHz above the lossy earth. Circles 606 on the graph represent the measured field intensities produced by the experimental polyphase waveguide probes.

및/또는 σ)은 경로-평균 구성 파라미터들로부터 상당히 벗어났을 가능성이 있다.As can be seen, the measured field intensities fall close to the theoretical Jennek induced field strength curve 600. A special mention of the field strength measured at r = 7 miles can be made. This intensities data was measured near the lake, which may explain the fact that the data is slightly off the theoretical Jenckian field intensity curve 600, i.e., the configuration parameters at that location

And / or &lt; RTI ID = 0.0 &gt; a) &lt; / RTI &gt;

The field strength measurements were performed using a NIST-traceable Potomac Instruments FIM-41 MF / HF field strength instrument. The measured field strength data is consistent with the presence of inductive or generic surface waves. From the experimental data, the measured field intensities observed at distances of less than 15 miles may not be due to the conventional Norton ground wave propagation, and may be attributable only to the induced surface wave propagation launched by the multiphase probe operating as described above It is obvious. Under the given 1.85 MHz experimental conditions, the Norton ground wave component ultimately outperforms the generic surface wave component beyond 20 miles.

Comparison of the measured GE surface wave data shown in Figure 9 at 59 MHz with the measured data in Figure 11 at 1.85 MHz shows the great advantage of using a polyphase waveguide probe according to various embodiments at lower frequencies .

의 고유 위상 부스트를 갖는 위상 전진 표면 전류를 유도하며, 그의 장들은 본 명세서에서 개시되는 바와 같이 손실 경계에 대한 복소 브루스터 각도에서 표면 조명을 합성한다는 것을 보여준다. 그 결과는 기하학적 확산으로 인해 1/d로서 감소하는 방사 장이 아니라

로서 감쇠하는 소실하는 단일 도체 방사상 송신선 모드로서 경계 표면에 의해 유도되는 원통 제넥-형태 파 전파의 효율적인 런칭이다.These experimental data show that the present polyphase waveguide probes comprising a plurality of properly phaseed and adjusted charge terminals, as taught herein,

Which leads to a phase forward surface current with a natural phase boost of &lt; RTI ID = 0.0 &gt; I, &lt; / RTI &gt; which shows that the surface illumination is synthesized at a complex Brewster angle to the loss boundary as described herein. The result is not a radiation field that decreases as 1 / d due to geometric spreading

Is an efficient launching of the cylindrical neck-shaped wave propagation induced by the boundary surface as a single-conductor radial transmission mode that attenuates as a result.

Referring now to Figures 12A, 12B and 13, examples of generalized receive circuits for using surface induced waves in wireless power delivery systems are shown. 12A and 12B illustrate a linear probe 703 and a tunable resonator 706. FIG. 13 shows magnetic coil 709 in accordance with various embodiments of the present disclosure. According to various embodiments, each of the linear probe 703, tunable resonator 706 and magnetic coil 709 may be formed in the form of a surface acoustic wave on the surface of the lossy conduction medium 203 (FIG. 6) Lt; / RTI &gt; may be used to receive power transmitted to the base station. As described above, in one embodiment, the lossy conduction medium 203 includes a ground medium.

12A, the open circuit terminal voltage at the output terminals 713 of the linear probe 703 depends on the effective height of the linear probe 703. [ For this reason, the terminal point voltage can be calculated as follows.

는 미터당 볼트 단위의 선형 프로브(703) 상의 벡터 내의 전기장의 강도이고,

은 선형 프로브(703)의 방향을 따른 적분의 요소이고,

는 선형 프로브(703)의 유효 높이이다. 전기 부하(716)가 임피던스 매칭 네트워크(719)를 통해 출력 단자들(713)에 결합된다.here,

Is the intensity of the electric field in the vector on linear probe 703 in volts per meter,

Is an element of integral along the direction of the linear probe 703,

Is the effective height of the linear probe 703. An electrical load 716 is coupled to the output terminals 713 via an impedance matching network 719.

When a surface acoustic wave as described above is applied to the linear probe 703, a voltage is generated across the output terminals 713 and this voltage is applied to the electric load (e. G., Via the conjugate impedance matching network 719) 716). To facilitate the flow of power to the electrical load 716, the electrical load 716 must be substantially impedance matched to the linear probe 703, as described below.

Referring to FIG. 12B, the tuned resonator 706 includes a charge terminal T _R raised above the lossy conduction medium 203. The charge terminal T _R has a magnetic capacity C _R. In addition, the loss-conducting medium 203, coupled between the charge port (T _R) with loss-conducting medium 203, according to the height of the charge terminal (T _R), above capacitor (not shown) may also be present. The coupling capacitance should preferably be minimized as much as possible, but this may not be entirely necessary in all cases of the polyphase waveguide probe 200.

The tunable resonator 706 also includes a coil L _R. One end of the coil (L _R) is coupled to the charge terminal (T _R), the other end of the coil (L _R) is coupled to a loss-conducting medium 203. Thus, a tuning resonator 706 (which may also be referred to as a tunable resonator (L _R -C _R )) includes a series tuned resonator as a charge terminal C _R , and the coil L _R is connected in series . The tuned resonator 706 is tuned by substantially eliminating the reactive impedance of the structure by adjusting the size and / or height of the charge terminal T _R and / or by adjusting the size of the coil L _R.

로서 계산된다. 튜닝형 공진기(706)의 전체 용량은 전하 단자(T_R)와 손실 전도 매체(203) 간의 용량도 포함할 수 있으며, 튜닝형 공진기(706)의 전체 용량은 인식될 수 있는 바와 같이 자기 용량(C_R) 및 임의의 결합 용량 양자로부터 계산될 수 있다. 일 실시예에 따르면, 전하 단자(T_R)는 임의의 결합 용량을 실질적으로 줄이거나 제거하기 위한 높이로 상승될 수 있다. 결합 용량의 존재는 전하 단자(T_R)와 손실 전도 매체(203) 간의 용량 측정들로부터 결정될 수 있다.For example, the reactance provided by the magnetic capacity (C _R )

. The total capacitance of the tuning resonator 706 may also include capacitance between the charge terminal T _R and the lossy conduction medium 203 and the total capacitance of the tuning resonator 706 may be the capacitance C _R ) and any coupling capacitances. According to one embodiment, the charge terminal T _R can be raised to a height to substantially reduce or eliminate any coupling capacitance. The presence of coupling capacitance can be determined from the capacitance measurements between the charge terminal (T _R ) and the lossy conduction medium (203).

로서 계산될 수 있으며, 여기서 L은 코일(L_R)의 집중 요소 인덕턴스(lumped-element inductance)이다. 코일(L_R)이 분산 요소인 경우, 그의 등가 단자점 유도 리액턴스는 전통적인 접근법들에 의해 결정될 수 있다. 튜닝형 공진기(706)를 튜닝하기 위해, 코일(L_R)에 의해 제공되는 유도 리액턴스가 튜닝형 공진기(706)에 의해 제공되는 용량 리액턴스와 동일하게 하여, 튜닝형 공진기(706)의 결과적인 순수 리액턴스가 동작 주파수에서 실질적으로 0이 되게 하도록 조정들이 행해질 것이다. 임피던스 매칭 네트워크(723)가 프로브 단자들(721)과 전기 부하(726) 사이에 삽입되어, 전기 부하(726)로의 최대 전력 전달을 위한 켤레 매치 조건을 달성할 수 있다.The inductive reactance provided by the individual element coils (L _R )

, Where L is the lumped-element inductance of the coil (L _R ). If the coil (L _R ) is a dispersive element, its equivalent terminal point induced reactance can be determined by conventional approaches. In order to tune the tuned resonator 706, the inductive reactance provided by the coil L _R is equal to the capacitive reactance provided by the tuning resonator 706, so that the resulting pure water of the tuned resonator 706 Adjustments will be made to make the reactance substantially zero at the operating frequency. An impedance matching network 723 may be inserted between the probe terminals 721 and the electrical load 726 to achieve a match match condition for maximum power delivery to the electrical load 726. [

As described above, in the presence of the surface acoustic wave generated at the frequencies of the tuning resonator 706 and the matching network 723, the maximum power will be transferred from the surface waveguide to the electrical load 726. [ That is, if a conjugate impedance match is established between the tuning resonator 706 and the electrical load 726, power will be transferred from the structure to the electrical load 726. For this reason, the electrical load 726 can be coupled to the tuning resonator 706 through magnetic coupling, capacitive coupling, or conduction (direct tap) coupling. The elements of the combined network may be concentrate components or distributed elements as can be appreciated. In the embodiment shown in FIG. 12B, magnetic coupling is used, where the coil L _S is arranged as a secondary coil with respect to the coil L _R acting as a transformer primary coil. The coil L _S can be linked to the coil L _R by recognizing it by geometrically winding it around the same core structure and adjusting the coupled magnetic flux. In addition, the tunable resonator 706 includes a series tunable resonator, but a parallel tunable resonator or even a dispersion-element resonator can be used.

)이 자기 코일(709)을 통과하여 자기 코일(709) 내에 전류를 유도하고 그의 출력 단자들(729)에서 단자점 전압을 생성하도록 배치될 수 있다. 단일 권선 코일에 결합된 유도 표면파의 자속은 다음과 같이 표현된다.13, magnetic coil 709 includes a receiving circuit coupled to an electrical load 736 via an impedance matching network 733. [ In order to facilitate reception and / or extraction of electric power from the surface acoustic wave, the magnetic coil 709 generates magnetic flux

May be arranged to pass through the magnetic coil 709 to induce a current in the magnetic coil 709 and to generate a terminal point voltage at its output terminals 729. [ The magnetic flux of the surface acoustic wave coupled to the single winding coil is expressed as follows.

는 결합 자속이고,

은 자기 코일(709)의 코어의 유효 상대 투자율이고,

은 자유 공간의 투자율이고,

는 입사 자기장 강도 벡터이고,

은 권선들의 교차 영역에 수직인 단위 벡터이고,

는 각각의 루프에 의해 둘러싸인 면적이다. 자기 코일(709)의 교차 영역에 걸쳐 균일한 입사 자기장에 대한 최대 결합을 위해 배향된 N-권선 자기 코일(709)의 경우, 자기 코일(709)의 출력 단자들(729)에서 나타나는 개방 회로 유도 전압은 다음과 같다.here,

Lt; / RTI &gt;

Is the effective relative permeability of the core of the magnetic coil 709,

Is the permeability of the free space,

Is an incident magnetic field strength vector,

Is a unit vector perpendicular to the intersection area of the windings,

Is the area surrounded by each loop. In the case of an N-winding magnetic coil 709 oriented for maximum coupling to a uniform incident magnetic field across the crossing region of the magnetic coil 709, the open circuit induction, which appears at the output terminals 729 of the magnetic coil 709, The voltage is as follows.

Here, the variables are defined above. The magnetic coil 709 is tuned to the induced wave frequency, optionally using an external capacitor across its output terminals 729 as a distributed resonator, and then coupled to the external electrical load 736 via the conjugate impedance matching network 733. [ ). &Lt; / RTI &gt;

The resultant circuit provided by the magnetic coil 709 and the electrical load 736 is appropriately adjusted through the impedance matching network 733 and is subjected to impedance matching using the current induced in the magnetic coil 709, The load 736 can be optimally supplied with power. The receiving circuit provided by the magnetic coil 709 provides an advantage in that it need not be physically connected to ground.

12A, 12B and 13, each of the receive circuits provided by the linear probe 703, the tuning resonator 706 and the magnetic coil 709 may be implemented in embodiments of the polyphase waveguide probes 200 described above Or the like. For this reason, the received energy can be used to power the electrical loads 716/726/736 via the matching network as can be seen. This is in contrast to signals that can be received at the receiver, transmitted in the form of a radiated electromagnetic field. Such signals have very low available power, and the receivers of such signals do not load the transmitters.

3) in which the receiving circuits provided by the linear probe 703, the tuning resonator 706 and the magnetic coil 709 are applied to the polyphase waveguide probe 200, Generating the surface acoustic waves to be applied to the circuits is also a feature of the present guided surface waves generated using the above-mentioned polyphase waveguide probes 200. [ This reflects the fact that the surface-acoustic wave generated by the given polyphase waveguide probe 200 described above includes a transmission line mode. Alternatively, the power source driving the radiated antenna producing the radiated electromagnetic waves is not loaded by the receivers irrespective of the number of receivers used.

Thus, a given polyphase waveguide probe 200 and receive circuits in the form of a linear probe 703, a tuned resonator 706 and / or a magnetic coil 709 may together constitute a wireless distribution system. When the distance of the transmission of the surface acoustic wave using the polyphase waveguide probe 200 as described above depends on the frequency, it is possible that the wireless power distribution can be achieved over a wide area, or even throughout the world.

Conventional wireless power transmission / distribution systems that are extensively studied today include "energy harvesting" from radiation fields and also sensor coupling to induction or response near fields. On the other hand, this wireless power system does not waste the power of radiated form that is lost forever unless it is intercepted. Presently disclosed wireless power systems are not limited to very short ranges, such as in conventional mutual reactance coupled near field systems. The wireless power system disclosed herein is probe coupled to a new surface induced transmission mode that is equivalent to delivering power to a load by a wave induction or to a load directly wired to a remote generator. Except for the power required to maintain the transmission field strength and the power dissipated in the surface waveguide at very low frequencies compared to the transmission losses at conventional high voltage power lines at 60 Hz, Only. When the electrical load demand is terminated, the source power generation is relatively low.

) 및 데드(dead) 네트워크 단자점 임피던스(

)에 의해 표현되는 테브난 등가로서 간주될 수 있다. 자기 코일(709)은 단락 회로 단자 전류 소스(

) 및 데드 네트워크 단자점 임피던스(

) 노턴 등가로서 간주될 수 있다. 각각의 전기 부하(716/726/736)(도 12a-b 및 도 13)는 부하 임피던스(

)에 의해 표현될 수 있다. 소스 임피던스(

)는 실수 및 허수 성분들 양자를 포함하며,

의 형태를 취한다.Referring now to FIG. 14A, a schematic diagram illustrating a linear probe 703 and a tuned resonator 706 is shown. Fig. 14B shows a schematic view showing the magnetic coil 709. Fig. Linear probe 703 and tunable resonator 706 each have an open circuit terminal voltage source (

) And dead network terminal point impedance (

). &Lt; / RTI &gt; The magnetic coil 709 is connected to a short circuit terminal current source (

) And dead network terminal point impedance (

) Norton equivalent. Each of the electrical loads 716/726/736 (Figs. 12A-B and 13) has a load impedance

). &Lt; / RTI &gt; Source Impedance (

) Includes both real and imaginary components,

.

According to one embodiment, each of electrical loads 716/726/736 is impedance matched to a respective receiving circuit. Specifically, each of the electric loads (716/726/736) is _{Z L '= Z s * =} R S - jX be equal to _{S, Z L' = R L} '+ j X L' Z L ' is expressed as And the load impedance Z _L 'provided is a complex conjugate of the actual source impedance Z _{S. The} load impedance Z _L ' is a complex conjugate of the actual source impedance Z _S. Furthermore, the principle of the pair match, which mentions that in a cascaded network when a match occurs at any terminal pair it will occur at every terminal pair, the real electrical load 716/726/736 also has its impedance Z _L '). Everitt, WL and GE Tanner, Communication Engineering , McGraw-Hill, 3 ^rd edition, 1956, p. 407 &lt; / RTI &gt; This ensures that each electrical load 716/726/736 is impedance matched to the respective receiving circuit and maximum power transfer is set for each electrical load 716/726/736.

In addition to the above, various embodiments of the present disclosure include, but are not limited to, the embodiments described in the following sections.

Clause 1. A method comprising transmitting energy carried in the form of an induced surface waveguide mode along the surface of a ground medium by exciting a polyphase waveguide probe.

2. The method of clause 1, wherein said step of transmitting energy carried in the form of said induced surface waveguide mode along said surface of said ground medium by excitation of said multiphase waveguide probe comprises the step of: &Lt; / RTI &gt; further comprising synthesizing a plurality of substantially matching chapters.

3. In Clause 1 or 2, the radial surface current density of the inductive surface waveguide mode is substantially

는

에 의해 주어지는 표면파 방사상 전파 상수이고,

는

에 의해 주어지는 수직 전파 상수이고,

이고,

는 상기 지상 매체의 전도율이고,

는 2πf와 동일하고, f는 상기 다상 도파관 프로브의 여기 주파수이고,

는 자유 공간의 유전율이고,

는 상기 지상 매체의 상대 유전율이고, 자유 공간 파수

은

와 동일하고,

는 상기 다상 도파관 프로브의 자유 공간 파장이고, j는

과 동일하고,

는 방사상 좌표이고, z는 상기 지상 매체에 수직인 수직 좌표이고,

는 방위 좌표이고,

는 순수 다상 프로브 전류이고,

는

시간 변화에 대한 복소 인수

를 갖는 제2 유형 및 1차의 핸켈 함수이고, t는 시간인 방법.Lt; RTI ID = 0.0 &gt;

The

Is a surface wave radiated wave constant given by?

The

Lt; / RTI &gt; is a vertical propagation constant given by &lt;

ego,

Is the conductivity of the ground medium,

Is the same as 2? F, f is the excitation frequency of the polyphase waveguide probe,

Is the permittivity of the free space,

Is the relative permittivity of the ground medium, and the free space wave number

silver

Lt; / RTI &gt;

Is the free space wavelength of the polyphase waveguide probe, j is

Lt; / RTI &gt;

Is a radial coordinate, z is a vertical coordinate perpendicular to the ground medium,

Is an azimuth coordinate,

Is the pure polyphase probe current,

The

Complex argument for time variation

A second type with a first order Hankel function and t is a time.

4. The method of any one of clauses 1 to 3, wherein said guide surface waveguide mode is substantially

및

And

는 방위 자기장 강도이고,

는 방사상 전기장 강도이고,

는 수직 전기장 강도이고,

는

에 의해 주어지는 표면파 방사상 전파 상수이고,

는

에 의해 주어지는 수직 전파 상수이고,

이고,

는 상기 지상 매체의 전도율이고,

는 2πf와 동일하고, f는 상기 다상 도파관 프로브의 여기 주파수이고,

는 자유 공간의 유전율이고,

는 상기 지상 매체의 상대 유전율이고, 자유 공간 파수

은

와 동일하고,

는 상기 다상 도파관 프로브의 자유 공간 파장이고, j는

과 동일하고,

는 방사상 좌표이고, z는 상기 지상 매체에 수직인 수직 좌표이고,

는 방위 좌표이고,

는 순수 다상 프로브 전류이고,

는

시간 변화에 대한 복소 인수

를 갖는 제2 유형 및 1차의 핸켈 함수이고, t는 시간인 방법.Lt; RTI ID = 0.0 &gt;

Is the azimuthal magnetic field strength,

Is the radial electric field strength,

Is the vertical electric field intensity,

The

Is a surface wave radiated wave constant given by?

The

Lt; / RTI &gt; is a vertical propagation constant given by &lt;

ego,

Is the conductivity of the ground medium,

Is the same as 2? F, f is the excitation frequency of the polyphase waveguide probe,

Is the permittivity of the free space,

Is the relative permittivity of the ground medium, and the free space wave number

silver

Lt; / RTI &gt;

Is the free space wavelength of the polyphase waveguide probe, j is

Lt; / RTI &gt;

Is a radial coordinate, z is a vertical coordinate perpendicular to the ground medium,

Is an azimuth coordinate,

Is the pure polyphase probe current,

The

Complex argument for time variation

A second type with a first order Hankel function and t is a time.

Item 5. The method of any of clauses 2-4, wherein the fields substantially combine the wavefront incident at the complex Brewster angle of the ground media to cause negligible reflection.

Item 6. The method of any one of clauses 1 to 5, wherein said polyphase waveguide probe comprises a plurality of charge terminals, said method comprising: adjusting the height of at least one charge terminal of said charge terminals, &Lt; / RTI &gt;

Clause 7. The method of any one of clauses 1-5, wherein the polyphase waveguide probe comprises a plurality of charge terminals, the method further comprising the step of tuning the polyphase waveguide probe by adjusting the distance between the charge terminals How to.

Clause 8. The method of any one of clauses 1 to 5, wherein said polyphase waveguide probe comprises a plurality of charge terminals, said method comprising the step of adjusting the size of at least one charge terminal of said charge terminals, &Lt; / RTI &gt;

9. The method of any one of clauses 1 to 5, wherein said polyphase waveguide probe comprises a plurality of charge terminals, said method comprising tuning said polyphase waveguide probe by adjusting a probe coupling circuit coupled to said charge terminals &Lt; / RTI &gt;

Item 10. An apparatus comprising a polyphase waveguide probe configured to generate a plurality of resultant chan- nels substantially mode-matched to a generic surface wave mode on a surface of a lossy conduction medium.

Item 11. The apparatus of Clause 10, wherein the loss conduction medium further comprises a ground medium.

Item 12. The apparatus of Clause 10 or 11, wherein the radiation resistance of the polyphase waveguide probe is substantially zero.

보다 작고, 여기서

는 상기 동작 주파수에서의 파장인 장치.13. The method of any one of clauses 10-12, wherein the height of the polyphase waveguide probe is greater than the height of the polyphase waveguide probe at an operating frequency

Smaller, where

Is a wavelength at the operating frequency.

14. The apparatus of any one of clauses 10-13, wherein the resulting chan- nels substantially combine the wavefront incident at a complex Brewster angle of the lossy conduction medium to cause a substantially zero reflection.

Item 15. The apparatus of any of clauses 10-14, wherein the excitation source is electrically coupled to the polyphase waveguide probe.

16. The method of any one of clauses 10-15, wherein the radial surface current density of the generator surface acoustic wave mode is substantially

는

에 의해 주어지는 표면파 방사상 전파 상수이고,

는

에 의해 주어지는 수직 전파 상수이고,

이고,

는 상기 손실 전도 매체의 전도율이고,

는 2πf와 동일하고, f는 상기 다상 도파관 프로브의 여기 주파수이고,

는 자유 공간의 유전율이고,

는 상기 손실 전도 매체의 상대 유전율이고, 자유 공간 파수

은

와 동일하고,

는 상기 다상 도파관 프로브의 자유 공간 파장이고, j는

과 동일하고,

는 방사상 좌표이고, z는 상기 손실 전도 매체에 수직인 수직 좌표이고,

는 방위 좌표이고,

는 순수 다상 프로브 전류이고,

는

시간 변화에 대한 복소 인수

를 갖는 제2 유형 및 1차의 핸켈 함수이고, t는 시간인 장치.Lt; RTI ID = 0.0 &gt;

The

Is a surface wave radiated wave constant given by?

The

Lt; / RTI &gt; is a vertical propagation constant given by &lt;

ego,

Is the conductivity of the lossy conduction medium,

Is the same as 2? F, f is the excitation frequency of the polyphase waveguide probe,

Is the permittivity of the free space,

Is the relative permittivity of the loss conductive medium, and the free space wave number

silver

Lt; / RTI &gt;

Is the free space wavelength of the polyphase waveguide probe, j is

Lt; / RTI &gt;

Is the radial coordinate, z is the vertical coordinate perpendicular to the loss conductive medium,

Is an azimuth coordinate,

Is the pure polyphase probe current,

The

Complex argument for time variation

And t is a time.

17. The method of any one of clauses 10-16, wherein said surface-

및

And

는 방위 자기장 강도이고,

는 방사상 전기장 강도이고,

는 수직 전기장 강도이고,

는

에 의해 주어지는 표면파 방사상 전파 상수이고,

는

에 의해 주어지는 수직 전파 상수이고,

이고,

는 상기 손실 전도 매체의 전도율이고,

는 2πf와 동일하고, f는 상기 다상 도파관 프로브의 여기 주파수이고,

는 자유 공간의 유전율이고,

는 상기 손실 전도 매체의 상대 유전율이고, 자유 공간 파수

은

와 동일하고,

는 상기 다상 도파관 프로브의 자유 공간 파장이고, j는

과 동일하고,

는 방사상 좌표이고, z는 상기 손실 전도 매체에 수직인 수직 좌표이고,

는 방위 좌표이고,

는 순수 다상 프로브 전류이고,

는

시간 변화에 대한 복소 인수

를 갖는 제2 유형 및 1차의 핸켈 함수이고, t는 시간인 장치.Lt; RTI ID = 0.0 &gt;

Is the azimuthal magnetic field strength,

Is the radial electric field strength,

Is the vertical electric field intensity,

The

Is a surface wave radiated wave constant given by?

The

Lt; / RTI &gt; is a vertical propagation constant given by &lt;

ego,

Is the conductivity of the lossy conduction medium,

Is the same as 2? F, f is the excitation frequency of the polyphase waveguide probe,

Is the permittivity of the free space,

Is the relative permittivity of the loss conductive medium, and the free space wave number

silver

Lt; / RTI &gt;

Is the free space wavelength of the polyphase waveguide probe, j is

Lt; / RTI &gt;

Is the radial coordinate, z is the vertical coordinate perpendicular to the loss conductive medium,

Is an azimuth coordinate,

Is the pure polyphase probe current,

The

Complex argument for time variation

And t is a time.

18. The method of any one of clauses 10-17, wherein the polyphase waveguide probe further comprises a plurality of charge terminals, wherein the polyphase waveguide probe is adapted to apply a plurality of voltage magnitudes and a plurality of phases on the charge terminals A further configured device.

19. The method of any one of clauses 10-18, wherein the polyphase waveguide probe further comprises a probe coupling circuit coupled to the charge terminals, the probe coupling circuit coupling the voltage magnitudes and the phases to the charge terminal Lt; / RTI &gt;

20. The apparatus of any one of clauses 10-19, wherein the phases as well as the voltage magnitudes vary as a function of the geographical position of the charge terminals relative to one another.

21. The apparatus of any of clauses 10-20, wherein said voltages as well as said phases vary as a function of the geographic location of each of said charge terminals relative to said lossy conduction medium.

22. The apparatus of any one of clauses 10-21, wherein the phases as well as the voltage magnitudes vary as a function of the physical size of the charge terminals.

23. The apparatus of any one of clauses 10-22, wherein the phases as well as the voltage magnitudes vary as a function of the electrical circuit.

24. The apparatus of any one of clauses 10-23, wherein the charge terminals are disposed along an axis.

25. An apparatus according to any one of clauses 10-24, wherein the excitation source is coupled in series with the polyphase waveguide probe.

Item 26. The apparatus of any of clauses 10-17, wherein the polyphase waveguide probe further comprises a coil coupled to both the first charge terminal and the second charge terminal.

Item 27. The method of any one of clauses 10-17, wherein the polyphase waveguide probe further comprises a first coil and a second coil, wherein the first coil is coupled to both the first charge terminal and the second charge terminal, And the second coil is coupled to the second charge terminal and the lossy conduction medium.

28. The multi-phase waveguide probe of any of claims 10-17, wherein the polyphase waveguide probe further comprises a coil having a first end coupled to the first charge terminal and a second end coupled to the loss conductive medium, And coupled to the second charge terminal and disposed along the coil.

29. The method of any one of clauses 10-17, wherein the polyphase waveguide probe further comprises a first coil and a second coil, wherein the first coil is coupled to both the first charge terminal and the lossy conduction medium, And the second coil is coupled to both the second charge terminal and the lossy conduction medium.

Item 30. A device according to any one of clauses 10-17, wherein the polyphase waveguide probe further comprises a coil coupled to the first charge terminal and the loss conductive medium, and a resistor coupled to the second charge terminal and the unit. .

Item 31. The apparatus of any of clauses 10-17, wherein the polyphase waveguide probe further comprises a coil coupled to both the first charge terminal and the ground screen.

Item 32. The method of any one of clauses 10-17, wherein the polyphase waveguide probe comprises: a first coil coupled to both the first charge terminal and the second charge terminal; And a second coil coupled to the loss conductive medium and the capacitor; Wherein the capacitance is further coupled to the second charge terminal.

Item 33. The apparatus of clause 32, wherein the capacity is variable capacity.

34. The method of any one of clauses 10-17, wherein the polyphase waveguide probe comprises: a first coil coupled to both the first charge terminal and the second charge terminal; And a second coil coupled to the terminal and the lossy conduction medium, the terminal being disposed relative to the second electronic terminal to cause a coupling capacitance between the terminal and the second charge terminal.

35. The method of any one of clauses 10-17, wherein the polyphase waveguide probe comprises: a first coil coupled to both the first charge terminal and the second charge terminal; And a second coil coupled to the terminal, the terminal being disposed with respect to the second charge terminal to cause a coupling capacitance between the terminal and the second charge terminal, the excitation source being coupled to the second coil and the second coil, And wherein the device is coupled to the lossy conduction medium.

Item 36. The apparatus of any one of clauses 10-17, wherein the polyphase waveguide probe further comprises a plurality of charge terminals, each of the terminals comprising a sphere or disc.

Item 37. The method of any one of clauses 10-17, wherein the polyphase waveguide probe has a coil having a first end coupled to a first charge terminal and a second end coupled to a second charge terminal; And a tab coupled to the loss conductive medium and disposed along the coil.

38. The apparatus of any of clauses 26-34, 36 and 37, further comprising an excitation source coupled to the primary coil, wherein the primary coil is mechanically coupled to the polyphase waveguide probe.

Item 39. A device comprising a polyphase waveguide probe configured to produce a plurality of resultant chanks, wherein the resulting chan- nels are substantially mode-matched to a gen- eral surface wave mode on the surface of the terrestrial medium.

40. The apparatus of clause 39, wherein the resulting chapters substantially combine waves incident at a complex Brewster angle of the ground media to cause a reflection of substantially zero.

41. The method of clause 39 or 40, wherein the radial surface current density of the generator surface acoustic wave mode is substantially

는

에 의해 주어지는 표면파 방사상 전파 상수이고,

는

에 의해 주어지는 수직 전파 상수이고,

이고,

는 상기 지상 매체의 전도율이고,

는 2πf와 동일하고, f는 상기 다상 도파관 프로브의 여기 주파수이고,

는 자유 공간의 유전율이고,

는 상기 지상 매체의 상대 유전율이고, 자유 공간 파수

은

와 동일하고,

는 상기 다상 도파관 프로브의 자유 공간 파장이고, j는

과 동일하고,

는 방사상 좌표이고, z는 상기 지상 매체에 수직인 수직 좌표이고,

는 방위 좌표이고,

는 순수 다상 프로브 전류이고,

는

시간 변화에 대한 복소 인수

를 갖는 제2 유형 및 1차의 핸켈 함수이고, t는 시간인 장치.Lt; RTI ID = 0.0 &gt;

The

Is a surface wave radiated wave constant given by?

The

Lt; / RTI &gt; is a vertical propagation constant given by &lt;

ego,

Is the conductivity of the ground medium,

Is the same as 2? F, f is the excitation frequency of the polyphase waveguide probe,

Is the permittivity of the free space,

Is the relative permittivity of the ground medium, and the free space wave number

silver

Lt; / RTI &gt;

Is the free space wavelength of the polyphase waveguide probe, j is

Lt; / RTI &gt;

Is a radial coordinate, z is a vertical coordinate perpendicular to the ground medium,

Is an azimuth coordinate,

Is the pure polyphase probe current,

The

Complex argument for time variation

And t is a time.

42. The method of any one of clauses 39 to 41, wherein said surface-

및

And

는 방위 자기장 강도이고,

는 방사상 전기장 강도이고,

는 수직 전기장 강도이고,

는

에 의해 주어지는 표면파 방사상 전파 상수이고,

는

에 의해 주어지는 수직 전파 상수이고,

이고,

는 상기 지상 매체의 전도율이고,

는 2πf와 동일하고, f는 상기 다상 도파관 프로브의 여기 주파수이고,

는 자유 공간의 유전율이고,

는 상기 지상 매체의 상대 유전율이고, 자유 공간 파수

은

와 동일하고,

는 상기 다상 도파관 프로브의 자유 공간 파장이고, j는

과 동일하고,

는 방사상 좌표이고, z는 상기 지상 매체에 수직인 수직 좌표이고,

는 방위 좌표이고,

는 순수 다상 프로브 전류이고,

는

시간 변화에 대한 복소 인수

를 갖는 제2 유형 및 1차의 핸켈 함수이고, t는 시간인 장치.Lt; RTI ID = 0.0 &gt;

Is the azimuthal magnetic field strength,

Is the radial electric field strength,

Is the vertical electric field intensity,

The

Is a surface wave radiated wave constant given by?

The

Lt; / RTI &gt; is a vertical propagation constant given by &lt;

ego,

Is the conductivity of the ground medium,

Is the same as 2? F, f is the excitation frequency of the polyphase waveguide probe,

Is the permittivity of the free space,

Is the relative permittivity of the ground medium, and the free space wave number

silver

Lt; / RTI &gt;

Is the free space wavelength of the polyphase waveguide probe, j is

Lt; / RTI &gt;

Is a radial coordinate, z is a vertical coordinate perpendicular to the ground medium,

Is an azimuth coordinate,

Is the pure polyphase probe current,

The

Complex argument for time variation

And t is a time.

43. The method of any one of clauses 39-42, wherein the polyphase waveguide probe further comprises a pair of charge terminals, wherein the polyphase waveguide probe is adapted to apply a plurality of voltage magnitudes and a plurality of phases on the charge terminals A further configured device.

44. The apparatus of claim 43, wherein the polyphase waveguide probe further comprises a distribution circuit coupled to the charge terminals.

Item 45. The apparatus of clause 44, wherein an electrical source is coupled to the decomposition circuit.

46. The apparatus of clause 44 or 45, wherein the distribution circuit further comprises a coil.

Item 47. The apparatus of clause 43, wherein the polyphase waveguide probe further comprises a coil coupled between the charge terminals.

48. The apparatus of clause 43, wherein the phases as well as the voltage magnitudes vary as a function of the geographical position of the charge terminals relative to one another.

49. The apparatus of clause 43, wherein the voltages as well as the phases vary as a function of the geographic location of each of the charge terminals relative to the ground media.

50. The apparatus of clause 43, wherein the voltages as well as the phases vary as a function of the physical size of the charge terminals.

51. The apparatus of clause 43, wherein the phases as well as the voltage magnitudes vary as a function of the electrical circuit.

Item 52. The apparatus of any one of clauses 39-44 and 46-51, wherein the excitation source is electrically coupled to the polyphase waveguide probe.

Clause 53. Placing the receiving circuit on the ground medium; And receiving, via the receiving circuit, energy carried in the form of a generic surface wave on the surface of the ground media.

54. The method of clause 53, wherein an electrical load coupled to the receiving circuit applies a load to an excitation source coupled to a polyphase waveguide probe producing the generator surface wave.

55. The method of claim 53 or 54, wherein the energy further comprises power, the method further comprising the step of applying the power to an electrical load coupled to the receiving circuit, / RTI &gt;

56. The method of any one of clauses 53-55, further comprising impedance matching the electrical load to the receiving circuit.

Clause 57. The method according to any one of clauses 53-56, further comprising setting a maximum power transfer from the receiving circuit to the electrical load.

Clause 58. The method according to any one of clauses 53-57, wherein the receiving circuit further comprises a magnetic coil.

Clause 59. The method according to any one of clauses 53-57, wherein the receiving circuit further comprises a linear probe.

60. The method of any one of clauses 53-57, wherein the receiving circuit further comprises a tuning resonator coupled to the ground media.

Item 61. An apparatus comprising receiving circuitry for receiving energy carried in the form of a generator surface wave along a surface of a lossy conduction medium.

62. The apparatus of claim 61, wherein the lossy conduction medium further comprises a ground medium.

Clause 63. The apparatus of clause 61 or 62, wherein an electrical load coupled to the receiving circuit applies a load to an excitation source coupled to a polyphase waveguide probe producing the generator surface wave.

Item 64. The method of clause 61 or 62, wherein the energy further includes power, the receiving circuit is coupled to an electrical load, the power is applied to the electrical load, and the power is a power source for the electrical load Devices used.

65. The device of clause 63 or 64, wherein the electrical load is impedance matched to the receiving circuit.

66. The apparatus of any one of clauses 61-65, wherein the receiving circuit further comprises a magnetic coil.

Item 67. The apparatus of any one of clauses 61-65, wherein the receiving circuit further comprises a linear probe.

Clause 68. The apparatus of any one of clauses 61-65, wherein the receiving circuit further comprises a tunable resonator.

69. The apparatus of clause 68, wherein the tunable resonator comprises a series tunable resonator.

70. The apparatus of clause 68, wherein the tunable resonator comprises a parallel tunable resonator.

71. The apparatus of clause 68, wherein the tunable resonator comprises a dispersion tunable resonator.

Clause 72. A polyphase waveguide probe for transmitting electrical energy in the form of a surface acoustic wave along a surface of a ground medium; And a receiving circuit for receiving the electrical energy.

73. The power transmission system of clause 72, wherein an electrical load coupled to the receiving circuit applies a load to the polyphase waveguide probe.

74. The power transmission system of clause 72, wherein an electrical load is coupled to the receiving circuit and the electrical energy is used as a power source for the electrical load.

Item 75. The electrical transmission system according to item 73 or 74, wherein the electrical load is impedance matched to the receiving circuit.

76. The power transmission system according to clause 73 or 74, wherein maximum power transfer from the receiving circuit to the electrical load is established.

77. The power transmission system according to any one of clauses 72-76, wherein the receiving circuit further comprises a magnetic coil.

78. The power transmission system of any one of clauses 72-76, wherein the receiving circuit further comprises a linear probe.

Clause 79. The power transmission system according to any one of clauses 72-76, wherein the receiving circuit further comprises a tuning resonator.

80. The power transmission system of any one of clauses 72-79, wherein the polyphase waveguide probe is configured to generate a plurality of resultant fields that are substantially impedance matched to the surface-acoustic-wave mode on the terrestrial medium.

Clause 81. The power transmission system of any one of clauses 72-80, wherein the radiation resistance of the polyphase waveguide probe is substantially zero.

보다 작고, 여기서

는 상기 동작 주파수에서의 파장인 전력 송신 시스템.Clause 82. In clauses 72-81, the height of the polyphase waveguide probe is greater than the height of the polyphase waveguide probe at an operating frequency

Smaller, where

Is a wavelength at the operating frequency.

83. The power transmission system of clause 80, wherein the resulting chan- nels substantially combine the wavefront incident at a complex Brewster angle of the lossy medium to cause a reflection of substantially zero.

84. The power transmission system of any one of clauses 72-83, wherein the excitation source is electrically coupled to the polyphase waveguide probe.

85. In Article 80, the radial surface current density of the surface acoustic wave mode is substantially

는

에 의해 주어지는 표면파 방사상 전파 상수이고,

는

에 의해 주어지는 수직 전파 상수이고,

이고,

는 상기 손실 매체의 전도율이고,

는 2πf와 동일하고, f는 상기 다상 도파관 프로브의 여기 주파수이고,

는 자유 공간의 유전율이고,

는 상기 손실 매체의 상대 유전율이고, 자유 공간 파수

은

와 동일하고,

는 상기 다상 도파관 프로브의 자유 공간 파장이고, j는

과 동일하고,

는 방사상 좌표이고, z는 상기 손실 매체에 수직인 수직 좌표이고,

는 방위 좌표이고,

는 순수 다상 프로브 전류이고,

는

시간 변화에 대한 복소 인수

를 갖는 제2 유형 및 1차의 핸켈 함수이고, t는 시간인 전력 송신 시스템.Lt; RTI ID = 0.0 &gt;

The

Is a surface wave radiated wave constant given by?

The

Lt; / RTI &gt; is a vertical propagation constant given by &lt;

ego,

Is the conductivity of the lossy medium,

Is the same as 2? F, f is the excitation frequency of the polyphase waveguide probe,

Is the permittivity of the free space,

Is the relative dielectric constant of the lossy medium,

silver

Lt; / RTI &gt;

Is the free space wavelength of the polyphase waveguide probe, j is

Lt; / RTI &gt;

Is the radial coordinate, z is the vertical coordinate perpendicular to the loss medium,

Is an azimuth coordinate,

Is the pure polyphase probe current,

The

Complex argument for time variation

And t is a time. &Lt; Desc / Clms Page number 21 &gt;

86. The method of clause 80, wherein said surface-acoustic-wave mode is substantially

및

And

는 방위 자기장 강도이고,

는 방사상 전기장 강도이고,

는 수직 전기장 강도이고,

는

에 의해 주어지는 표면파 방사상 전파 상수이고,

는

에 의해 주어지는 수직 전파 상수이고,

이고,

는 상기 손실 매체의 전도율이고,

는 2πf와 동일하고, f는 상기 다상 도파관 프로브의 여기 주파수이고,

는 자유 공간의 유전율이고,

는 상기 손실 매체의 상대 유전율이고, 자유 공간 파수

은

와 동일하고,

는 상기 다상 도파관 프로브의 자유 공간 파장이고, j는

과 동일하고,

는 방사상 좌표이고, z는 상기 손실 매체에 수직인 수직 좌표이고,

는 방위 좌표이고,

는 순수 다상 프로브 전류이고,

는

시간 변화에 대한 복소 인수

를 갖는 제2 유형 및 1차의 핸켈 함수이고, t는 시간인 전력 송신 시스템.Lt; RTI ID = 0.0 &gt;

Is the azimuthal magnetic field strength,

Is the radial electric field strength,

Is the vertical electric field intensity,

The

Is a surface wave radiated wave constant given by?

The

Lt; / RTI &gt; is a vertical propagation constant given by &lt;

ego,

Is the conductivity of the lossy medium,

Is the same as 2? F, f is the excitation frequency of the polyphase waveguide probe,

Is the permittivity of the free space,

Is the relative dielectric constant of the lossy medium,

silver

Lt; / RTI &gt;

Is the free space wavelength of the polyphase waveguide probe, j is

Lt; / RTI &gt;

Is the radial coordinate, z is the vertical coordinate perpendicular to the loss medium,

Is an azimuth coordinate,

Is the pure polyphase probe current,

The

Complex argument for time variation

And t is a time. &Lt; Desc / Clms Page number 21 &gt;

It should be emphasized that the above-described embodiments of the present disclosure are only possible examples of implementations described for a clear understanding of the principles of the present disclosure. Many changes and modifications may be made to the above-described embodiments without departing substantially from the teachings and principles of this disclosure. All such modifications and variations are intended to be included within the scope of this disclosure and protected by the following claims. In addition, all optional and preferred features and modifications of the described embodiments and dependent claims are useful in all aspects of the disclosure taught herein. Moreover, optional and preferred features and modifications of the described embodiments, as well as individual features of the dependent claims, may be combined and interchanged where applicable. For this reason, the various embodiments described above disclose elements that can be optionally combined in various ways depending on the desired implementation.

Claims (27)

Transmitting the energy carried in the form of an induced surface waveguide mode along the surface of the ground medium by exciting the polyphase waveguide probe
&Lt; / RTI &gt; The method according to claim 1,
Wherein said step of transferring energy carried in the form of said guided surface waveguide mode along said surface of said ground medium by exciting said multiphase waveguide probe comprises providing a plurality of sheets substantially matching said guiding surface waveguide mode of said ground media Wherein the fields substantially combine the wavefront incident at the complex Brewster angle of the terrestrial medium to cause negligible reflection. Phase waveguide probe configured to generate a plurality of resultant chan- nels that are substantially mode-matched to a generic surface-wave mode on the surface of the lossy conduction medium.
/ RTI &gt; The method of claim 3,
Wherein the lossy conduction medium further comprises a ground medium. The method according to claim 3 or 4,
Wherein the radiation resistance of said polyphase waveguide probe is substantially zero. 6. The method according to any one of claims 3 to 5,
Wherein the height of the polyphase waveguide probe is greater than the height of the polyphase waveguide probe at an operating frequency

Smaller, where

Is a wavelength at the operating frequency. 7. The method according to any one of claims 3 to 6,
Wherein the resulting fields substantially combine the wavefront incident at the complex Brewster angle of the lossy conduction medium to cause a reflection of substantially zero. 8. The method according to any one of claims 3 to 7,
Wherein the source is electrically coupled to the polyphase waveguide probe. 9. The method according to any one of claims 3 to 8,
Wherein the radial surface current density of the generator surface acoustic wave mode is substantially

Lt; RTI ID = 0.0 &gt;

The

Is a surface wave radiated wave constant given by?

The

Lt; / RTI &gt; is a vertical propagation constant given by &lt;

ego,

Is the conductivity of the lossy conduction medium,

Is the same as 2? F, f is the excitation frequency of the polyphase waveguide probe,

Is the permittivity of the free space,

Is the relative permittivity of the loss conductive medium, and the free space wave number

silver

Lt; / RTI &gt;

Is the free space wavelength of the polyphase waveguide probe, j is

Lt; / RTI &gt;

Is the radial coordinate, z is the vertical coordinate perpendicular to the loss conductive medium,

Is an azimuth coordinate,

Is the pure polyphase probe current,

The

Complex argument for time variation

And t is a time. 10. The method according to any one of claims 3 to 9,
The surface acoustic wave mode is substantially

And

Lt; RTI ID = 0.0 &gt;

Is the azimuthal magnetic field strength,

Is the radial electric field strength,

Is the vertical electric field intensity,

The

Is a surface wave radiated wave constant given by?

The

Lt; / RTI &gt; is a vertical propagation constant given by &lt;

ego,

Is the conductivity of the lossy conduction medium,

Is the same as 2? F, f is the excitation frequency of the polyphase waveguide probe,

Is the permittivity of the free space,

Is the relative permittivity of the loss conductive medium, and the free space wave number

silver

Lt; / RTI &gt;

Is the free space wavelength of the polyphase waveguide probe, j is

Lt; / RTI &gt;

Is the radial coordinate, z is the vertical coordinate perpendicular to the loss conductive medium,

Is an azimuth coordinate,

Is the pure polyphase probe current,

The

Complex argument for time variation

And t is a time. And a multi-phase waveguide probe configured to generate a plurality of resultant fields,
Wherein the resulting chapters are substantially mode-matched to a first surface-acoustic-wave mode on the surface of the ground media. 12. The method of claim 11,
Wherein the resulting fields substantially combine waves incident at a complex Brewster angle of the ground media to cause a reflection of substantially zero. 13. The method according to claim 11 or 12,
Wherein the polyphase waveguide probe further comprises a pair of charge terminals and wherein the polyphase waveguide probe is further adapted to impose a plurality of voltage magnitudes and a plurality of phases on the charge terminals. 14. The method of claim 13,
And wherein the polyphase waveguide probe further comprises a coil coupled between the charge terminals. Disposing a receiving circuit for the ground media; And
Receiving, via the receiving circuit, energy carried in the form of a generic surface wave on the surface of the ground medium
&Lt; / RTI &gt; 16. The method of claim 15,
Wherein an electrical load coupled to the receiving circuit applies a load to an excitation source coupled to a polyphase waveguide probe producing the generator surface wave. 16. The method of claim 15,
Wherein the energy further comprises power, the method further comprising applying the power to an electrical load coupled to the receiving circuit, wherein the power is used as a power source for the electrical load. 18. The method according to any one of claims 15 to 17,
And impedance matching the electrical load to the receiving circuit. 19. The method according to any one of claims 16 to 18,
And setting a maximum power transfer from the receiving circuit to the electrical load. A receiving circuit for receiving energy carried in the form of a surface acoustic wave along a surface of a lossy conduction medium.
/ RTI &gt; 21. The method of claim 20,
Wherein the lossy conduction medium further comprises a ground medium. 22. The method according to claim 20 or 21,
Wherein an electrical load coupled to the receiving circuit applies a load to an excitation source coupled to a polyphase waveguide probe producing the generator surface wave. 23. The method according to any one of claims 20 to 22,
Wherein the receiving circuit further comprises one of a magnetic coil, a linear probe, or a tunable resonator. A polyphase waveguide probe for transmitting electrical energy in the form of a surface acoustic wave along the surface of the ground medium; And
A receiving circuit for receiving the electrical energy
&Lt; / RTI &gt; 25. The method of claim 24,
And wherein an electrical load coupled to the receiving circuit applies a load to the polyphase waveguide probe. 25. The method of claim 24,
Wherein an electrical load is coupled to the receiving circuit and the electrical energy is used as a power source for the electrical load. 27. The method of claim 25 or 26,
Wherein maximum power transfer from the receiving circuit to the electrical load is established.

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