KR20130087982A - Compressed sensing based fast gnss and spread spectrum signal acquisition method and apparatus thereof - Google Patents

Abstract

PURPOSE: A compression sensing method and a device thereof are provided to reduce the hardware size of a receiver for obtaining a signal. CONSTITUTION: A global navigation satellite system (GNSS) and band spread signal acquisition device includes a signal detecting unit (300). A signal searching unit searches a GNSS positioning signal or a band spread signal. A signal searching unit comprises a parallel correlator (310), a signal generator (320), and a decoding module (330). A signal generator generates a compression sensing matrix including a matrix size (M×L) by using a measurement matrix including a matrix size (M<L) and a correlation matrix including a matrix size (L×L). A parallel correlator executes correlation between a received signal and a compression sensing matrix by constructing the M correlator in parallel. The decoding module obtains a Doppler frequency and a code phase of the received signal by using the compression correlation result of the correlator.

Description

Compression sensing method and device therefor for fast signal acquisition of NSS and spread signal TECHNICAL FIELD

Embodiments of the present invention relate to fast signal acquisition of Global Navigation Satellite System (GNSS) and Spread Spectrum signal receivers. In particular, fast signal search and search using compression sensing techniques. and acquisition methods and apparatus.

GNSS collectively refers to a positioning system using a satellite network such as GPS, and measures the time of arrival (TOA) of a radio wave from an antenna of the satellite and arrives at the GPS receiver. This is to calculate the relative position of the GPS receiver by locating the satellites. The signal from the GNSS satellite to the ground is the spread spectrum signal used in a typical wireless communication system. The first requirement for the receiver of all communication and satellite navigation systems using spread spectrum signals to start is the search for satellite signals and the acquisition of satellite information.

In acquiring the initial position of the GPS receiver, the GPS receiver remains powered off for a long period of time (usually more than 4 hours), and then powers on again after all the GPS satellite related data that has been received has been lost. The initial position measurement is called cold start and the time taken is called TTFF (time to first fix). In the case of GPS, the cold start of the GPS receiver to detect the L1 (= 1.575 GHz) frequency C / A code signal takes several steps, in particular the code phase of the satellite signal (corresponding to the temporal delay of the signal). Searching for hypothesis range and Doppler frequency The finding of the Doppler frequency and code phase of an actual received signal requires the highest hardware complexity. In other words, in the case of the C / A code sequence loaded on the L1 frequency of the GPS, 1023 chips are used, so the code phase to be searched by the receiver in the initial stage is 2046 units in 0.5 chip units. For receivers, the Doppler frequency hypothesis is set every 500Hz from -5KHz to + 5KHz, so a total of 21 Doppler hypotheses are generated, so all 442966 hypotheses (about 43000) have to be verified. One hypothesis verification is verified through a correlator with a correlation length of at least 1 msec, so a receiver with only one correlator requires about 43 seconds of signal search time, whereas a receiver using 43000 correlators is about 1 msec. You can search for a signal within. Here, the correlation length of 1 msec is a good case that the strength of the received signal is sufficiently high, and if the signal strength is not good, the correlation length should have a value larger than 1 msec. Therefore, the Doppler frequency hypothesis unit should be smaller than 500 Hz. The number of Doppler frequency hypotheses increases in inverse proportion to the correlator length. The reason why the GPS receiver must search and acquire the C / A signal is to track the GPS satellite signal received at the present moment in real time. The signal acquisition obtains two things; (1) precise synchronization between the received satellite signal and the signal generated internally in the receiver-ie, code synchronization and (2) frequency synchronization with the received GPS satellite signal. The receiver of a general spread spectrum communication system also performs a signal search for each code phase hypothesis in order to achieve code synchronization. (In a ground system, the Doppler frequency is not large, so most accurate frequency synchronization is not required).

Figure 1 shows all code phase hypotheses (0.5 chip unit hypothesis in 1023 chips-2046 total) and all Doppler frequency hypotheses (-5 Hz to +5 KHz in 500 Hz increments for a typical GPS receiver to search for one satellite signal. This is the result of searching all 43,000 hypotheses. The x-y plane in FIG. 1 is 42000 hypothetical planes, and the output value of the correlator is shown on the Z axis for each point (hypothesis) on the hypothetical plane. The output of the correlator is a value obtained by operating a correlator having a correlation length of 1 msec for each hypothesis (= code and Doppler frequency hypothesis), and using a correlator having a longer correlation length to detect a weak signal. If so, the Doppler frequency hypothesis should be searched in smaller units.

The process of performing signal search and acquiring the detected signal by verifying many hypotheses as described above is very inefficient because it uses a large amount of hardware resources at the beginning of the receiver's startup. Various techniques have been developed to reduce this inefficiency or to speed up signal detection. The simplest technique is to use multiple parallel correlators. For example, 42000 parallel correlators complete signal search quickly in less than 1 msec, but have very high correlators for only one use, resulting in high hardware inefficiency. Another method is to perform a fast correlation by multiplying the received signal with the PRN code generated in the receiver in the frequency domain using the Fast Fourier Transform (FFT). Since all code signals generated in s have to be Fourier-transformed into the frequency domain, and the result multiplied in the frequency domain must be inverse Fourier transformed into the time domain, the computational load is very high. Such a high computational FFT-based signal search method can be implemented using a high speed DSP chip, but there is a problem of high cost due to the DSP chip.

Another method is Qualcomm Inc.'s A-GPS (Assisted GPS, or collectively Assisted GNSS) technology. 2 shows a comparison of time to first fix (TTFF) in cold start in an A-GPS receiver and a conventional GPS receiver. As shown in FIG. 2, in the case of the existing GPS, the TTFF takes at least 30 seconds (up to 12.5 minutes), while the A-GPS receiver receives the TTFF by assisting the A-GPS server connected to the mobile communication base station. It can be completed in just a few seconds. A-GPS has a significant performance improvement over the existing GPS technology, but has some limitations and problems. First, since help information must be obtained from a server connected to a mobile communication network, it can be implemented only when mobile communication and wireless connection are possible, and the code phase range that the GPS receiver must search is based on the time synchronization between the base station and the mobile phone and the location of the mobile phone. It depends on the accuracy. Therefore, in the case of Asynchronous Cellular Networks such as 3rd Generation or 3.5th Generation and 4th Generation, the base station time is inaccurate and the size of the code phase search area becomes the entire 1023 chip which is the search area of the existing GPS receiver. GPS performs the same signal search as conventional GPS.

One of the breakthroughs in recent signal processing technologies is the Compressed Sensing technology, which is a very small amount of signal measurements that can be obtained from traditional methods of extracting information contained in signals from large amounts of signal measurements. FIG. 2 is a technology capable of extracting information contained in a signal. Compression detection techniques are currently used in many systems for processing images and processing large amounts of data.However, most of the techniques for obtaining signal measurements use a random measurement matrix, resulting from signal measurements. The amount of computation required to recover the original signal is so high that it cannot be used as a real-time system. In addition, no signal detection and acquisition techniques have been developed in spread spectrum systems using compression sensing techniques.

The present invention proposes a technique that enables the minimization of receiver hardware and the extremely high speed of signal acquisition in signal acquisition of a GNSS and spread spectrum communication receiver.

The present invention proposes to implement an optimal structured deterministic compressed sensing scheme for a GNSS and a spread spectrum receiver to minimize hardware, and to search for an initial signal of a communication terminal using a conventional GNSS receiver or spread spectrum signal. The goal is to reduce the number of correlators needed for the system by reducing the number of hardware resources required for signal detection.

The present invention provides an implementation method capable of rapid signal detection by dramatically reducing the amount of computation required for signal detection by presenting a structural measurement matrix without using a random measurement matrix used in a general compression sensing technique.

Compression detection technology implemented in the present invention is a technology that can quickly extract the necessary information from the minimum measurement value in real time, a technique that can be utilized in a system for detecting and acquiring a signal requiring very high speed and its implementation method to provide.

According to an aspect of the present invention, a global navigation satellite system (GNSS) and a spread spectrum signal acquisition apparatus includes a signal search unit for searching at least one received signal of a GNSS positioning signal or a spread spectrum signal, the signal The search unit uses a correlation matrix having a matrix size of [L × L] and a measurement matrix having a matrix size (M <L) of [M × L] whose number of rows is smaller than the number of columns. A signal generator for generating a compression sensing matrix having a magnitude; A parallel correlator having M correlators configured in parallel to perform correlation between the received signal and the compression sensing matrix; And a decoding module for obtaining a code phase and a Doppler frequency of the received signal by using the compression correlation result of the parallel correlator.

The signal generator may generate the compression sensing matrix by matrix multiplication of the correlation matrix and the measurement matrix.

Each row of the correlation matrix may be a spreading code having a code delay of an arbitrary time unit and the same as a spreading code signal of the received signal.

The parallel correlator may generate the compression correlation result having a matrix size of [M × 1] by performing correlation between the M rows constituting the compression sensing matrix and the received signal.

Each row of the compression correlation result is obtained by multiplying each of the L columns constituting the measurement matrix by the product of the correlation matrix and the received signal by a coefficient, and as a result, may be a linear sum of all columns of the measurement matrix. have.

The decoding module inputs M rows of the compression sensing matrix to the parallel correlator for all Doppler frequency hypotheses, obtains the M compression correlation results from the parallel correlator, and then obtains the M compression correlation results obtained from the parallel correlator. From the Doppler frequency and the code phase of the received signal can be obtained.

And the first signal generator to the signal generator by using the measurement matrix having a matrix size of [M ₁ × L] generating a first compressed sensing matrix having a matrix size of _{[M 1 × L], [} M 2 × L] (where M ₁ + M ₂ = M), using a measurement matrix having a matrix size of [M ₂ × L] to a second order signal generator that generates a second order compression sensing matrix having a matrix size of [M ₂ × L]. And a parallel correlator comprising a M ₁ correlator, a _first parallel correlator configured to receive M ₁ rows of the _first compression sensing matrix, and perform correlation with the received signal; and M ₂ correlators. And a second order parallel correlator configured to receive M ₂ rows of the second order compression sensing matrix and perform correlation with the received signal.

The decoding module estimates the Doppler frequency closest to the received signal from the M ₁ compression correlation result obtained for all Doppler frequency hypotheses in the first parallel correlator, and then estimates the estimated Doppler frequency in the second parallel correlator. The final code phase of the received signal can be obtained from the M ₂ compression correlations obtained for the received Doppler frequencies.

According to another aspect of the present invention, a method for obtaining GNSS and spread spectrum signals for finding at least one received signal among a GNSS positioning signal or a spread spectrum signal and obtaining a code phase and a Doppler frequency index of the received signal is described in [L × L]. A compression sensing matrix having a matrix size of [M × L] is generated by using a correlation matrix having a matrix size and a measurement matrix having a matrix size (M <L) of [M × L] where the number of rows is smaller than the number of columns. Doing; Performing correlation between the received signal and the compression sensing matrix through M correlators configured in parallel for each Doppler frequency hypothesis; And obtaining a Doppler frequency of the signal received from the correlation result and obtaining a code phase of a spreading code.

According to the embodiment of the present invention, it is possible to search and acquire signals several times faster than conventional GNSS and receiver technologies of the spread spectrum communication system, It is possible to realize an optimal receiver in terms of time and cost.

FIG. 1 shows a result of searching a two-dimensional GPS signal hypothesis search region consisting of all code phase hypotheses and all Doppler frequency hypotheses for a general GPS receiver to search for one satellite signal.
2 is a comparative analysis of the time required for initial position detection at cold start of a GPS receiver in Qualcomm's A-GPS and a conventional GPS method.
3 is a block diagram illustrating an internal configuration of a signal search unit based on a compression sensing technique according to an embodiment of the present invention.
4 is a block diagram illustrating an internal configuration of a signal search unit based on a compression sensing technique according to another embodiment of the present invention.
5 illustrates an auto-correlation function (ACF) of a general GPS and Galileo signal.

Hereinafter, two exemplary embodiments of the present invention will be described in detail with reference to the accompanying drawings. However, the embodiments of the present invention may be modified into various other similar forms, and the scope of the present invention is not limited to the embodiments described below. For example, Global Navigation Satellite Systems (GNSS) has developed a full range of satellite navigation systems including US Global Positioning System (US GPS, Glonass in Russia, Galileo in Europe, Beidou in China, QZSS in Japan, etc.) ).

Although GNSS is represented by GPS in the present invention, since most GNSS uses a spread spectrum signal in the same way as GPS, most GNSS receivers are similar in implementation to GPS receivers, Unless otherwise stated. In addition, since the receiver of a mobile communication system using a spread spectrum signal performs the same function, the field to which the present invention is applied is utilized in a mobile communication system using a spread spectrum signal, satellite communication, and a navigation system such as GPS (GNSS). Can be. One difference is that in spread spectrum mobile communication systems, only the code phase is searched without considering the Doppler frequency error of the received signal, while the GNSS searches not only the code phase of the received signal but also the Doppler frequency of the received signal. do. Therefore, in the present invention, the GNSS receiver is regarded as a more general receiver, and an implementation example thereof is described. The implementation of the GNSS receiver proposed in the present invention can be applied to a spread spectrum mobile communication terminal as it is.

Accordingly, the present invention will be described with reference to the embodiments shown in the drawings, which are merely exemplary and will be understood by those skilled in the art that various modifications and variations of the embodiments are possible therefrom. And, such modifications should be considered to be within the technical protection scope of the present invention. Accordingly, the true scope of the present invention should be determined by the technical idea of the appended claims.

FIG. 3 is a first embodiment of the present invention, based on a compressed sensing technique in which a general GNSS receiver searches for a received GNSS signal and acquires code phase information based on the present invention. An example of the implementation of a signal searcher is shown.

As shown in FIG. 3, the GNSS and the spread spectrum signal acquisition apparatus according to the first embodiment include a signal search unit 300. In this case, the signal search unit 300 includes parallel correlators ( 310, a signal generator 320, and a decoding (signal information search) module 330.

First, the signal received through the antenna 301 of the receiver provided in the GNSS and the spread spectrum signal acquisition device is a chip speed through the frequency down converter (302) and ADC (Analogue to Digital Converter) (303) A sampled signal obtained by sampling at a sampling frequency twice as high as a chip rate is input to the signal search unit 300. In general, since the code phase of the signal received at the receiver is searched in units of 0.5 chips, it is assumed in the present invention that the output of the ADC 303 is sampled at 2 chips per chip. The signal input to the signal search unit 300 is first passed through a parallel correlator 310 composed of M correlators and then output to a decoding module 330 which is a digital signal processor as a correlation output of each correlator. At this time, the parallel correlator 310 correlator 1 (311), correlator 2 (312),. Correlator M (313), the correlator input of the parallel correlator 310 according to the present invention is different from the correlation scheme for the search for a conventional conventional spread signal.

Hereinafter, a specific process of obtaining a measurement value of a GPS or spread spectrum signal by the compression detection technique according to the present invention will be described.

In general, a correlator for detecting a spread spectrum signal performs a correlation between an internally generated signal and a received signal, and the internally generated signal has the same spreading code as a spreading code of the received signal. A signal made up of a code. A receiver having a parallel correlator generates a plurality of spreading code signals having different time delays (code phases) from the internally generated signals and performs correlation between each signal and the received signal.

However, the correlation scheme according to the invention performs a correlation between the received signal and a special signal produced internally. The particular signal is each row of the A matrix and the A matrix has a magnitude of [M × L]. The internally generated special signal is a signal generated through the signal generator block 320 and is made through the following process. At this time, the signal generator block 320 may be composed of a PN generator 322, Ψ generator 323, Φ generator 324, A generator 325.

(F는 도플러 주파수 인덱스)를 갖는 확산코드 신호의 상관행렬(Correlation Matrix)로서 다음 수학식 1과 같은 압축 행렬로 나타낼 수 있다.First, the information in block 321 is transmitted to the signal generator 320 as including basic information (matrix size of A matrix and spreading code information) for making A matrix, and starts with a value of F = 1. Based on the information in block 321, the PN generator 322 generates a spreading code signal of the signal to be acquired. Generation of spreading codes is generally implemented using a linear feedback shift register (LFSR). The matrix Ψ produced by Ψgenerator 323 via PN generator 322 is the Doppler frequency.

F is a correlation matrix of a spreading code signal having a Doppler frequency index and may be represented by a compression matrix shown in Equation 1 below.

Where Ψ ₁ and Ψ ₂ are as follows.

라고 할 때(R_c는 chip rate), 상기 Ψ₁의 각 행은 0.5T_c 단위의 서로 다른 코드(위상)지연을 갖는 확산코드를 나타낸다. 상기 Ψ₁은 크기 [L×L]을 가지며, 이하 L=2N을 나타낸다. 또한,

는 현재 탐색하고자 하는 수신신호의 도플러 주파수를 나타내며, F는 탐색하고자 하는 도플러 주파수의 정수 인덱스로서,

를 가장 작은 도플러 주파수 값이라고 할 때, 미리 정해진 작은 주파수 간격

에 대하여

로 나타낸다. 이렇게 F를 증가시켜 다양한 도플러 주파수를 탐색하려는 이유는 지상 수신기에 수신되는 위성 신호의 도플러 주파수를 처음부터 정확히 예측할 수 없기 때문이다. 일반적으로 지상 GPS 수신기가 탐색하는 도플러 주파수 범위는 주로 -5KHz에서 +5KHz까지 10KHz 대역이고 1ms의 상관길이(correlation length)를 가정할 때

가 500Hz가 되므로,

를 -5KHz로 한다면 F는 0~21까지의 정수 값을 갖는다. 그러나,

의 값은 수신기의 상관길이(correlation length)에 따라서 달라지므로 (반비례) 이하에서는 F=1,2,3,…, F_max로 표현한다.In Equation 1, p [k] is the value of the kth chip (code) of the spreading code of the signal to be searched (k = 1, 2, 3, ... N), and a value of +1 or -1. Has Where N (>> 1) is the number of codes corresponding to one period of the spreading code. (For reference, the C / A code of the GPS L1 signal is known as N = 1023, N = 32768 in IS-95 (CDMA) mobile communication, and 38400 in 3G mobile communication WCDMA.) The reason why p [k] is repeated two times in order to synchronize the receiver to generate two samples per chip from the received signal. That is, the time length of one chip of the spreading code

(R _c is the chip rate), each row of Ψ ₁ is 0.5T _c Represents a spreading code with different code (phase) delays in units. Ψ ₁ has a size [L × L], hereinafter L = 2N. Also,

Is the Doppler frequency of the received signal to be searched, F is the integer index of the Doppler frequency to be searched,

Is the smallest Doppler frequency value, the predetermined small frequency interval

about

Respectively. The reason for increasing the F to search for various Doppler frequencies is because it is impossible to accurately predict the Doppler frequencies of satellite signals received from terrestrial receivers from the beginning. In general, the Doppler frequency range searched by terrestrial GPS receivers is mainly in the 10KHz band from -5KHz to + 5KHz and assumes a correlation length of 1ms.

Becomes 500 Hz,

If is set to -5KHz, F has an integer value from 0 to 21. But,

Since the value of depends on the correlation length of the receiver (inversely), F = 1,2,3,. Expressed as F _max .

Parallel correlators in conventional receivers receive received signal samples of the same length as each row of Ψ ₁ and perform correlation between the two inputs. In addition, if applied to the embodiment of the general GNSS receiver of the present invention, only Ψ = Ψ ₁ and Ψ ₂ can be utilized in the form of being directly multiplied by the signal received in the RF block of the GNSS receiver, so that only Ψ = Ψ ₁ is used. (I.e., the role of Ψ ₂ is to correct the Doppler frequency of the received GNSS signal so that the RF receiver can be implemented through preprocessing.) However, the compression according to an embodiment of the present invention (Fig. 3) is performed. In the implementation of the signal search unit 300 of the GNSS or spread spectrum signal using a detection technique, an internally generated signal input to each parallel correlator 310 is generated through the A generator 325, and the A generator 325 is The correlation matrix Ψ is made and the matrix product of the measurement matrix Φ generated in the Φ generator 324 of the following equation (2).

Here, matrix m ₁ th row and l-th column in the element value and the m-th row and l ₂ in the second column the value of the elements of Φ ₂ Φ ₁ is expressed as follows.

,

(여기서, M₀₁ 및 M₀₂는 양의 정수),

는 0보다 큰 실수(주로 3~7 사이의 값)이고, WH_M1는 [M₁×M₁]의 크기를 갖는 Walsh-Hadamard 행렬이며

는 천장값함수(ceil function)이다. 따라서 Φ₁은 [M₁×M₁] 크기의 월쉬(Walsh) 행렬의 각 열(column)이 β번 연속으로 반복된 형태의 행렬이다. 예를 들어 β=8이고 L=2046인 경우, M₁=256가 되므로 Φ₁은 다음과 같이 구해진다. (일반적으로

을 사용한다.)And,

,

Where M ₀₁ and M ₀₂ are positive integers,

Is a real number greater than 0 (usually between 3 and 7), and WH _M1 is a Walsh-Hadamard matrix with the size [M ₁ × M ₁ ]

Is the ceiling function. Therefore, Φ ₁ is a matrix in which each column of the Walsh matrix having the size [M ₁ × M ₁ ] is repeated β times in succession. For example, when β = 8 and L = 2046, since M ₁ = 256, Φ ₁ is obtained as follows. (Generally

Is used.)

은 Φ₁의 열 개수에 맞추기 위하여 β보다 작은 회수로 반복되었다. 그러나, Φ₁을 이루는 월쉬코드 열들은 대부분 β번 반복된다.Here, wi is an arbitrary Walsh code index ( ₁ ≦ Wi ≦ M ₁ ) and the Walsh code column W _wi having wi as the Walsh code index has a size of [M ₁ × 1]. Also, the Walsh code in the last column

Was repeated a number less than β to match the number of columns of Φ ₁ . However, most Walsh code strings forming Φ ₁ are repeated β times.

(여기서

는 transpose)라고 할 때, M(=M₁+M₂)개의 병렬상관기(310)를 구성하는 각 상관기(311~313)는 상기 압축감지행렬 A의 모든 행(row)과 수신되는 신호 간의 상관을 수행하여 측정치(Y)를 얻는다. 즉, 측정치 벡터는 Y=AX으로서, Y는 [(M₁+M₂)×1]의 크기를 갖는다.As a result, the measurement matrix Φ becomes a matrix of size [(M ₁ + M ₂ ) × L] (wherein (M ₁ + M ₂ ) << L). Therefore, the finally generated compression sensing matrix A (= ΦΨ) has a matrix size of [(M ₁ + M ₂ ) × L]. The received signal of [L × 1] size obtained by sampling (sampling) twice per chip of the received signal r (t)

(here

Is a transpose, each correlator 311 to 313 constituting M (= M ₁ + M ₂ ) parallel correlators 310 correlates between all rows of the compression sensing matrix A and the received signal. Is performed to obtain the measured value Y. That is, the measurement vector is Y = AX, and Y has a size of [(M ₁ + M ₂ ) × 1].

When the correlation output of Ψ with respect to the sampled received signal X is expressed as X ₁ , the output Y of the (M ₁ + M ₂ ) parallel correlators is represented by Equation 3 below.

에서의 Y₁측정치)을 얻는데 사용하고, 모든 도플러 주파수에 대하여 얻어진 F_max개의 Y₁으로부터 수신 신호와 주파수가 가장 가까운 도플러 주파수를 추정한 후, 추정된 도플러 주파수에 대해서만 Φ₂Ψ로부터 얻은 M₂개의 행을 이용하여 Y₂을 얻어 보다 정확한 코드위상을 추정해낸다. 이렇게 되면, 도 3의 방식 (모든 도플러 주파수에 대하여 (M₁+M₂)개의 측정치 Y을 얻는 방식)과는 달리 모든 도플러 주파수에 대하여 M₁개의 측정치 Y₁을 얻고 최종 추정된 도플러 주파수와 대략적 코드 위상 지연 값에 대해서만 1회에 한하여 M₂개의 측정치 Y₂을 얻는다. 따라서, 전자의 경우 총합 (M₁+M₂)F_max번의 상관함수를 구동해야 하지만, 후자의 경우에는 총합 M₁F_max+M₂번의 상관함수를 구동하는 것이 가능하다.Unlike FIG. 3, in the second embodiment of the present invention illustrated in FIG. 4, only M ₁ rows of A are used _first without using all (M ₁ + M ₂ ) rows of A according to an implementation algorithm. It is also possible to estimate the Doppler frequency and approximate code phase and perform signal search faster by using the remaining M ₂ rows of A later. That is, M ₁ rows made of Φ ₁ Ψ measured for different Doppler frequencies Y ₁ (F) (= Doppler frequency

Was used to obtain the Y ₁ measurement) in, and estimates a received signal and the frequency is closest to the Doppler frequency from the F _max of Y ₁ obtained for all Doppler frequencies, only the estimated Doppler frequency obtained from Φ ₂ Ψ M ₂ Use Y rows to get Y ₂ to estimate the more accurate code phase. This results in M ₁ measurements Y ₁ for all Doppler frequencies and approximate the final estimated Doppler frequency, unlike the scheme of FIG. 3 (where (M ₁ + M ₂ ) measurements Y are obtained for all Doppler frequencies). M ₂ measurements Y ₂ are obtained only once for the code phase delay value. Therefore, in the former case, the correlation function of the total (M ₁ + M ₂ ) F _max times must be driven, while in the latter case, the correlation function of the total M ₁ F _max + M ₂ times can be driven.

Conventional receivers using conventional parallel correlators should use L (= 2N) parallel correlators for all Doppler frequencies (F _max total) to obtain a total of LF _max correlation measurements. in the signal search section 300 using a compression technique in accordance with detected that it can also perform the same function as a measure for obtaining a minimum M ₁ + M ₂ F _max parallel compression Any deception (wherein, M _1, M ₂ << L). In the present invention, since M ₁ and M ₂ is a small value of 10-25% of L, the receiver according to the present invention receives GNSS (GPS) received using about 4-10 times fewer parallel correlators than conventional receivers. Alternatively, a spread spectrum communication signal may be detected and acquired.

과 같은 형식이 된다. 여기서, α는 수신 신호의 강도에 관계되는 실수이며 θ는 신호의 주파수 위상으로

이다. (만약, 칩 당 1회의 샘플링 주파수를 갖는 수신기라면 상기 L의 크기는 2N이 아니라 N이 되며 X₁의 원소들 중에서는 3개가 아닌 오직 1개의 원소만이 특별히 큰 절대 값을 갖는다.) 상기 절대 값이 작은 값들을 v로 표현하고 절대 값이 큰 연이은 값들을

로 표현하면 X₁은 v와 V가 섞여있는 벡터가 된다. 이때, 수신기는 X₁벡터에서 원소 V2가 위치한 인덱스(index)

를 찾아냄으로써 코드위상 탐색을 완료하게 된다. (이하의 설명에서 V2의 인덱스 값을

라고 표시한다. 또한 수신기가 예측한 도플러 주파수가 수신되는 신호의 도플러 주파수와 맞지 않는 경우에는 신호 상관도가 낮아지므로 X₁의 모든 요소의 값이 v로 표현될 수 있다.) 압축감지 기술을 사용하지 않고 병렬상관기를 구비한 일반적인 수신기에서는 상관행렬 Ψ만을 사용하므로 L개의 상관기 출력 X₁에서 단순히 가장 절대 값이 큰 원소를 찾고 그 원소의 인덱스(index)

를 찾는다.For example, suppose there is no frequency error due to the Doppler frequency in the received GPS signal (or if the receiver selects a frequency that matches the Doppler frequency of the received signal), then X ₁ (= ΨX) is the L parallels used by the existing receiver. As seen from the correlator output, most elements of the vector X ₁ (size [L × 1]) have a small absolute value, with only three consecutive elements of L having a particularly large absolute value (among others). The middle value has the largest triangle value). For example,

Is of the form Where α is a real number related to the strength of the received signal and θ is the frequency phase of the signal

to be. (If a receiver has one sampling frequency per chip, the size of L is N, not 2N, and only _one element of X ₁ has a particularly large absolute value, not three.) Represent successive values with small values as v and absolute values

X ₁ is a mixture of v and V. In this case, the receiver is an index (index) where the element V2 is located in the X ₁ vector

The code phase search is completed by finding. (In the description below, the index value of V2

Is displayed. In addition, if the Doppler frequency predicted by the receiver does not match the Doppler frequency of the received signal, the signal correlation becomes low, so the values of all elements of X ₁ can be expressed as v.) Parallel correlation without using a compression detection technique. In general, a receiver with a group uses only the correlation matrix Ψ and simply finds the element with the largest absolute value in the L correlator outputs X ₁ and indexes the element.

Find it.

Hereinafter, information necessary for signal acquisition from a measurement value of a GNSS or a spread spectrum signal obtained by a compression sensing technique according to the present invention (for GPS, a received signal in case of a mobile communication system having a code phase value, a Doppler frequency value of a received signal, and a spread spectrum method) The algorithm of the decoding module 330 that obtains the code phase value of &quot;

으로부터 도플러 주파수와 대략적인 코드위상 값을 얻고

로부터 정확한 코드위상을 얻는 방식이다. 제일 먼저, FWHT(Fast Walsh Hadamard Transform)함수(331)는 (탐색하는 도플러 주파수 값을 사용한 Ψ를 이용하여 얻어진 A로 측정한) M개의 병렬상관기(310) 상관출력 Y(크기 [M×1])를 입력 받고 그 중 Y₁에 대한 M₁개 FWHT (M₁ point-FWHT)를 수행하여 M₁개 결과 값을 얻고 그 결과 값에 절대값을 취하여 M₁개의 Z₁(F,wi)을 얻는다. (여기서 Z₁(F,wi)은 탐색하는 도플러 주파수를 나타내는 인덱스 F와 M₁개의 FWHT 출력 결과에 대한 인덱스 1≤wi≤M₁를 나타낸다.) 상기 M₁개의 출력 Z₁(F,wi) (1≤wi≤M₁)은 제1차판정함수(332)에서 미리 설정된 제1임계치(TH₁)과 비교되어 제1임계치(TH₁) 이상이 되는 출력 Z₁(F,wi)이 있는지 판단한다. 만일 제1차판정함수(332)의 판정식(Z₁(F,wi)>TH₁)을 만족하는 Z₁(F,wi)이 없는 경우 도플러주파수조절함수(335)로 진행하여 도플러 주파수 인덱스를 1만큼 증가시켜 다른 도플러 주파수 신호에 대한 병렬상관기(310) 출력Y(1~M|F)을 얻는다. 만일 제1차판정함수(332)의 제1차판정식(Z₁(F,wi)>TH₁)을 만족하는 Z₁(F,wi)이 존재하는 경우, 그 FWHT 출력 인덱스 wi를 Wi로 지정한다. 상세탐지함수(333)은 다음의 세 가지 입력: (1) FWHT함수(331)으로부터 Y₂를 (2) 제1차판정함수(332)로부터 Wi를 (3) 상기 신호생성기(320)으로부터

를 입력 받는다. 상세탐지함수(333)은 제일 먼저 입력 받은 Wi를 월쉬코드 인덱스로 갖는 Φ₁의 모든 (최대 β개의) 열 인덱스들(column indices)을 원소로하는 집합 Iw를 계산한다.First, in the first embodiment of FIG. 3, each row of the matrix A (total M rows) with respect to the Doppler frequency (expressed by the index F) to be searched is assigned to each correlator 311, 312, and 313 of the parallel correlator 310. After M correlation outputs Y (1 ~ M | F) are obtained by performing correlation with the sampled signal input and received

From the Doppler frequency and the approximate code phase

This is how to get the correct code phase from. Firstly, the Fast Walsh Hadamard Transform (FWHT) function 331 is the correlation output Y of M parallel correlators 310 (measured with A obtained using Ψ using Doppler frequency values to search). ) And M ₁ FWHT (M ₁ for Y ₁ ) point-FWHT) to obtain the M ₁ result and take the absolute value of the M ₁ to obtain Z ₁ (F, wi). (Wherein Z ₁ (F, wi) represents the index to the index 1≤wi≤M ₁ F and M ₁ of FWHT output indicating a Doppler frequency of searching.) The outputs M ₁ Z ₁ (F, wi) (1≤wi≤M ₁₎ is that the first decision function (332) a first threshold (TH ₁₎ is compared with the first threshold value becomes equal to or greater than the output (TH ₁₎ Z ₁ (F, wi) preset in To judge. If the first difference is determined to go to board formal (Z ₁ (F, wi)> TH ₁₎ Z ₁ (F, wi), the Doppler frequency control function 335. If not satisfying the function 332, the Doppler frequency index Is increased by 1 to obtain parallel correlator 310 output Y (1 ~ M | F) for another Doppler frequency signal. If there exists Z ₁ (F, wi) that satisfies the _first decision formula (Z ₁ (F, wi)> TH ₁ ) of the first decision function 332, the FWHT output index wi is designated as Wi. do. The detailed detection function 333 has three inputs: (1) Y ₂ from the FWHT function 331, (2) Wi from the first decision function 332, and (3) from the signal generator 320.

Get input. The detailed detection function 333 first calculates a set Iw having all (maximum β) column indices of Φ ₁ having Wi as the Walsh code index as an element.

Here, the set Iw can be expressed as Equation 4.

At this time, the set Iw has a maximum maximum β values between [1, L].

간의 상관(correlation)을 수행하고 그 상관 결과들의 절대 값을 취하여 Z₂(F,k)를 얻는다. (경우에 따라서는 잡음(Noise)의 영향으로 상기 Wi가 Nw(>1)개 발견될 수 있는데 이때, Iw는 모든 Wi를 월쉬코드 인덱스로 갖는 Φ₁의 모든 (최대 Nw×β개의) 열 인덱스들(column indices)을 원소로 하는 집합이 된다.) 상기 상세탐지함수(333)의 출력은 제2차판정함수(334)로 전달되어 미리 설정된 제2임계치(TH₂)에 대하여 제2차판정식 (Z₂(F,k)>TH₂)을 만족하는 k를 찾는다. 이때, 제2차판정식을 만족하는 Z₂(F,k)가 발견되면 제2차판정함수(334)는 그 인덱스 k를 현재 수신되는 신호의 코드위상(code phase) 추정 값

으로 출력하고 현재 탐색한 도플러 주파수를 실제 수신신호의 도플러 주파수 추정 값

으로 출력한다. 상기 제2차판정함수(334)에서 제2차판정식을 만족하는 Z₂(F,k)가 발견되지 않는 경우에, 도플러주파수조절함수(335)로 진행하여 탐색하려는 도플러 주파수를 바꿔 새로운 도플러 주파수에서 위의 신호탐색 과정을 반복 수행한다.And, the detailed detection function 333 is Y ₂ and

Correlation is performed and Z ₂ (F, k) is obtained by taking absolute values of the correlation results. (In some cases, due to the influence of noise, Nw (> 1) Wis may be found, where Iw is an index of all (up to Nw × β) columns of Φ ₁ having all Wis as Walsh code indices.) And the output of the detailed detection function 333 is transferred to the second determination function 334 so as to generate a second threshold value TH ₂ for a preset second threshold value TH ₂ . Find k that satisfies (Z ₂ (F, k)> TH ₂ ). At this time, if Z ₂ (F, k) is found that satisfies the second decision equation, the second decision function 334 uses the index k to estimate the code phase of the currently received signal.

Doppler frequency estimate of the actual received signal

Will output If Z ₂ (F, k) that satisfies the second decision equation is not found in the second decision function 334, the Doppler frequency control function 335 is performed to change the Doppler frequency to be searched to change the new Doppler frequency. Repeat the above signal search process.

가 실제 수신되는 도플러 주파수와 거의 같은 경우에만 사용된다.The first embodiment of the present invention with respect to FIG. 3 obtains a correlation output Y of magnitude [M × 1] every time for all Doppler frequencies, but the determination is performed using only Y _{1 in} the first determination function 332. Y ₂ is input to the detailed detection function 333 only when the first determination is satisfied. Therefore, in most cases Y ₂ is not used and the Doppler frequency

Is used only if is nearly equal to the actual Doppler frequency received.

In order to reduce such unnecessary measurement, Figure 4 shows a signal search unit 400 according to a second embodiment of the present invention. Functions 401, 402, 403, 422 and 423 of FIG. 4 perform the same functions as the functions 301, 302, 303, 322 and 323 of FIG.

First, generation of the first compression detection matrix A ₁ starts by receiving the following information. Set Stage to 1, and set Ψ and Φ ₁ to [L × L] and [M ₁ × L], respectively, and specify the Doppler frequency index value to determine the Doppler frequency to be searched. Start with 1). In addition, in order to specify the PN code information to be generated, the PN code information (PRN number) of the GNSS is received in the case of CDMA or WCDMA mobile communication, and receives the PN code information of the base station to be found. The Φ ₁ generator 424 generates a measurement matrix Φ _{1 in} which each column of the Walsh matrix WH _{M1 having} the size of [M ₁ × M ₁ ] is repeated β times in succession. Make. Accordingly, the output of the A ₁ generator 425 may be expressed as Equation 5 as the product of the correlation matrix Ψ and Φ ₁ output from the Ψ generator 423 as the _first compression detection matrix A ₁ (where Ψ). Is a correlation matrix, which is made according to the expression in Equation 1).

과 Wi를 블록 (451)로 출력한다. 블록 (451)에서는 상세한 코드위상(code phase) 값을 추정하기 위한 정보와 제2차압축감지행렬 A₃를 생성하기 위한 몇 가지 기본 정보를 블록 (450)으로 전달한다. (경우에 따라서는 잡음(Noise)의 영향으로 상기 Wi가 Nw(>1)개 발견될 수 있는데 이때, Iw는 모든 Wi를 월쉬코드 인덱스로 갖는 Φ₁의 모든 (최대 Nw×β개의) 열 인덱스들(column indices)을 원소로 하는 집합이 된다.) 상기 블록 (451)에서 블록 (450)으로 전달되는 정보는 상기 제1차판정함수(432)의 결과인 현재 탐색된 도플러 주파수 추정값

과 상기 측정행렬 Φ₁에서 Wi를 월쉬코드 인덱스로 갖는 모든 (최대 β개) 열들의 열 인덱스(column indices)들의 집합(Iw), A₃의 크기에 대한 정보 - M₃ 및 L, 그리고 PN코드생성을 위한 GNSS의 PRN정보 (CDMA 및 WCDMA이동통신의 경우 단말기가 탐색하고자 하는 PN코드 생성정보) 등으로 이루어진다. 상기 Iw는 수학식 4와 같이 만들어진다. 또한, 블록 (451)에서 변수 'Stage'는 2로 그 값이 지정된다(Stage=2). 상기 블록 (451)로부터 입력된 정보를 기반으로 A₃생성블록(450)에서는 먼저 PN생성기(452)로부터 PN코드를 생성한다 (함수 422는 함수 322와 동일). 상기 생성된 PN코드와 현재 탐색한 도플러 주파수 추정값

은 Ψ생성기(453)로 입력되어 도플러 주파수

에서의 상관행렬 Ψ를 만든다. 상기 블록 (451)로부터 입력된 Iw의 총 원소 개수를 N_I라고 할 때, Iw는 Φ₃생성기(454)로 입력되고 측정행렬 Φ₃는 [M₃×L]의 크기를 가지며 (여기서 M₃=2^x로서 x는

가 되는 최소 양의 정수(smallest positive integer)) 다음과 같이 만들어진다. 측정행렬 Φ₃는 Iw의 원소 값을 인덱스로 갖는 열(column)을 제외한 모든 열(column)이 제로벡터(0)로 구성되어 있으며, 나머지 열(column)들은 길이 M₃의 서로 다른 월쉬코드를 열로 갖는다. 따라서, Φ₃의 m₃번째 행과 l번째 열

은 다음과 같이 표현된다.Each row of the first compression detection matrix has a size of [1 × L] as shown in FIG. 3 and is input to each correlator 411, 412, 413 of the first parallel correlator 410, respectively. It is correlated with the sampled received signal (sample signal magnitude = [1 × L]). The output Y ₁ of each correlator 411, 412, 413 is the output of the first parallel correlator 410 for any Doppler frequency (index F) to be searched, and is expressed as Y ₁ (1 to M ₁ | F). Has a size of [M ₁ × 1]. FWHT function 431 is equal to _{Y 1 (1 ~ M 1 |} F) in the first embodiment of Figure 3 receives the M ₁ perform gae -FWHT (M ₁ -point Fast Walsh Hadamard Transform), and the M Z ₁ (F, wi) is obtained by taking the absolute value of M ₁ output of ₁ -FWHT. (Wherein Z ₁ (F, wi) represents an index wi (1≤wi≤M ₁₎ for index F and M ₁ of FWHT output indicating a Doppler frequency of searching.) The results M ₁ Z ₁ (F, wi) (1≤wi≤M ₁₎ is the first decision function (432) is input to a preset first threshold (TH ₁₎ and the first threshold (TH ₁₎ or more Z ₁ (F compared, wi Is determined. If the first proceeds to the first chapan formal (Z ₁ (F, wi)> TH ₁₎ Z ₁ (F, wi), the Doppler frequency control function 435. If not satisfying the difference determination function 432 doppler The frequency index is increased by 1 to repeat the signal measurement and determination (perform functions 420, 410, 431, 432) for the other Doppler frequency signals. If there exists Z ₁ (F, wi) that satisfies the first decision formula (Z ₁ (F, wi)> TH ₁ ) of the first decision function 432, the index wi is designated as Wi and the current Searched Doppler Frequency Estimates

And Wi to block 451. In block 451, information for estimating a detailed code phase value and some basic information for generating a second compression detection matrix A ₃ are passed to block 450. (In some cases, due to the influence of noise, Nw (> 1) Wis may be found, where Iw is an index of all (up to Nw × β) columns of Φ ₁ having all Wis as Walsh code indices.) Information transmitted from the block 451 to the block 450 is a current searched Doppler frequency estimate that is a result of the first decision function 432.

And information on the size of the column indices (Iw), A ₃ of all (maximum β) columns having Wi as the Walsh code index in the measurement matrix Φ ₁ , M ₃ and L, and the PN code. PRN information of the GNSS for generation (in the case of CDMA and WCDMA mobile communication, PN code generation information to be searched by the terminal). The Iw is made as shown in Equation 4. Also, in block 451 the variable 'Stage' is assigned a value of 2 (Stage = 2). Based on the information input from the block 451, the A ₃ generation block 450 first generates a PN code from the PN generator 452 (function 422 is the same as function 322). The generated PN code and the estimated Doppler frequency estimate

Is input to generator 453 so that the Doppler frequency

Create a correlation matrix Ψ at. When the total number of elements of Iw input from the block 451 is N _I , Iw is input to the Φ ₃ generator 454 and the measurement matrix Φ ₃ has a size of [M ₃ × L] (where M ₃ = 2 ^x where x is

Where the smallest positive integer is The measurement matrix Φ ₃ consists of zero vectors ( 0 ) of all columns except the column whose element value of Iw is an index, and the rest of the columns are different Walsh codes of length M ₃ . Have with heat. Therefore, the Φ _{3 3} m-th row and l-th column

Is expressed as:

이고, Wi가 2인 경우,

이 되며, Φ₃는 다음과 같다.Here, WH _M3 is a Walsh-Hadamard matrix having a size of [M ₃ × M ₃ ]. E.g,

, If Wi is 2,

Φ ₃ is as follows.

은 WH_M3의 n번째 열을 나타낸다.here,

Represents the nth column of WH _M3 .

. 상기 제2차압축감지행렬 A₃의 모든 행(row)들은 제2차병렬상관기(440)으로 입력되어 A₃의 각 행(row)과 수신된 샘플 신호간의 상관이 수행되고 그 병렬 상관의 결과로 (현재 탐색하는 도플러 주파수

와 수신 신호의 코드위상

의 대략적 추정을 Iw 구간 내로 가정했을 때의) 압축감지 측정치 Y₃를 얻는다. (여기서 Y₃는 크기 [M₃×1]를 가지며 Y₃(1~M₃|F,Iw)과 같이 표현할 수 있다. 또한, 상기 제2차병렬상관기(440)은 앞의 제1차병렬상관기(410)으로 대체될 수 있다.) 상기 출력 Y₃는 다시 FWHT함수(431)로 입력되고 FWHT함수(431)은 M₃개-FWHT (M₂-point Fast Walsh Hadamard Transform)를 수행하고 상기 FWHT함수(431)의 M₃개 출력 결과에 절대값을 취하여 M₃개의 Z₃(F,k)를 얻는다. (여기서 Z₃(F,k)은 탐색하는 도플러 주파수를 나타내는 인덱스 F와 M₃개의 FWHT 출력의 인덱스 1≤k≤M₃를 나타낸다.) 상기 M₃개의 결과 Z₃(F,k)는 제2차판정함수(434)에서 미리 설정된 제2임계치(TH₃)과 비교되어 제2임계치(TH₃) 이상이 되는 결과 Z₃(F,k)가 있는지 판단한다. 만일 제2차판정함수(434)의 제2차판정식(Z₃(F,k)>TH₃)을 만족하는 Z₃(F,k)가 없는 경우 도플러주파수조절함수(435)로 진행하여 도플러 주파수 인덱스를 1만큼 증가시켜 다른 도플러 주파수 신호에 대한 신호측정과 판정을 다시 반복한다(함수 420, 410, 431, 432를 수행). 만일 제2차판정함수(434)의 제2차판정식(Z₃(F,k)>TH₃)을 만족하는 Z₃(F,k)가 존재하는 경우, 그 FWHT 출력 인덱스 k를 현재 수신되는 신호의 코드위상(code phase) 추정치

로 지정하고 현재 탐색한 도플러 주파수 추정값

과

를 최종 신호 획득 결과로 출력한다.Finally, the A ₃ generator 455 makes the second compression detection matrix A ₃ the product of the measurement matrix Φ ₃ and the correlation matrix Ψ.

. All rows of the second compression detection matrix A ₃ are input to the second parallel correlator 440 to perform correlation between each row of A ₃ and the received sample signal and as a result of the parallel correlation. Doppler frequency currently searching

And phase of received signal

The compression detection measurement Y ₃ , assuming an approximate estimate of in the Iw interval, is obtained. (Wherein Y ₃ has a size [M ₃ × 1] and may be expressed as Y ₃ (1 to M ₃ | F, Iw). Also, the second parallel correlator 440 may be the first parallel correlator 440). 410). The output Y ₃ is input again to the FWHT function 431, and the FWHT function 431 performs M ₃ -FWHT (M ₂ -point Fast Walsh Hadamard Transform) and the FWHT function. M ₃ Z ₃ (F, k) is obtained by taking the absolute value of the M ₃ output results of (431). (Wherein Z ₃ (F, k) represents the index of the index F 1≤k≤M ₃ and M ₃ of FWHT output indicating a Doppler frequency of searching.) The results M ₃ Z ₃ (F, k) is the a second threshold previously set in the second function determination (434) (TH ₃₎ that is the second threshold value (TH ₃₎ above are compared to the results to determine if the Z ₃ (F, k). Ten thousand and one second control proceeds to the second chapan formal _{(Z 3 (F, k)} > TH 3) Z 3 when there is no (F, k), the Doppler frequency adjustment function (435) that satisfies the difference determination function 434 doppler The frequency index is increased by 1 to repeat the signal measurement and determination on another Doppler frequency signal (perform functions 420, 410, 431, 432). If there exists Z ₃ (F, k) that satisfies the second determination formula (Z ₃ (F, k)> TH ₃ ) of the second determination function 434, the FWHT output index k is currently received. Code phase estimate of the signal

Doppler frequency estimate currently specified and

and

Is output as a final signal acquisition result.

In the case of signals such as Galileo in Europe, in GPS, X ₁ shows a triangular shape where the code phase of the signal fits, but since the BOC (Binary Offset Carrier) method is used, the correlation result is not triangular ( It takes the form of a sinc function where the +) and (-) values intersect. 5 shows a correlation result between a general GPS and a Galileo signal. In FIG. 5, the horizontal axis is a relative code phase difference between a spreading code of a received signal and a spreading code generated internally by a receiver, and a value thereof is expressed in units of chips. Although the signal acquisition technique using the compression detection technique of GPS and spread spectrum signal proposed in the present invention can be used as it is in the general GNSS signal, but if you want to acquire the signal of the Galileo navigation satellite using the most common BOC (1,1) All odd rows of all measurement matrices Φ (ie, Φ ₁ , Φ ₂ and Φ _3, etc. as shown in FIGS. 3 and 4) with 2N (= L) rows made in 0.5 chip (T _c ) units multiply -1 by columns or all even columns. For example, when β = 8 and L = 2046 as an implementation example of FIG. 3, M ₁ = 256, and Φ ₁ for Galileo signal acquisition is obtained as follows.

또는

가 되며, 상기 도 4의 설명에서 예시한 Φ₃의 구현 예에서는 다음과 같이 변형하여 사용한다.

or

In the embodiment of Φ ₃ illustrated in the description of FIG. 4, the present invention is modified as follows.

또는

가 된다. 즉, 0.5칩(T_c) 단위로 L(=2N)개의 열을 갖는 측정행렬의 짝수 또는 홀수 열에 -1을 곱하여 생성한다. 따라서, BOC(1,1)를 이용하는 Galileo 신호의 경우 도 3 및 도 4의 구현에서 모든 측정행렬 Φ(Φ₁, Φ₂ 및 Φ₃ 등)의 모든 홀수열(odd columns) 또는 모든 짝수열(even columns)에 -1을 곱하면 된다.

or

. That is, it is generated by multiplying -1 by an even or odd column of a measurement matrix having L (= 2N) columns in units of 0.5 chips (T _c ). Thus, for Galileo signals using BOC (1,1), all odd columns or all even columns of all measurement matrices Φ (Φ ₁ , Φ ₂ and Φ _3, etc.) in the implementations of FIGS. even columns).

As such, according to two embodiments of the present invention, a separate communication device is not required as in the conventional A-GPS, and signals can be searched and acquired several times faster than the receiver technology of the conventional GNSS and spread spectrum communication systems. Since the hardware of the receiver for searching and signal acquisition is greatly consumed, the optimal receiver can be implemented with minimum time and hardware cost.

Methods according to an embodiment of the present invention may be implemented in the form of program instructions that can be executed by various computer means and recorded in a computer readable medium. The computer readable medium may include program instructions, data files, data structures, etc. alone or in combination. The program instructions recorded on the medium may be those specially designed and constructed for the present invention or may be available to those skilled in the art of computer software. In addition, the above-described file system can be recorded in a computer-readable recording medium.

As described above, the present invention has been described by way of limited embodiments and drawings, but the present invention is not limited to the above embodiments, and those skilled in the art to which the present invention pertains various modifications and variations from such descriptions. This is possible.

Therefore, the scope of the present invention should not be limited to the described embodiments, but should be determined by the equivalents of the claims, as well as the claims.

300, 400: signal search unit
310: parallel correlator
320: signal generator
330, 430: decoding module
410: first parallel correlator
440: second parallel correlator

Claims (16)

A signal search unit for searching for at least one received signal of a global navigation satellite system (GNSS) positioning signal or a spread spectrum signal,
The signal search unit,
A matrix size of [M × L] is determined by using a correlation matrix having a matrix size of [L × L] and a measurement matrix having a matrix size (M <L) of [M × L] where the number of rows is smaller than the number of columns. A signal generator for generating a compression sensing matrix having;
A parallel correlator having M correlators configured in parallel to perform correlation between the received signal and the compression sensing matrix; And
A decoding module for acquiring a code phase and a Doppler frequency of the received signal using the compression correlation result of the parallel correlator
GNSS and spread spectrum signal acquisition device comprising a. The method of claim 1,
The signal generator,
Generating the compression sensing matrix by matrix multiplication of the correlation matrix and the measurement matrix
And the GNSS and spread-spectrum signal acquisition apparatus. The method of claim 1,
Wherein each row of the correlation matrix is made up of a spreading code having a code delay of an arbitrary unit of time equal to a spreading code signal of the received signal
And the GNSS and spread-spectrum signal acquisition apparatus. The method of claim 1,
The parallel correlator,
Performing the correlation between the M rows constituting the compression sensing matrix and the received signal to generate the compression correlation result having a matrix size of [M × 1]
And the GNSS and spread-spectrum signal acquisition apparatus. 5. The method of claim 4,
Each row of the compression correlation result is
Wherein the L columns of the measurement matrix are obtained by multiplying the result of multiplying the correlation matrix and the received signal by a coefficient, resulting in a linear sum of all columns of the measurement matrix.
And the GNSS and spread-spectrum signal acquisition apparatus. The method of claim 1,
The decoding module includes:
M rows of the compression sensing matrix are input to the parallel correlator for all Doppler frequency hypotheses to obtain the M compression correlation results in the parallel correlator, and then the received signals from the M compression correlation results obtained in the parallel correlator. Obtaining the Doppler frequency and code phase of
And the GNSS and spread-spectrum signal acquisition apparatus. The method of claim 1,
The signal generator,
Using a measuring matrix having a matrix size of [M ₁ × L] and the first signal generator for generating a first compressed sensing matrix having a matrix size of [M ₁ × L],
Second order to generate a second order compression sensing matrix having a matrix size of [M ₂ × L] using a measurement matrix having a matrix size of [M ₂ × L], where M ₁ + M ₂ = M Signal generator
&Lt; / RTI &gt;
The parallel correlator,
The first parallel correlator that M is configured as _a single correlation receiving the first M ₁ rows in the compression sensing matrix performing a correlation with the received signal,
M consists of a group of _two correlation receiving the second row of the M ₂ matrix compression sensing the second parallel correlator for performing a correlation with the received signal
GNSS and spread spectrum signal acquisition device consisting of. The method of claim 7, wherein
The decoding module includes:
After estimating the Doppler frequency closest to the received signal and the frequency from the M ₁ compression correlation results obtained for all Doppler frequency hypotheses in the first parallel correlator, the second parallel correlator estimates the Doppler frequency. Obtaining the final code phase of the received signal from the M ₂ compression correlation results obtained for
And the GNSS and spread-spectrum signal acquisition apparatus. A method of obtaining a code phase and a Doppler frequency index of a received signal by finding at least one received signal of a GNSS positioning signal or a spread spectrum signal,
A matrix size of [M × L] is determined by using a correlation matrix having a matrix size of [L × L] and a measurement matrix having a matrix size (M <L) of [M × L] where the number of rows is smaller than the number of columns. Generating a compressed sense matrix having;
Performing correlation between the received signal and the compression sensing matrix through M correlators configured in parallel for each Doppler frequency hypothesis; And
Obtaining a Doppler frequency of the received signal from the correlation result and obtaining a code phase of a spreading code
GNSS and spread spectrum signal acquisition method comprising a. 10. The method of claim 9,
Generating the compression sensing matrix,
Generating the compression sensing matrix by matrix multiplication of the correlation matrix and the measurement matrix
GNSS and spread spectrum signal acquisition method characterized in that. 10. The method of claim 9,
Each row of the correlation matrix is composed of a spreading code equal to a code signal of the received signal and having a code delay of an arbitrary time unit
GNSS and spread spectrum signal acquisition method characterized in that. 10. The method of claim 9,
Performing correlation between the received signal and the compression detection matrix,
Generating the correlation result having a matrix size of [M × 1] by performing correlation between the M rows constituting the compression sensing matrix and the received signal
GNSS and spread spectrum signal acquisition method characterized in that. The method of claim 12,
Each row of the compression correlation result is
Wherein the L columns of the measurement matrix are obtained by multiplying the result of multiplying the correlation matrix and the received signal by a coefficient, resulting in a linear sum of all columns of the measurement matrix.
GNSS and spread spectrum signal acquisition method characterized in that. 10. The method of claim 9,
Acquiring a code phase and a Doppler frequency index of the received signal,
For every Doppler frequency hypothesis, M rows of the compression sensing matrix are input to the M correlators to obtain the M compressed correlation results in the M correlators, and then from the M compressed correlation results obtained in the M correlators. Acquiring Doppler frequency and code phase of the received signal
GNSS and spread spectrum signal acquisition method characterized in that. 10. The method of claim 9,
Generating the compression sensing matrix,
Using a measuring matrix having a matrix size of [M ₁ × L] comprising: generating a first compressed sensing matrix having a matrix size of [M ₁ × L]; And
Generating a second order compression sensing matrix having a matrix size of [M ₂ × L] using a measurement matrix having a matrix size of [M ₂ × L], where M ₁ + M ₂ = M
GNSS and spread spectrum signal acquisition method comprising a. 16. The method of claim 15,
Acquiring a code phase and a Doppler frequency index of the received signal,
All Doppler frequency hypotheses the first compression detection matrix of M M ₁ compression correlation frequency as the received signal from the result obtained for _one line from the M _one correlator to the input to the M correlators of the M _single correlator with respect to Estimating the nearest Doppler frequency; And
The M ₂ rows of the second order compression sensing matrix are input to M ₂ correlators of the M correlators with respect to the estimated Doppler frequency, and the received signals are obtained from the M ₂ compression correlation results obtained from the M ₂ correlators. Obtaining the final code phase
GNSS and spread spectrum signal acquisition method comprising a.

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