KR101440692B1 - 2-dimensional compressed correlator for fast gnss and spread spectrum signal acquisition and robust tracking and apparatus thereof - Google Patents

Abstract

GNSS and spread spectrum signal acquisition apparatus and GNSS and spread spectrum signal tracking apparatus are disclosed. The GNSS and the spread spectrum signal acquisition apparatus may include a signal search unit for searching at least one received signal of a Global Navigation Satellite System (GNSS) positioning signal or a spread spectrum signal, As the number of stages of the search unit increases, the degree of compression of a hypothesis range for searching for the received signal can be reduced to search for the received signal.

Description

{2-DIMENSIONAL COMPRESSED CORRELATOR FOR FAST GNSS AND SPRAY SPECTRUM SIGNAL ACQUISITION AND ROBUST TRACKING AND APPARATUS THEREOF}

Embodiments of the present invention relate to fast signal acquisition and robust signal tracking of a Global Navigation Satellite System (GNSS) receiver and, in particular, to rapid signal acquisition using a 2D Compressed Correlator search and acquisition, and robust tracking.

GNSS collectively refers to a positioning system using a satellite network such as GPS. It measures the time (TOA: Time of Arrival) that a radio wave is transmitted from a satellite antenna to arrive at a GPS receiver, And the relative position of the GPS receiver is calculated. The signal from the GNSS satellite to the ground is the spread spectrum signal used in a typical wireless communication system. The first requirement for the receiver of all communication and satellite navigation systems using spread spectrum signals to start is the search for satellite signals and the acquisition of satellite information. After acquisition of the signal, the receiver performs continuous signal tracking. A correlator is used for searching, acquiring, and tracking the spread spectrum signal. The correlator takes a correlation between a received signal and a signal generated inside the receiver, accumulates the correlation result, And a matched filter technique for determining whether or not to receive the signal.

In the case of the L1 (= 1.575 GHz) frequency C / A signal of the GPS, for example, the GPS cold start is performed in various stages, in particular, the code phase of the satellite signal, ) Hypothesis range and Doppler frequency The step of finding the Doppler frequency and the code phase of the actual received signal by searching the hypothesis area requires the highest hardware complexity. Further, in the case of a C / A code sequence carried on the L1 frequency of the GPS, the code phase is composed of 1023 code phases. Therefore, the code phase to be searched by the receiver in the initialization step is verified to a total of 2046 hypotheses hypothesis testing. In addition, assuming that a general pedestrian GPS receiver performs correlation of 1023 chip lengths (1 msec in time), it is necessary to set Doppler frequency hypothesis every 500 Hz from -5 KHz to +5 KHz, resulting in a total of 21 Doppler hypotheses And therefore 42966 (about 43000) hypotheses should be verified. Therefore, since a receiver having only one correlator takes a time of 1 msec per hypothesis, a signal seek time of 43 seconds is required for the entire hypothesis verification. On the contrary, the receiver using 43,000 correlators requires signal search within about 1 msec can do. In this case, the correlation length of 1 msec corresponds to a case where the intensity of the received signal is sufficiently high, and when the signal strength is not good, the correlation length should be larger than 1 msec and the Doppler frequency hypothesis unit should be set to be smaller than 500 Hz The number of Doppler frequency hypotheses increases in inverse proportion to the length of the correlator.

Figure 1 shows a GPS receiver having a correlation length of 1 msec correlates all code phase hypotheses (0.5 chip unit hypothesis in 1023 chips: total 2046 samples) and all Doppler frequency hypotheses (In total, 21 from -5 KHz to +5 KHz in 500 Hz steps). The x-y plane in FIG. 1 is 42000 hypothetical planes, and the output value of the correlator is shown on the Z axis for each point (hypothesis) on the hypothetical plane. The output of the correlator represents a degree of correlation obtained by operating a correlator having a correlation length of 1 msec for each hypothesis (= code and Doppler frequency hypothesis). From the hypothesis that the degree of correlation is higher than a predetermined threshold value, The Doppler frequency hypothesis should be searched in smaller increments if a correlator with a longer correlation length is used to detect a weak signal.

The process of verifying such a large number of hypotheses and performing a signal search and acquiring a detected signal is a very inefficient process in terms of hardware resource utilization since a lot of hardware resources are used intensively at the initial stage of the receiver . Various techniques have been developed to reduce this inefficiency or to speed up signal detection. The simplest technique is to use multiple parallel correlators. For example, 43000 parallel correlators complete the signal search within 1msec quickly, but there is a high inefficiency of the hardware because there are many correlators for only one use.

Another recent related technology is the rapid detection and acquisition of GNSS signals based on compression sensing (CS-GNSS technique). The CS-GNSS technique is similar to the recently introduced folding technique for verifying a plurality of code phases at once, but the folding technique and the CS-GNSS technique are both one-dimensional Technology.

The present invention utilizes a two-dimensional compression correlator in signal acquisition and signal tracking of GNSS and spread-spectrum communication receivers to minimize receiver hardware, speed up signal acquisition, and robustness of signal tracking for noise To improve the quality of the image.

The first object of the present invention is to enable simultaneous saving of hardware resources and rapid signal acquisition in a GNSS and a spread spectrum receiver which consume the most inefficient hardware resources.

A second object of the present invention is to maintain or increase the sensitivity in GNSS signal acquisition and signal tracking.

A third object of the present invention is to provide a broader linear tracking region in the signal trackers (DLL and PLL) of the GNSS and the spread spectrum receiver, thereby being robust against noise and reducing signal loss due to multipath signals .

The two-dimensional compression correlation technique implemented in the present invention is a technique that can be utilized for acquiring a signal requiring a very high speed and for robust signal tracking, and provides a technique and an implementation method in the present invention.

According to an embodiment of the present invention, the GNSS and the spread spectrum signal acquisition apparatus may include a signal search unit for searching at least one received signal of a Global Navigation Satellite System (GNSS) positioning signal or a spread spectrum signal, At this time, the signal search unit is composed of a multi-stage search unit, and as the stage of the search unit increases, the degree of compression for a hypothesis range for searching for the received signal is reduced to search for the received signal.

According to an embodiment of the present invention, the GNSS and the spread spectrum signal acquisition apparatus may include a signal search unit for searching for a received signal of at least one of a GNSS positioning signal and a spread spectrum signal, The received signal can be searched using hypotheses and a hypothesized region in which a plurality of individual code phase delay hypotheses are compressed.

According to an embodiment of the present invention, the GNSS and the spread spectrum signal tracking apparatus may include a signal tracking unit for tracking at least one received signal of a GNSS positioning signal or a spread spectrum signal, A line-time compression code correlator for performing correlation with a line-time code signal preceding a signal by a specific phase; And a post-temporal compression code correlator that performs a correlation with the time code signal after a specific phase is delayed from the currently received code signal.

According to the embodiment of the present invention, it is possible to search and acquire signals several times faster than conventional GNSS and receiver technologies of the spread spectrum communication system, It is possible to realize an optimal receiver in terms of time and cost.

According to the embodiment of the present invention, a two-dimensional compression correlator is applied to a code phase tracker and a phase tracker for signal tracking to have a wider linear characteristic, thereby realizing a system robust against instantaneous signal blur due to multipath and noise .

FIG. 1 shows a result of searching a two-dimensional GPS signal hypothesis search region consisting of all code phase hypotheses and all Doppler frequency hypotheses for a general GPS receiver to search for one satellite signal.
2 is a block diagram illustrating an internal configuration of a receiver signal search unit using a conventional GNSS correlator.
3 is a block diagram illustrating an internal configuration of a primary signal search unit in a receiver based on a two-dimensional compression correlation technique, according to an embodiment of the present invention.
4 is a block diagram illustrating an internal configuration of a secondary signal search unit in a receiver based on a two-dimensional compression correlation technique, according to an embodiment of the present invention.
5 is a flowchart showing a simplified operation of the primary signal search unit and the secondary signal search unit in an embodiment of the present invention.
6 is a block diagram illustrating an internal configuration of a code signal tracking apparatus using a compression correlator in an embodiment of the present invention.
7 is a graph illustrating a result of comparing a discriminator value of a code signal tracking apparatus using a compression correlator with a discriminator value of an existing EL correlator in an embodiment of the present invention.

DESCRIPTION OF THE PREFERRED EMBODIMENTS One exemplary embodiment of the present invention will now be described in detail with reference to the accompanying drawings. However, the embodiments of the present invention may be modified into various other similar forms, and the scope of the present invention is not limited to the embodiments described below. For example, Global Navigation Satellite Systems (GNSS) has developed a full range of satellite navigation systems including US Global Positioning System (US GPS, Glonass in Russia, Galileo in Europe, Beidou in China, QZSS in Japan, etc.) ).

Although GNSS is represented by GPS in the present invention, since most GNSS uses a spread spectrum signal in the same way as GPS, most GNSS receivers are similar in implementation to GPS receivers, Unless otherwise stated. In addition, since the receiver of the satellite communication system using the spread spectrum signal performs the same function, the field to which the present invention is applied can be utilized in a navigation system such as a GPS (GNSS) as well as a satellite communication using a spread signal . In the spread spectrum mobile communication system, only the code phase is searched without considering the Doppler frequency error of the received signal, but the compression correlation technique according to the present invention can be utilized. Therefore, in the present invention, the GNSS receiver is regarded as a more general receiver, and an implementation example thereof will be described. The GNSS receiver implementation of the present invention can be applied to spread spectrum satellite communication and a mobile communication terminal as it is.

Therefore, the present invention is described with reference to the embodiments shown in the drawings, but it is to be understood that various changes and modifications may be made without departing from the scope of the present invention. It should be understood that such modifications are within the technical scope of the present invention. Accordingly, the true scope of the present invention should be determined by the technical idea of the appended claims.

Figure 2 shows a signal search of a conventional GNSS receiver using a correlator. In addition, FIG. 2 shows a serial search method as a basic example of a method using a correlator. The implementation of the two-dimensional compression correlation technique proposed in the present invention is similar to that of the general signal search unit shown in FIG. 2, so that the description of FIG. 2 is preceded.

, 중심주파수

, 코드위상지연

을 갖는 위성 s의 코드신호 c(t)이다. 도시하는 바와 같이, I채널 및 Q 채널로 입력되는 수신신호

,

에는 각각

,

로 표현되는 AWGN (Additive White Guassian Noise, 백색잡음)가 포함되어 있다. 상기 수신신호

,

는 1차 곱셈기(201, 202)에서 수신기가 블록 204에서 생성한 특정 주파수

를 갖는 Carrier신호 및 90도 위상 변환된 Carrier 신호와 각각 곱해진다 (여기서

는 수신기가 알고 있는 값). 상기 1차곱셈기의 출력은 각각 대역통과필터(BPF: 211 및 212)와 샘플러(ADC: 221 및 222)를 거쳐

와

를 발생시킨다. (여기서 Ts는 샘플의 간격이고 n은 각 샘플링 시점의 시간 인덱스). 상기

와

는 다시 2차곱셈기 (231, 232)에서 수신기가 생성한 특정 코드위상지연(또는, 코드위상 code phase)

를 갖는 특정 위성 s의 코드신호 c(t)와 곱해진다. 상기 2차곱셈기(231, 232)의 출력은 각각

와

로 표시되며 누적기(241, 242)에서 N개의 시간적으로 연속된 출력이 누적되어 각각

와

를 발생한다. 상기

와

는 각각 제곱기(251, 252)에서 제곱되고 합산기(261)에서 그 값이 합해져서 판단변수(Decision Variable)

를 만든다. 상기 판단변수 Z는 1차판별기(262)에서 탐지임계치(Detection Threshold)

와 비교되어 Z값이

보다 큰 경우에는 GNSS 신호를 찾은 것으로 판단하여 이때의 도플러 주파수 값을 나타내는 인덱스

와 코드위상지연 값

를 출력한다. 상기 탐지임계치는 상관기의 동위상상관길이(Coherent Correlation Length) 및 비동위상누적수(Non-Coherent Accumulation Length), 탐지확률(Probability of Detection), 오보확률(Probability of False Alarm) 및 도플러 주파수 가설 간격 (Doppler frequency search step)등에 따라 결정된다. 만일, Z값이

보다 작은 경우에는 현재의 가설 (

로 표현되는 도플러 주파수 가설과

로 표현되는 코드위상지연 가설)이 틀린 것으로 판단하고

또는

를 하나 증가하여 새로운 가설에서의 신호 탐색을 수행한다. 이러한 신호 탐색은 일반적으로

의 도플러 주파수 가설 구간과

의 코드 위상 지연 가설 구간에 대하여 이루어진다. (GPS L1 C/A 신호를 이용하는 GPS 수신기라면 일반적으로 1msec의 상관길이를 가지는데, 칩당 2회 샘플링을 하고 C/A 신호의 주기가 1023이므로,

,

,

이 된다).First, the GNSS signal received from the antenna is converted to an IF (intermediate frequency) signal through a radio frequency (RF) processing block. In Fig. 2, a received signal already converted to an IF is depicted as an input of a signal search unit. The received signals are amplitude A, phase

, Center frequency

, Code phase delay

Is the code signal c (t) of the satellite < RTI ID = 0.0 > As shown in the figure, the reception signal inputted through the I channel and the Q channel

,

Respectively

,

(Additive White Guassian Noise, white noise). The received signal

,

0.0 > 201 < / RTI > and 202,

And a carrier signal having a phase shifted by 90 degrees, respectively

Is the value known to the receiver). The outputs of the first-order multipliers are respectively passed through band pass filters (BPFs) 211 and 212 and samplers (ADCs) 221 and 222

Wow

. (Where Ts is the interval of samples and n is the time index of each sampling point). remind

Wow

(Or code phase code phase) generated by the receiver in the second-order multipliers 231 and 232,

Is multiplied by a code signal c (t) of a particular satellite s having < RTI ID = 0.0 > The outputs of the second-order multipliers 231 and 232 are

Wow

And the N temporally consecutive outputs from the accumulators 241 and 242 are accumulated,

Wow

. remind

Wow

Are squared by the squarers 251 and 252, respectively, and their values are summed in a summer 261 to obtain a decision variable,

. The judgment variable Z is detected by the primary discriminator 262,

And the Z value is

It is determined that the GNSS signal is found, and the index indicating the Doppler frequency value at this time

And the code phase delay value

. The detection threshold may include a Coherent Correlation Length and a Non-Coherent Accumulation Length, a Probability of Detection, a Probability of False Alarm, and a Doppler frequency hypothesis spacing Doppler frequency search step). If the Z value is

The current hypothesis (

Doppler frequency hypothesis and

The code phase delay hypothesis expressed as < RTI ID = 0.0 >

or

And the signal search in the new hypothesis is performed. This signal search is generally

The Doppler frequency hypothesis interval

Of the code phase delay. (If the GPS receiver using the GPS L1 C / A signal has a correlation length of 1 msec in general, sampling is performed twice per chip and the cycle of the C / A signal is 1023,

,

,

.

보다 작은 것으로 판정되면, 2차판별기(271)에서 현재 시험된 코드위상지연 가설이 마지막 가설인지 판별한다. 즉,

인지 그 여부를 판별하고 그렇지 않은 경우(

)에는 다음 코드위상(지연) 가설을 시험해보도록 블록234에서

값을 하나 더 증가시키고 증가된

값을 현재의 코드위상지연 값으로 갖는 GNSS 위성의 PRN 코드신호를 블록233에서 발생시킨다. (참고로, 초기 도플러 주파수 및 코드위상지연 가설은 블록205에서 지정하는 바와 같이

및

이다.) 따라서, 현재 시각 t=nTs에서 발생되는 신호는 도시하는 바와 같이,

로 표현될 수 있다. 만일 2차판별기에서

로 판별된 경우에는 블록 272에서

으로 초기화 하고 블록234로 진행한다. 동시에 2차판별기에서

로 판별된 경우에는, 3차판별기(281)로 진행하여

인지 그 여부를 판별하는데, 만일

인 경우에는 블록235로 진행하여

값을 1만큼 더 증가시켜 새로운 도플러 주파수 가설에서 신호 탐색을 시작하도록 만든다. 만일

인 경우에는 블록282에서

으로 초기화 시키고 블록235로 진행한다. 블록203은 블록204에서 발생한 내부 생성 Carrier신호가 90도 위상 변환되도록 만드는 위상변환기이다.In general, the GNSS receiver is implemented in a manner that verifies the code phase delay hypothesis of all signals within a set Doppler frequency hypothesis. Therefore, in the primary discriminator 262,

The secondary discriminator 271 determines whether the currently tested code phase delay hypothesis is the last hypothesis. In other words,

Whether or not it is not

) Is sent to block 234 to test the next code phase (delay) hypothesis

The value is incremented by one more

Lt; RTI ID = 0.0 > 233 < / RTI > with the current code phase delay value. (For reference, the initial Doppler frequency and code phase delay hypothesis may be computed as specified in block 205

And

. Therefore, as shown in the signal generated at the current time t = nTs,

. ≪ / RTI > If the second discriminator

It is determined in block 272

And proceeds to block 234. At the same time,

, The process proceeds to the tertiary discriminator 281

If it is determined that

The process proceeds to block 235

Value by one more to start the signal search in the new Doppler frequency hypothesis. if

Then at block 282

And proceeds to block 235. [ Block 203 is a phase shifter that causes the internally generated Carrier signal generated at block 204 to be 90 degrees phase shifted.

As described above with reference to FIG. 2, since the conventional GNSS signal search unit performs signal correlation on each individual hypothesis indicating one Doppler frequency and one code phase delay, 21 Doppler frequencies and 2046 codes If phase delay is possible, we have to test all the hypotheses about 43000, and verify the detection parameters in each hypothesis.

3 shows a primary signal search unit (1 ^st StageSignalSearcher) according to a specific embodiment of the present invention. 2, the primary signal search unit detects a single Doppler frequency and code phase delay hypothesis pair every time a conventional GNSS receiver detects a single Doppler frequency hypothesis and a plurality of code phase delay hypotheses, ) Using a two-dimensional compression correlator capable of searching for a desired signal. In the present invention, the Doppler frequency and the code phase detected in accordance with the implementation of FIG. 3 are further approximated to provide a method of performing the secondary signal search unit of FIG. 4 additionally. According to the present invention, in the first and second signal search methods, the Doppler frequency and the code phase delay of the received signal are roughly found within the reduced hypothesis range, and then, And detects the accurate Doppler frequency and code phase delay, thereby greatly reducing the overall acquisition time of the GNSS signal (Acquisition Time). In the following description, since the plurality of blocks have the same functions as those of FIG. 2 while explaining the implementation of FIG. 3, the description of the same blocks is simplified as much as possible.

및

는 도 2의 입력신호와 동일하며 1차곱셈기(301, 302)와 위상변환기(303)의 기능은 각각 도 2에서 보인 1차곱셈기(201, 202)와 위상변환기(203)와 동일하다. 상기 1차곱셈기(301, 302)에 입력되어 수신되는 신호와 곱해지는 수신기 내부 발생 Carrier신호는 도 2와는 다르게 복합적Carrier신호(Combined carrier signal)이다. 도 3에서의 블록311과 블록312는 도 2에서 소개한 대역통과필터(BPF: 211 및 212)와 샘플러(ADC: 221 및 222)를 단순히 2개의 블록으로 축약하여 표현한 것이다. 따라서, 블록 311 및 312의 출력은

와

이며,

와

는 다시 2차곱셈기(331, 332)에서 수신기가 생성한 복합코드위상(combined code phase) 지연을 갖는 특정 위성 s의 복합코드신호 (Combined code phase signal) c(t)와 곱해진다. 상기 2차곱셈기(331, 332)의 출력은 각각

와

(또는 I와 Q)로 표시되며 누적기(341, 342)에서 N개의 시간적으로 연속된 2차곱셈기 출력이 누적되어 각각

와

를 발생한다. 상기

와

는 각각 제곱기(351, 352)에서 제곱되고 1차합산기(361)에서 합산되어 판단변수(Decision Variable)

를 만든다.First, the signals received on the I channel and Q channel

And

And the functions of the first-order multipliers 301 and 302 and the phase shifter 303 are the same as those of the first-order multipliers 201 and 202 and the phase shifter 203 shown in FIG. 2, respectively. 2, a mixed carrier signal is generated by multiplying the received signal by the first-order multipliers 301 and 302. Blocks 311 and 312 in FIG. 3 are simplified representations of the band-pass filters (BPFs 211 and 212) and samplers (ADCs 221 and 222) shown in FIG. Thus, the outputs of blocks 311 and 312

Wow

Lt;

Wow

Is multiplied again by a combined code phase signal c (t) of a specific satellite having a combined code phase delay generated by the receiver in the second-order multipliers 331 and 332. The outputs of the second-order multipliers 331 and 332 are

Wow

(Or I and Q) and N time-continuous sequential second-order multiplier outputs at accumulators 341 and 342 are accumulated

Wow

. remind

Wow

Are squared in squarers 351 and 352, summed in a first summing adder 361,

.

과 비교되어 Z값이

보다 큰 경우에는 GNSS 신호를 찾은 것으로 판단하여 이때의 탐지된 대략적인 도플러 주파수 값을 나타내는 인덱스

와 대략적인 코드위상지연을 나타내는 값

를 출력한다 (이 출력은 도 4에 도시하는 2차신호탐색부로 입력되어 세밀한 가설로 분해된 후 좀더 세밀한 신호 탐색을 위한 기준이 된다.) 상기 1차탐지임계치(Detection Threshold)

는 도 2의 탐지임계치와 같이 다양한 변수에 의하여 적절한 값이 결정된다. (반복 설명 생략) 만일, Z값이

보다 작은 경우에는 현재의 가설이 틀린 것으로 판단하고

또는

를 한 단계 더 증가시켜 새로운 복합 가설에서의 신호 탐색을 수행한다. 상기 1차판별기(362)에서 Z값이

보다 작은 것으로 판정되면, 2차판별기(371)에서 현재 시험된 복합코드위상지연 가설이 마지막 코드위상지연 가설(last code phase hypothesis)을 포함하는지 판별한다. 즉,

인지 여부를 판별하고 그렇지 않은 경우에는 블록 372에서

값을 K_t만큼 더 증가시키고 그 증가된

값을 코드위상지연 값으로 갖는 GNSS 위성의 PRN 코드신호를 PRN 신호발생기(373)에서 발생시킨다. 상기 PRN 신호발생기(373)의 출력과 그 출력이 다시 [1, 2, …, K_t-1]Ts만큼 시간적으로 지연된 PRN신호들이 모두 2차합산기(375)로 출력된다. 여기서 GPS와 같은 BPSK(Binary Phase Shift Keying)을 갖는 신호의 경우 상기 PRN신호들은 모두 갖은 부호로 더해져서 상기 2차합산기(375)의 출력 cc(t)는 c(n-k_t)+c(n-k_t-1)+c(n-k_t-2)+.. +c(n-k_t-K_t+1)로 표현할 수 있다. 이와는 다르게, Galileo의 BOC(n,n)과 같은 신호들은 각 시간 지연 값마다 부호가 바뀌어 더해져서 상기 2차합산기(375)의 출력 cc(t)는 c(n-k_t)-c(n-k_t-1)+c(n-k_t-2)-.. +(-1) ^{Kt -1}c(n-k_t-K_t+1)로 표현할 수 있다. 따라서, 상기 2차합산기(375)의 출력은 연속된 서로 다른 코드위상지연을 갖는 다수개의 PRN신호가 합산된 복합신호이다. 만일, 상기 2차판별기(371)에서

로 판별된 경우에는 블록 374에서

으로 초기화 하고 PRN신호발생기(373)로 진행한다. 또한, 동시에 3차판별기(281)로 진행하여

여부를 판별한다. 만일

인 경우에는 블록 3821, 3822 그리고 3823으로 진행하여 복합신호의 주파수 인덱스

의 값들을 모두

만큼 증가시키고 각각 4차판별기(3841, 3842, 3843)으로 진행하여

값들이

보다 작은지 판별한다. 만일

보다 큰 값을 갖는 주파수 인덱스가 있다면, 해당 도플러 주파수를 갖는 Carrier 신호는 발생하지 않는다. 따라서, 도시하는 바와 같이, 주파수 인덱스가

보다 작은 경우 블록3851 또는 블록3852 또는 블록3853로 진행하여 Carrier발생기(Carrier Generator)에서 입력된 주파수 인덱스 (

등)에 해당하는 Carrier신호들을 발생시킨다. 상기 블록3851 또는 블록3852 또는 블록3853에서 발생된 Carrier 신호들은 블록3862 및 블록3863에 도시된 바와 마찬가지로 블록3851의 출력 Carrier신호는 위상 증감없이 3차합산기(304)로 입력되고 블록3852의 출력 Carrier신호는 1차위상변환기(3862)에서 90도 위상감소 되어 3차합산기(304)로 입력되며 그 다음 Carrier신호는 180도 위상감소 되는 방식으로 위상변환 시켜 3차합산기(304)로 입력한다. 따라서, 블록3853의 출력 Carrier신호는 K_f차위상변환기(3863)에서 위상이 90*(

만큼 감소되어 3차합산기(304)로 입력된다. 결과적으로, 상기 3차합산기의 출력은 수학적으로

으로 표현될 수 있다. 참고로, 도 3에 도시하는 1차 신호탐색부의 초기 코드위상지연 및 도플러 주파수 값은 블록374와 블록3831~3833의 값을 따른다. 참고로, 도 3의 1차신호탐색부에서

,

의 값을 가질 수 있으므로 일반적인 도 2의 신호탐색부보다 최소 9배 이상의 빠른 탐색이 가능하다.The decision variable Z is set to a first detection threshold (1 ^st DetectionThreshold) at the first discriminator 362,

And the Z value is

It is determined that the GNSS signal is found, and an index indicating the detected approximate Doppler frequency value

And a value representing the approximate code phase delay

(This output is input to the secondary signal search unit shown in FIG. 4 and decomposed into a fine hypothesis and then becomes a reference for more precise signal search). The primary detection threshold (detection threshold)

The appropriate value is determined by various parameters such as the detection threshold of FIG. (Repeat description omitted) If the Z value is

It is judged that the current hypothesis is wrong

or

And the signal search in the new hybrid hypothesis is performed. If the Z value in the primary discriminator 362 is

The secondary discriminator 371 determines whether the currently tested composite code phase delay hypothesis includes the last code phase hypothesis. In other words,

And if not, at block 372

Lt; RTI ID = 0.0 _> Kt _{< /} RTI >

The PRN code signal of the GNSS satellite having the value of the code phase delay value is generated in the PRN signal generator 373. [ The output of the PRN signal generator 373 and its output are [1, 2, ... , K _t -1] Ts are all output to the quadratic adder 375. In the case of a signal having BPSK (Binary Phase Shift Keying) such as GPS, the PRN signals are added to all of the codes so that the output cc (t) of the quadratic adder 375 is c (nk _t ) + c _t- 1) + c (nk _t -2) + .. + c (nk _t -K _t +1). Alternatively, the signal, such as a BOC (n, n) are summed in the Galileo codes are changed for each time delay value output cc (t) of said second summer (375) is c (nk _t) -c _(t nk -1) + c (nk _t -2) - .. + (- 1) ^{Kt -1} c (nk _t -K _t +1). Thus, the output of the quadrature summer 375 is a composite signal of a plurality of PRN signals summed with successive different code phase delays. If the secondary discriminator 371

It is determined in block 374

And proceeds to the PRN signal generator 373. At the same time, the process proceeds to the tertiary discriminator 281

. if

The process proceeds to blocks 3821, 3822 and 3823, and the frequency index of the composite signal

All of the values

And proceeds to the fourth order discriminators 3841, 3842, and 3843, respectively,

The values

. if

If there is a frequency index having a larger value, a carrier signal having the corresponding Doppler frequency does not occur. Therefore, as shown in the figure,

The process proceeds to block 3851, block 3852 or block 3853, and the frequency index ("

And so on). The carrier signals generated in the block 3851 or the block 3852 or the block 3853 are input to the tertiary adder 304 without increasing or decreasing the phase of the output carrier signal of the block 3851 as shown in the blocks 3862 and 3863, The signal is phase-reduced by 90 degrees in the first-order phase shifter 3862, input to the third-order adder 304, and then phase-shifted in such a manner that the carrier signal is phase-reduced by 180 degrees and input to the third-order adder 304 . Therefore, the output signal of the Carrier Blocks 3853 K _f is the phase difference from the phase shifter (3863) 90 * (

And input to the third order summing unit 304. [ As a result, the output of the third order summer is mathematically

. ≪ / RTI > For reference, the initial code phase delay and Doppler frequency values of the primary signal search unit shown in FIG. 3 follow the values of block 374 and blocks 3831-3833. For reference, in the primary signal search unit of FIG. 3

,

, It is possible to search at least 9 times faster than the conventional signal search unit of FIG.

=500Hz 그리고 1msec의 상관길이를 예를 들었으므로,

만큼의 상대적 도플러 주파수 차이를 가지고 있는 상기 Carrier발생기(3851~3853)에서 발생된 이웃하는 Carrier신호들은, 상기 누적기(341, 342)에서 누적되는 출력신호에 90^o씩의 위상 증가가 발생한다. 이를 보상하고 Z_I,Z_Q의 크기가 최대가 되도록 하기 위하여 상기 위상변환기(3862 및 3863)를 사용하는 것이다. 따라서, 만일 상기 Carrier발생기(3851~3853)에서 발생된 이웃하는 Carrier신호들의 상대적 도플러 주파수 편차

와 상기 누적기(341, 342)의 시간적 누적길이의 곱이 본 발명의 예제(각각 500Hz, 1msec)와는 달리 0.5가 되지 않는다면 상기 위상 변환기(3862, 3863)에서의 위상 증감도 적절하게 달라져야 한다.The reason why the phase converters 3862 and 3863 are used for the outputs of the carrier generators 3851 to 3853 is that the carrier signals generated by the carrier generators 3851 to 3853 are accumulated in the accumulators 341 and 342 To compensate for the average phase change that occurs. In the present invention,

= 500 Hz and a correlation length of 1 msec is given as an example,

Neighboring carrier signals generated from the carrier generators 3851 to 3853 having a relative Doppler frequency difference as much as 90o of the output signals accumulated in the accumulators 341 and 342 are phase-shifted by 90 ^° . And the phase converters 3862 and 3863 are used to compensate for this and to maximize the magnitude of Z _I and Z _Q. Therefore, if the relative Doppler frequency deviation of the neighboring carrier signals generated by the carrier generators 3851 to 3853

And phase accumulation lengths of the accumulators 341 and 342 do not become 0.5, unlike the examples of the present invention (500 Hz, 1 msec), the phase shifts in the phase converters 3862 and 3863 must be appropriately changed.

Figure 4 shows a second signal search section ((2 ^nd StageSignalSearcher)) presented in this invention. The basic structure of the secondary signal search unit is very similar to the implementation structure of the signal search unit of the general GNSS receiver described in FIG. For example, blocks 401, 402, 403, 404, 411, 412, 421, 422, 431, 432, 433, 441, 442, 451, 452, 461, 462 of FIG. , 203, 204, 211, 212, 221, 222, 231, 232, 233, 241, 242, 251, 252, 261 and 262, respectively.

과

)을 입력 받아 그 값을 각각

및

로 치환하고

과

가 나타내는 도플러 주파수와 코드위상지연 가설 영역에서의 신호탐색을 수행한다. 이때, 입력된

과

가 나타내는 가설 영역은 주파수 인덱스

및 코드위상지연 인덱스

로 표현할 때,

및

이므로, 블록405에서 도시하는 바와 같이 초기 도플러 주파수와 코드위상지연 가설 인덱스는 각각

와

로 지정된다. 블록404는 주어진 초기 도플러 주파수 가설(

에 해당하는 Carrier신호를 생성하여 1차곱셈기(401)에 공급하며, 같은 Carrier신호가 위상변환기(403)을 통과하여 90도 위상증가 된 후 1차곱셈기(402)로 공급된다. 블록433은 주어진 초기 코드위상지연 가설(

)에 따른 코드위상지연(code phase)를 갖는 PRN코드신호 c^s[n-

]를 발생하여 2차곱셈기(431 및 432)에 공급한다. 이렇게 하여 얻어진 판단변수 Z는 도2에서 도시한 탐지임계치(Detection Threshold)와 동일한 값

과 비교될 수 있는데, 이는 2차신호탐색부(도 4)는 도 2에 도시한 신호탐색부와 동일한 상관 방식을 갖기 때문이다. 즉, 2차신호탐색부는 일반적인 GNSS 수신기가 갖는 신호탐색부와 동일한 구현방식을 갖기 때문에 탐색 성능도 동일하므로 같은 탐지임계치를 사용할 수 있다. (도 4의 2차신호탐색부와 도 2의 신호탐색부가 다른 점은 단지 도 4의 2차신호탐색부는 주어진 좁은 2차원 가설영역에서만 신호를 탐색한다는 것이다.) 1차판별기 (462)에서

으로 판별이 난 경우에는 현재 검증한 도플러 주파수와 코드위상지연 값 (

과

)을 최종 탐색결과로 출력하고

로 판별이 난 경우에는 블록471로 진행하여 현재의 코드위상지연

가 최대치인지를 파악한다. 따라서,

인 경우에는 블록434로 진행하여

를 1만큼 증가시키고 블록 433에서 증가된

값에 해당하는 PRN코드신호가 발생되도록 한다. 만일

인 경우에는 블록 472로 진행하여

로

를 초기화 시키고 동시에 3차판별기(481)로 진행하여 현재의 도플러 주파수

가 주어진 가설 범위의 최대치인지를 파악한다. 따라서,

인 경우에는 블록482로 진행하여

를 1만큼 증가시키고 오실레이터(404)에서 증가된

값에 해당하는 Carrier신호가 발생되도록 한다. 만일,

인 경우에는 모든 가설 영역에서의 탐색을 완료한 것이므로 탐색을 종료한다.The secondary signal search unit shown in FIG. 4 outputs the output of the primary signal search unit of FIG. 3

and

) And inputs the values

And

≪ / RTI &

and

And the signal phase of the code phase delay hypothesis area. At this time,

and

The hypothesis area represented by the frequency index

And code phase delay index

When expressed as,

And

The initial Doppler frequency and the code phase delay hypothesis index are

Wow

. Block 404 illustrates a given initial Doppler frequency hypothesis

And supplies the carrier signal to the first-order multiplier 401. The same carrier signal passes through the phase shifter 403 and is phase-increased by 90 degrees and then supplied to the first-order multiplier 402. [ Block 433 illustrates a given initial code phase delay hypothesis < RTI ID = 0.0 >

) PRN code signal having a code phase delay (code phase) in accordance with c ^s [n-

To the second-order multipliers 431 and 432, respectively. The judgment variable Z thus obtained is the same value as the detection threshold shown in FIG. 2

Because the secondary signal search unit (FIG. 4) has the same correlation scheme as the signal search unit shown in FIG. 2. That is, since the secondary signal search unit has the same implementation scheme as that of the signal search unit of a general GNSS receiver, the same detection threshold can be used because the search performance is the same. (The secondary signal search unit of FIG. 4 differs from the signal search unit of FIG. 2 only in that the secondary signal search unit of FIG. 4 searches signals in a given narrow two-dimensional hypothetical region only.) In the primary decision unit 462

, The presently verified Doppler frequency and the code phase delay value (

and

) As a final search result

The process proceeds to block 471 where the current code phase delay

Is the maximum value. therefore,

The process proceeds to block 434

Lt; RTI ID = 0.0 > 433 < / RTI >

So that a PRN code signal corresponding to the value is generated. if

The process proceeds to block 472

in

And proceeds to the tertiary discriminator 481 to determine the current Doppler frequency

Is the maximum value of the given hypothesis range. therefore,

The process proceeds to block 482

Lt; RTI ID = 0.0 > 1 < / RTI >

So that a corresponding carrier signal is generated. if,

, The search is completed because the search in all the hypothesized areas has been completed.

과

)를 얻는다. 이후, 블록502에서는 입력된 대략적인 정보로부터 상세 탐색을 수행할 가설영역(

및

)을 찾아내어 블록503으로 전달하고 블록503에서는 도 4에서 도시하는 2차신호탐색부를 수행하여 정확한 도플러 주파수와 코드위상지연 정보(

과

)를 최종 출력한다.FIG. 5 shows a simplified operation of the primary signal search unit and the secondary signal search unit according to the present invention. First, in block 501, the primary signal search unit shown in FIG. 3 is performed to obtain a rough Doppler frequency and code phase delay information

and

). Then, at block 502, a hypothesis area (

And

), And transfers it to the block 503. In the block 503, the secondary signal search unit shown in FIG. 4 is performed to obtain an accurate Doppler frequency and code phase delay information

and

).

The above-described compression hypothesis technique can be used not only for signal acquisition but also for signal tracking.

6 shows an example of a code signal tracking apparatus using a compression correlator. The code signal tracking device of FIG. 6 is the result of applying blocks 373 and 375 of FIG. 3 to a code phase tracker.

After the output of the Early Compressed Code generation block 601 is multiplied by the received signal and integrated (605), the value determined in the Code Loop Discriminator (604) And enters the input of the generator 611 to generate a line-of-sight time code signal (a code signal that is ahead of the currently received code estimate value by a specific phase).

The output of the Late Compressed Code generation block 602 is also multiplied by the received signal and integrated (606), and the value determined by the code loop discriminator 604 is input to the post-visual code generator 612 And generates a later time code signal (a code signal delayed by a specific phase from the currently received code estimation value).

The present code generation block 603 generates a current code phase estimation value of the received signal, which is also multiplied by the received signal and integrated (607) and output as a result of the tracker.

At this time, in the block diagram of a line time compression code generation unit (601) n is an odd number more than 2, and α _i (1≤i≤n) is the real or imaginary non-zero delay elements D _i (1≤i≤n) is 1 chip. In the block of the post-compression code generation unit 602, n is an odd number of 2 or more, α _i (1≤i≤n) is a real number or imaginary number other than 0, and the delay element D _i (1≤i≤n) .

The linear-time-compression-code generating unit 601 multiplies each of the signals delayed from the early code sequence generator 611 by each of the specific coefficients 621 and adds them (631) do. (1) when P _P is the output of the current time code generation unit 603 and P _E is the output (output of 631) of the linear time compression code generation unit 601 and D = D _i = 1 chip, It is as follows.

Each of the signals delayed from the late code sequence generator 612 is multiplied by a respective specific coefficient 622 and added (632) do. (2) when P _P is the output of the current time code generation unit 603 and P _L is the output of the post-view compression code generation unit 602 (output of 632) and D = D _i = 1 chip, It is as follows.

From equations (1) and (2), when the output of the line-of-sight code generator 611 is P _e and the output of the current time code generator 603 is P _P , the relational expression is as shown in Equation (3).

Also, when the output of the post-time code generator 612 is P ₁ and the output of the current time code generator 603 is P _P , the relational expression is as shown in Equation 4.

In the block diagram of a preferred example of a linear time line combination generate compression code 601 proposed in the present invention, n is 3, _{α 1 = 1, α 2 =} 2, and _{α 3 = 1, D i (} 1≤i≤ n) becomes one chip. In the block of the post-visual-compression-code generation unit 602, n is 3, and α ₁ = 1, α ₂ = 2, α ₃ = 1, and D _i (1≤i≤n) is 1 chip.

Figure 7 compares the code tracking device discriminator value and the existing E-L correlator discriminator value according to the embodiment of the code tracking device using the compression correlator. As shown in FIG. 7, when the code tracking device using the compression correlator is applied, it can be seen that the trackable linear section is wider than that of the existing E-L correlator, and tracking in the high-speed state is easy.

As described above, according to the embodiment of the present invention, since it is possible to search and acquire signals several times faster than the receiver technology of the conventional GNSS and spread spectrum communication system, the hardware of the receiver for signal search and signal acquisition is consumed very little, It is possible to implement an optimal receiver in which the time and hardware cost are minimized.

The methods according to embodiments of the present invention may be implemented in the form of program instructions that can be executed through various computer means and recorded in a computer-readable medium. The computer-readable medium may include program instructions, data files, data structures, and the like, alone or in combination. The program instructions recorded on the medium may be those specially designed and constructed for the present invention or may be available to those skilled in the art of computer software. Furthermore, the above-described file system can be recorded on a computer-readable recording medium.

While the invention has been shown and described with reference to certain preferred embodiments thereof, it will be understood by those of ordinary skill in the art that various changes in form and details may be made therein without departing from the spirit and scope of the invention as defined by the appended claims. This is possible.

Therefore, the scope of the present invention should not be limited to the described embodiments, but should be determined by the equivalents of the claims, as well as the claims.

Claims (12)

In a GNSS and spread-spectrum signal acquisition apparatus,
A signal search section for searching for a received signal of at least one of a GNSS positioning signal and a spread spectrum signal,
Lt; / RTI >
Wherein the signal search unit comprises:
And a compression correlator for compressing a part of all the frequency and code phase hypotheses for searching for the received signal to search for a combined hypothesis, and includes a first and second stage signal search unit,
The primary signal search unit finds the position of the hypothesis of the actual signal from the compression hypothesis within the reduced hypothesis range using the compression correlator,
Finding a correct hypothesis of the actual signal within the reduced hypothesis range using a correlator without compression in the secondary signal searcher
And the GNSS and spread-spectrum signal acquisition apparatus. The method according to claim 1,
Wherein the individual hypotheses searched by the secondary signal search unit include a plurality of individual hypotheses included in the compound hypothesis searched by the primary signal search unit
And the GNSS and spread-spectrum signal acquisition apparatus. The method according to claim 1,
The compression correlator comprises:
A plurality of frequency signals corresponding to consecutive individual frequency hypotheses are generated, and the composite carrier signal generated by linearly combining the plurality of frequency signals,
A plurality of code signals corresponding to successive individual code phase hypotheses are generated and linearly combined to generate a composite code signal
Linear combination to generate a composite signal corresponding to a complex hypothesis,
Correlating the generated composite signal with the received signal
And the GNSS and spread-spectrum signal acquisition apparatus. The method according to claim 1,
The compression correlator comprises:
A plurality of frequency signals corresponding to consecutive individual frequency hypotheses are generated, and the composite carrier signal generated by linearly combining the plurality of frequency signals,
(N, n) signals by linearly combining the generated odd-numbered code signals, linearly combining even-numbered code signals, and then linearly combining the odd-numbered code signals with the opposite signs to generate a plurality of code signals corresponding to successive individual code phase hypotheses. A composite signal corresponding to the hypothesis is generated,
Correlating the generated composite signal with the received signal
And the GNSS and spread-spectrum signal acquisition apparatus. The method according to claim 1,
Wherein the signal search unit comprises:
A first discriminator for comparing the output of the compression correlator with a first threshold value to determine whether to detect a signal,
And a second discriminator for comparing the outputs of the correlator for a plurality of individual hypotheses included in the hybrid hypothesis with the second threshold in the secondary signal search unit to determine whether to detect a signal,
While performing the primary signal search for all Doppler frequencies and all code phases
Performing the secondary signal search unit when there is signal detection in the first discriminator
And the GNSS and spread-spectrum signal acquisition apparatus. A plurality of signal search units for searching for a received signal of at least one of a GNSS positioning signal and a spread spectrum signal,
Lt; / RTI >
The plurality of signal search units
A compression correlator for searching for a composite hypothesis by compressing a part of the mode frequency and code phase hypotheses for searching the received signal
A plurality of code phase hypotheses are searched at one time,
And a signal search unit of two stages,
The primary signal search unit drives the plurality of compression correlators in parallel to find a hypothetical position of an actual signal within a reduced hypothesis range,
Finding a correct hypothesis of the actual signal within the reduced hypothesis range using a correlator in a secondary signal searcher
And the GNSS and spread-spectrum signal acquisition apparatus. The method according to claim 6,
The individual hypotheses searched by the secondary signal search unit include a plurality of individual hypotheses included in the compression hypothesis searched in the compression correlator that outputs the highest correlation result in the primary signal search unit
And the GNSS and spread-spectrum signal acquisition apparatus. The method according to claim 6,
The compression correlator comprises:
A plurality of frequency signals corresponding to consecutive individual frequency hypotheses are generated, and the composite carrier signal generated by linearly combining the plurality of frequency signals,
A plurality of code signals corresponding to successive individual code phase hypotheses are generated and linearly combined to generate a composite code signal
Linear combination to generate a composite signal corresponding to a complex hypothesis,
Correlating the generated composite signal with the received signal
And the GNSS and spread-spectrum signal acquisition apparatus. The method according to claim 6,
The compression correlator comprises:
A plurality of frequency signals corresponding to consecutive individual frequency hypotheses are generated, and the composite carrier signal generated by linearly combining the plurality of frequency signals,
(N, n) signals by linearly combining the generated odd-numbered code signals, linearly combining even-numbered code signals, and then linearly combining the odd-numbered code signals with the opposite signs to generate a plurality of code signals corresponding to successive individual code phase hypotheses. A composite signal corresponding to the hypothesis is generated,
Correlating the generated composite signal with the received signal
And the GNSS and spread-spectrum signal acquisition apparatus. The method according to claim 6,
Wherein the signal search unit comprises:
A first discriminator for comparing a correlation result of the largest magnitude from all outputs of the plurality of compression correlators driven in parallel in the primary signal search unit and a first threshold value to determine whether to detect a signal,
And a second discriminator for comparing the outputs of the correlator for a plurality of individual hypotheses included in the compression hypothesis searched by the secondary signal searcher with a second threshold value to determine whether to detect a signal
While performing the primary signal search for all Doppler frequencies and all code phases
Performing the secondary signal search unit when there is signal detection in the first discriminator
And the GNSS and spread-spectrum signal acquisition apparatus. A signal tracking unit for tracking at least one received signal of a GNSS positioning signal or a spread spectrum signal,
Lt; / RTI >
The signal-
A line-time compression code correlator for performing a correlation with a line-time code signal preceding by a specific phase with respect to a currently received code signal; And tracking the received signal using a post-temporal compression code correlator that performs a correlation with the time code signal after being delayed by a certain phase relative to the currently received code signal
/ RTI > and the spread spectrum signal tracking device. 12. The method of claim 11,
The linear time code signal and the back time code signal are generated by linearly combining a plurality of continuous individual code phase signals
/ RTI > and the spread spectrum signal tracking device.

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