WO2013115481A1 - Compressed sensing method and device for quickly obtaining global navigation satellite system (gnss) and spread spectrum signals - Google Patents

Abstract

Disclosed are a compressed sensing method and device for quickly obtaining global navigation satellite system (GNSS) and spread spectrum signals. The device for obtaining GNSS and spread spectrum signals includes: a signal searching unit for searching either the GNSS signal or the spread spectrum signal, wherein the signal searching unit includes a signal generator for generating a compressed sensing matrix with a size of M × L by using a correlation matrix with a size of L × L and a measuring matrix and a size of M × L; a parallel correlator comprising M correlators arranged in parallel, for performing correlation between a received signal and the compressed sensing matrix; and a decoding module for obtaining a Doppler frequency index and a code phase of the received signal by using the compressed correlation of the parallel correlator.

Description

Compression detection method and device therefor for fast signal acquisition of GS and spread signals

Embodiments of the present invention relate to fast signal acquisition of Global Navigation Satellite System (GNSS) and Spread Spectrum signal receivers. In particular, fast signal search and search using compression sensing techniques. and acquisition methods and apparatus.

GNSS collectively refers to a positioning system using a satellite network such as GPS, and measures the time of arrival (TOA) of a radio wave from an antenna of the satellite and arrives at the GPS receiver. This is to calculate the relative position of the GPS receiver by locating the satellites. All GNSS satellites transmit signals to the ground which are spread spectrum signals used in general wireless communication systems. The first conditions for starting up the receivers of all communication and satellite navigation systems using spread signals are the search for satellite signals and the acquisition of satellite information.

In the initial position acquisition of the GPS receiver, the case where the GPS receiver is powered on again after a long power-off state is performed to perform the initial position measurement is called cold start. This time is called TTFF (Time to first fix). The cold start of a GPS receiver to detect GPS L1 frequency C / A signals takes several steps, in particular the code phase of the satellite signal (corresponding to the temporal delay of the signal), the hypothesis range and the Doppler frequency. (Doppler frequency) Searching for a hypothesis region requires the most hardware complexity. In other words, the GPS L1 frequency C / A code consists of 1023 codes, so the code phase to be searched is 2046 in 0.5 chip units, and for ground pedestrian GPS receivers, it is generally Doppler every 500 Hz from -5KHz to + 5KHz. Since the frequency hypothesis is set, a total of 21 Doppler hypotheses are generated, and a total of about 42000 hypotheses must be verified. One hypothesis verification is usually performed through a correlator with a correlation length of 1 msec, so a receiver using one correlator requires approximately 42 seconds of signal search time and a receiver using 42000 correlators within approximately 1 msec. You can explore The reason why a GPS receiver should search for and acquire a C / A signal is because the PRN code of the GPS satellite signal received at the present moment and the exact code synchronization and the Doppler frequency of the GPS satellite. To keep track of them without losing them. The receiver of a general spread spectrum communication system also performs a signal search for each code phase hypothesis in order to achieve code synchronization.

Figure 1 shows all code phase hypotheses (2046 total) and all Doppler frequency hypotheses (21 total in 500 Hz increments from -5 KHz to +5 KHz), a total of about 42000 hypotheses for a typical GPS receiver to search for one satellite signal. , The result of searching all of them. The x-y plane of FIG. 1 is about 42000 hypothesis planes, and the output value of the correlator is shown on the Z axis for each point (each hypothesis) on the hypothesis plane. The output of the correlator is a value obtained by operating a correlator having a correlation length of 1 msec for each hypothesis (= code and Doppler frequency hypothesis), and using a correlator having a longer correlation length to detect a weak signal. If so, the Doppler frequency hypothesis should be searched in smaller units.

The process of performing signal search and acquiring the detected signal by verifying many hypotheses as described above is very inefficient because it uses a large amount of hardware resources at the beginning of the receiver's startup. Various techniques have been developed to reduce this inefficiency or make signal detection faster. The simplest technique is to utilize multiple parallel correlators. For example, 42000 parallel correlators complete signal search quickly in less than 1 msec, but hardware inefficiency is high because only a large number of correlators are used for one time. Another method is to perform a fast correlation by multiplying the received signal with the PRN code generated in the receiver in the frequency domain using the Fast Fourier Transform (FFT). Since all code signals generated in s have to be Fourier-transformed into the frequency domain, and the result multiplied in the frequency domain must be inverse Fourier transformed into the time domain, the computational load is very high. Such a high computational FFT-based signal search method can be implemented using a high speed DSP chip, but there is a problem of high cost due to the DSP chip.

Another method is Qualcomm Inc.'s Assisted GPS, or collectively Assisted GNSS (A-GPS) technology. 2 shows a comparison of time to first fix (TTFF) in cold start in an A-GPS receiver and a conventional GPS receiver. As shown in FIG. 2, the A-GPS receiver may finish the TTFF in a few seconds with assistance of an A-GPS server connected to the mobile communication base station. A-GPS has a significant performance improvement over the existing GPS technology, but has some limitations and problems. First, since help information must be obtained from a server connected to a mobile communication network, it can be implemented only when mobile communication and wireless connection are possible, and the code phase range that the GPS receiver must search is based on the time synchronization between the base station and the mobile phone and the location of the mobile phone. It depends on the accuracy. Therefore, in the case of Asynchronous Cellular Networks such as 3rd Generation or 3.5th Generation and 4th Generation, the base station time is inaccurate and the size of the code phase search area becomes the entire 1023 chip which is the search area of the existing GPS receiver. GPS performs the same signal search as conventional GPS.

One of the major breakthroughs in recent signal processing technologies is the Compressed Sensing technology, which obtains a large amount of data and extracts a very small amount of data from a conventional method of extracting information contained in a signal. FIG. 2 is a technology capable of extracting information contained in a signal. Compression detection technology is currently used in many systems for image processing and processing large amounts of data.However, the technique of obtaining a measurement value from a compressed signal uses a random measurement matrix so that the original signal is obtained from the measurements obtained. The amount of computation required to restore is very high and is not used as a real time system. In addition, signal detection and acquisition techniques have not been developed in spread spectrum systems using compression sensing techniques.

The present invention proposes a technique for minimizing receiver hardware and speeding up signal acquisition by utilizing a compression sensing technique in signal acquisition and signal tracking of a GNSS and a spread spectrum communication receiver.

The present invention proposes to implement an optimal compressed sensing scheme for a GNSS and a spread spectrum receiver to minimize the hardware, so that the correlator required for the initial signal search of a communication terminal using a conventional GNSS receiver or a spread spectrum signal is provided. It aims to reduce the number of hardware resources required for signal detection by reducing the number of times.

The second aim of the present invention is to maintain or increase the sensitivity in GNSS signal acquisition and signal tracking.

The third object of the present invention is to have a wider linear tracking area in the signal tracker (DLL) of the GNSS and spread spectrum receivers, making it more robust to noise and less signal lost by multipath signals.

In addition, the present invention provides an implementation method capable of rapid signal detection by presenting a structural measurement matrix without using a random measurement matrix used in a general compression detection technique.

According to an aspect of the present invention, there is provided a signal search unit for searching for at least one received signal of a Global Navigation Satellite System (GNSS) positioning signal or a spread spectrum signal, wherein the signal search unit comprises a multi-stage As it increases, the compression of the hypothesis decreases and is searched.

Each multi-stage signal searcher can be implemented in two ways, one of which is as follows.

According to an aspect of the present invention, a global navigation satellite system (GNSS) and a spread spectrum signal acquisition apparatus includes a signal search unit for searching at least one received signal of a GNSS positioning signal or a spread spectrum signal, the signal Each stage of the search section uses a correlation matrix having a matrix size of [L × L] and a measurement matrix having a matrix size (M <L) of [M × L] where the number of rows is smaller than the number of columns. A signal generator for generating a compression sensing matrix having a matrix size of]; A parallel correlator having M correlators configured in parallel to perform correlation between the received signal and the compression sensing matrix; And a decoding module for obtaining a code phase and a Doppler frequency of the received signal by using the compression correlation result of the parallel correlator.

The signal generator may generate the compression sensing matrix by matrix multiplication of the correlation matrix and the measurement matrix.

Each row of the correlation matrix may be a spreading code having a code delay of an arbitrary time unit and the same as a spreading code signal of the received signal.

The parallel correlator may generate the compression correlation result having a matrix size of [M × 1] by performing correlation between the M rows constituting the compression sensing matrix and the received signal.

Each row of the compression correlation result is obtained by multiplying each of the L columns constituting the measurement matrix by the product of the correlation matrix and the received signal by a coefficient, and as a result, may be a linear sum of all columns of the measurement matrix. have.

The decoding module inputs M rows of the compression sensing matrix to the parallel correlator for all Doppler frequency hypotheses, obtains the M compression correlation results from the parallel correlator, and then obtains the M compression correlation results obtained from the parallel correlator. From the Doppler frequency and the code phase of the received signal can be obtained.

And the first signal generator to the signal generator by using the measurement matrix having a matrix size of [M ₁ × L] generating a first compressed sensing matrix having a matrix size of _{[M 1 × L], [} M 2 × L] (where M ₁ + M ₂ = M), using a measurement matrix having a matrix size of [M ₂ × L] to a second order signal generator that generates a second order compression sensing matrix having a matrix size of [M ₂ × L]. And a parallel correlator comprising a M ₁ correlator, a _first parallel correlator configured to receive M ₁ rows of the _first compression sensing matrix, and perform correlation with the received signal; and M ₂ correlators. And a second order parallel correlator configured to receive M ₂ rows of the second order compression sensing matrix and perform correlation with the received signal.

The decoding module estimates the Doppler frequency closest to the received signal from the M ₁ compression correlation result obtained for all Doppler frequency hypotheses in the first parallel correlator, and then estimates the estimated Doppler frequency in the second parallel correlator. The final code phase of the received signal can be obtained from the M ₂ compression correlations obtained for the received Doppler frequencies.

According to another aspect of the present invention, a method for obtaining GNSS and spread spectrum signals for finding at least one received signal among a GNSS positioning signal or a spread spectrum signal and obtaining a code phase and a Doppler frequency index of the received signal is described in [L × L]. A compression sensing matrix having a matrix size of [M × L] is generated by using a correlation matrix having a matrix size and a measurement matrix having a matrix size (M <L) of [M × L] where the number of rows is smaller than the number of columns. Doing; Performing correlation between the received signal and the compression sensing matrix through M correlators configured in parallel for each Doppler frequency hypothesis; And obtaining a Doppler frequency of the signal received from the correlation result and obtaining a code phase of a spreading code.

Another implementation method of the multi-stage signal search unit is as follows.

Reduced Doppler frequency range predicted by the received signal by performing a single correlation with the received signal on a complex hypothesis including a plurality of Doppler frequencies and a plurality of code phase delays And a primary signal searcher for obtaining an initial code phase delay range; And a final Doppler frequency value and code corresponding to the received signal through correlation with the received signal for each of the Doppler frequency hypothesis and the code phase delay hypothesis belonging to a predetermined hypothesis region including the initial Doppler frequency range and the code phase delay range. Provided are a GNSS and a spread spectrum signal acquisition device including a secondary signal searcher for obtaining a phase delay value.

The primary signal search unit uses the sum of the carrier signals corresponding to each of the plurality of Doppler frequency hypotheses and the sum of the code signals corresponding to each of the plurality of code phase delay hypotheses to determine the determination variable for the complex hypothesis. After the decision variable is obtained, when the determination variable is larger than the detection threshold, the received signal is determined to be found, and the corresponding Doppler frequency range and code phase delay range are determined by the initial Doppler frequency range and the initial code phase delay. You can output it as a range.

The primary signal search unit may include a complex carrier signal obtained by phase-converting a carrier signal corresponding to each of a plurality of Doppler frequency hypotheses (compound Doppler frequency hypotheses) and a plurality of code phase delay hypotheses (compound code phase delay hypotheses). Correlation between the complex correlation signal according to the complex code signal obtained by summing corresponding code signals and the received signal may be performed.

The detection threshold may include: Coherent Correaltion Length, Non-Coherent Accumulation Length, Probability of Detection, and Probability of False Alarm. ) May be determined by at least one of the Doppler frequency hypothesis intervals.

The primary signal searcher may perform a signal search for a new complex hypothesis by increasing the Doppler frequency or code phase delay when the determination variable is smaller than the detection threshold.

The secondary signal searcher may further refine hypotheses belonging to the initial Doppler frequency range and the initial code phase delay range (the Doppler frequency hypotheses belonging to the initial Doppler frequency range and the code phase delay hypotheses belonging to the initial code phase delay range, respectively). After the decision variables for each pair of hypotheses are obtained, the received Doppler frequency value and the code phase delay value are determined by determining that the received signal is found when the decision variable is larger than a detection threshold. It can output as frequency value and code phase delay value.

According to another aspect of the present invention, a method of obtaining a code phase (delay) and a Doppler frequency of the received signal by finding at least one received signal of a GNSS positioning signal or a spread spectrum signal may include a plurality of Doppler frequencies and a plurality of code phases. A first signal search step of obtaining a reduced initial Doppler frequency range and an initial code phase delay range predicted by the received signal through correlation with the received signal with respect to a complex hypothesis including (delay); And the Doppler frequency hypothesis and the code phase delay hypothesis of the predetermined hypothesis range within the predetermined hypothesis range including the initial Doppler frequency range and the initial code phase delay range through correlation with the received signal. And a second signal search step of obtaining a final Doppler frequency value and a code phase delay value.

According to another aspect of the present invention, the GNSS and spread spectrum signal acquisition apparatus includes a signal search unit for searching for at least one received signal of a Global Navigation Satellite System (GNSS) positioning signal or a spread spectrum signal, The signal search unit is configured to perform a single correlation with the received signal on a complex hypothesis including a plurality of Doppler frequencies and a plurality of code phase delays. A primary signal searcher for obtaining a reduced Doppler frequency range) and an initial code phase delay range; And a final Doppler frequency value and code corresponding to the received signal through correlation with the received signal for each of the Doppler frequency hypothesis and the code phase delay hypothesis belonging to a predetermined hypothesis region including the initial Doppler frequency range and the code phase delay range. It may include a secondary signal searcher for obtaining a phase delay value. In this case, the primary signal searcher may perform a signal search through a compressed delay lock loop for a complex code signal in which code signals corresponding to each of a plurality of code phase delay hypotheses are added together.

According to an embodiment of the present invention, a receiver for signal search and signal acquisition is possible because it does not require a communication device like the existing A-GPS and can perform signal search and acquisition several times faster than the receiver technology of the conventional GNSS and spread spectrum communication system. Very little hardware is used, which can be optimized for time and cost.

According to an embodiment of the present invention, a two-dimensional compression correlator can be applied to a code phase tracker and a phase tracker for signal tracking to have a wider linear characteristic, thereby realizing a system robust to instantaneous signal shaking due to multipath and noise. .

FIG. 1 shows a result of a typical GPS receiver searching a two-dimensional GPS signal hypothesis search region consisting of all code phase hypotheses and all Doppler frequency hypotheses in order to search for one satellite signal.

2 is a comparative analysis of the time required for initial position detection at cold start of a GPS receiver in Qualcomm's A-GPS and a conventional GPS method.

FIG. 3 is a block diagram illustrating an internal configuration of a signal search unit based on a structured compression detection technique according to an embodiment of the present invention.

FIG. 4 is a block diagram illustrating an internal configuration of a signal search unit based on a structured compression sensing technique according to another embodiment of the present invention.

5 illustrates an auto-correlation function (ACF) of a general GPS and Galileo signal.

6 is a block diagram illustrating an internal configuration of a receiver signal search unit using a conventional GNSS correlator.

FIG. 7 is a block diagram illustrating an internal configuration of a primary signal search unit in a receiver based on a 2D compression correlation technique according to an embodiment of the present invention.

FIG. 8 is a block diagram illustrating an internal configuration of a secondary signal searcher in a receiver based on a 2D compressed correlation technique according to an embodiment of the present invention.

9 is a flowchart illustrating a brief operation of the primary signal search unit and the secondary signal search unit according to an embodiment of the present invention.

FIG. 10 is a block diagram of implementing a code tracker based on a 2D compression correlation technique according to an embodiment of the present invention.

11 illustrates a comparison between a code tracking device discriminator value and a conventional E-L correlator discriminator value according to an embodiment of a code tracking device using a compression correlator.

Hereinafter, exemplary embodiments of the present invention will be described in detail with reference to the accompanying drawings. However, embodiments of the present invention may be modified in various other similar forms, and the scope of the present invention is not limited to the embodiments described below. For example, Global Navigation Satellite Systems (GNSS) are all types of satellite navigation systems, including the Global Positioning System (GPS) in the United States (GPS in the United States, Glonass in Russia, Galileo in Europe, Beidou in China, and QZSS in Japan). Is referred to generically.

In the present invention, the GNSS is represented by GPS. However, since all GNSSs use spread spectrum signals in the same manner as the GPS, the implementation examples of all GNSS receivers are similar to the implementation examples of the GPS receiver, so unless specifically distinguished. Use the same meaning. In addition, since the receiver of a mobile communication system using a spread spectrum signal performs the same function, the field to which the present invention is applied is utilized in a mobile communication system using a spread spectrum signal, satellite communication, and a navigation system such as GPS (GNSS). Can be.

Accordingly, the present invention will be described with reference to the embodiments shown in the drawings, which are merely exemplary and will be understood by those skilled in the art that various modifications and variations of the embodiments are possible therefrom. And, such modifications should be considered to be within the technical protection scope of the present invention. Therefore, the true technical protection scope of the present invention will be defined by the technical spirit of the appended claims.

FIG. 3 is a first embodiment of the present invention, based on a compressed sensing technique in which a general GNSS receiver searches for a received GNSS signal and acquires code phase information based on the present invention. An example of the implementation of a signal searcher is shown.

As shown in FIG. 3, the GNSS and the spread spectrum signal acquisition apparatus according to the first embodiment include a signal search unit 300. In this case, the signal search unit 300 includes parallel correlators ( 310, a signal generator 320, and a decoding (signal information search) module 330.

First, the signal received through the antenna 301 of the receiver provided in the GNSS and the spread spectrum signal acquisition device is a chip speed through the frequency down converter (302) and ADC (Analogue to Digital Converter) (303) A sampled signal obtained by sampling at a sampling frequency twice as high as a chip rate is input to the signal search unit 300. In general, since the code phase of the signal received at the receiver is searched in units of 0.5 chips, it is assumed in the present invention that the output of the ADC 303 is sampled at 2 chips per chip. The signal input to the signal search unit 300 is first passed through a parallel correlator 310 composed of M correlators and then output to a decoding module 330 which is a digital signal processor as a correlation output of each correlator. At this time, the parallel correlator 310 is correlator 1 (311), correlator 2 (312),... Correlator M (313), the correlator input of the parallel correlator 310 according to the present invention is different from the correlation scheme for the search for a conventional conventional spread signal.

Hereinafter, a specific process of obtaining a measurement value of a GPS or spread spectrum signal by the compression detection technique according to the present invention will be described.

In general, a correlator for detecting a spread spectrum signal performs a correlation between an internally generated signal and a received signal, and the internally generated signal has the same spreading code as a spreading code of the received signal. A signal made up of a code. A receiver having a parallel correlator generates a plurality of spreading code signals having different time delays (code phases) from the internally generated signals and performs correlation between each signal and the received signal.

However, the correlation scheme according to the invention performs a correlation between the received signal and a special signal produced internally. The particular signal is each row of the A matrix and the A matrix has a magnitude of [M × L]. The internally generated special signal is a signal generated through the signal generator 320 and is made through the following process. In this case, the signal generator 320 may include a PN generator 322, Ψ generator 323, Φ generator 324, A generator 325.

First, the information in block 321 is passed to the signal generator 320 as including basic information (matrix size of A matrix and spreading code information) for making the matrix A, starting with a value of F = 1. Based on the information in block 321, PN generator 322 generates a spreading code signal of the signal to be acquired. Generation of spreading codes is generally implemented using a linear feedback shift register (LFSR). The matrix Ψ produced by the Ψ generator 323 via the PN generator 322 is a correlation matrix of a spreading code signal having the Doppler frequency f _F (F is the Doppler frequency index). It can be represented as.

[Equation 1]

Ψ = Ψ ₁ · Ψ _2,

here

In Equation 1, p [k] is the value of the kth chip (code) of the spreading code of the signal to be searched (k = 1, 2, 3, ... N), and a value of +1 or -1. Has Where N (>> 1) is the number of codes corresponding to one period of the spreading code. (For reference, the C / A code of the GPS L1 signal is known as N = 1023, N = 32768 in IS-95 (CDMA) mobile communication, and 38400 in 3G mobile communication WCDMA.) The reason why p [k] is repeated two times in order to synchronize the receiver to generate two samples per chip from the received signal. That is, the time length of one chip of the spreading code

(R _C is the chip rate), each row of Ψ ₁ represents a spreading code having a different code (phase) delay in units of 0.5T _C. Ψ ₁ has a size [L × L], hereinafter L = 2N. Also, f _F represents the Doppler frequency of the received signal to be searched, and F is an integer index of the Doppler frequency to be searched. When f ₀ is the smallest Doppler frequency value, F f is a predetermined small frequency interval δ f.

Represented by The reason for increasing the F to search for various Doppler frequencies is because it is impossible to accurately predict the Doppler frequencies of satellite signals received from terrestrial receivers from the beginning. In general, the Doppler frequency range searched by terrestrial GPS receivers is mainly in the 10KHz band from -5KHz to + 5KHz and assumes a correlation length of 1ms, so that δf becomes 500Hz, so if f ₀ is -5KHz, F is 0. It has an integer value up to 21. However, since the value of δf varies depending on the correlation length of the receiver (inversely), F = 1, 2, 3,... Expressed by, F _max .

Parallel correlators in conventional receivers receive received signal samples of the same length as each row of Ψ ₁ and perform correlation between the two inputs. However, in the implementation of the signal search unit 300 of the GNSS or spread spectrum signal using the compression detection technique according to an embodiment of the present invention (in Fig. 3), the internally generated signal input to each parallel correlator 310 is an A generator ( Generated through 325, the A generator 325 is made of the matrix product of the correlation matrix Ψ produced above and the measurement matrix Φ generated in the Φ generator 324 of the following equation (2).

[Equation 2]

,

Where m _1st row of matrix Φ ₁ and

₂ m-th row of the elements and the value Φ ₂ in the second column and

The element value located in the first column is expressed as follows.

,

, And

Where M ₀₁ and M ₀₂ are positive integers,

,

,

,

,

Is a real number greater than zero (usually a value between 3 and 7) and WH _M1 is a Walsh-Hadamard matrix with a size of [M ₁ × M ₁ ]

Is the ceiling function. Therefore, Φ ₁ is a matrix in which each column of the Walsh matrix having the size [M ₁ × M ₁ ] is repeated β times in succession. For example, when β = 8 and L = 2046, since M ₁ = 256, Φ ₁ is obtained as follows. (Generally use β≥4.)

Here, wi is an arbitrary Walsh code index ( ₁ ≦ Wi ≦ M ₁ ) and the Walsh code column W _wi having wi as the Walsh code index has a size of [M ₁ × 1]. Also, the Walsh code in the last column

Was repeated a number less than β to match the number of columns of Φ ₁ . However, most Walsh code strings forming Φ ₁ are repeated β times.

As a result, the measurement matrix Φ becomes a matrix of size [(M ₁ + M ₂ ) × L] (wherein (M ₁ + M ₂ ) << L). Therefore, the finally generated compression sensing matrix A (= ΦΨ) has a matrix size of [(M ₁ + M ₂ ) × L]. The received signal of size [L × 1] obtained by sampling (sampling) twice per chip of the received signal r (t) is X = [r (1), r (2), r (3),... , r (L)] ^T (where

Is a transpose, each correlator 311 to 313 constituting M (= M ₁ + M ₂ ) parallel correlators 310 correlates between all rows of the compression sensing matrix A and the received signal. Is performed to obtain the measured value Y. That is, the measurement vector is Y = AX, and Y has a size of [(M ₁ + M ₂ ) × 1].

When the correlation output of Ψ with respect to the sampled received signal X is expressed as X ₁ , the output Y of the (M ₁ + M ₂ ) parallel correlators is represented by Equation 3 below.

[Equation 3]

Unlike FIG. 3, in the second embodiment of the present invention illustrated in FIG. 4, only M ₁ rows of A are used _first without using all (M ₁ + M ₂ ) rows of A according to an implementation algorithm. It is also possible to estimate the Doppler frequency and approximate code phase and perform signal search faster by using the remaining M ₂ rows of A later. That is, M ₁ rows made of Φ ₁ Ψ are used to obtain measurements Y ₁ (F) for different Doppler frequencies (= Y ₁ measurements at Doppler frequency f _F ) and F _{max values} obtained for all Doppler frequencies. After estimating the Doppler frequency closest to the received signal and the frequency from Y ₁ , we obtain Y ₂ using M ₂ rows obtained from Φ ₂ Ψ only for the estimated Doppler frequency to estimate the more accurate code phase. This results in M ₁ measurements Y ₁ for all Doppler frequencies and approximate the final estimated Doppler frequency, unlike the method of FIG. 3 (where (M ₁ + M ₂ ) measurements Y are obtained for all Doppler frequencies). M ₂ measurements Y ₂ are obtained only once for the code phase delay value. Therefore, in the former case, the correlation function of the total (M ₁ + M ₂ ) F _max times must be driven, while in the latter case, the correlation function of the total M ₁ F _max + M ₂ times can be driven.

Conventional receivers using conventional parallel correlators should use L (= 2N) parallel correlators for all Doppler frequencies (F _max total) to obtain a total of LF _max correlation measurements. in the signal search section 300 using a compression technique in accordance with detected that it can also perform the same function as a measure for obtaining a minimum M ₁ + M ₂ F _max parallel compression Any deception (wherein, M _1, M ₂ << L). In the present invention, since M ₁ and M ₂ is a small value of 10-25% of L, the receiver according to the present invention receives GNSS (GPS) received using about 4-10 times fewer parallel correlators than conventional receivers. Alternatively, a spread spectrum communication signal may be detected and acquired.

For example, suppose there is no frequency error due to the Doppler frequency in the received GPS signal (or if the receiver selects a frequency that matches the Doppler frequency of the received signal), then X ₁ (= ΨX) is the L parallels used by the existing receiver. As seen from the correlator output, most elements of the vector X ₁ (size [L × 1]) have a small absolute value, with only three consecutive elements of L having a particularly large absolute value (among others). The middle value has the largest triangle value).

For example, X ₁ =

[-0.1, 0.2, 0, -0.1, ... 0.1, 5, 10, 5, -0.1, -0.1, 0.2, ... 0.1]. here,

Is the real number related to the strength of the received signal, and θ is the frequency phase of the signal.

And

to be. (If a receiver has one sampling frequency per chip, the size of L is N, not 2N, and only _one element of X ₁ has a particularly large absolute value, not three.) value representing a value to v, and the subsequent value of the absolute value is larger v = [V1, V2, V3 ] (| V2 |> | V1 |, | V2 |> | V3 |) is expressed as X ₁ is v and It is a vector with V mixed together. In this case, the receiver is an index (index) where the element V2 is located in the X ₁ vector

The code phase search is completed by finding. (In the description below, the index value of V2

Is displayed. In addition, if the Doppler frequency predicted by the receiver does not match the Doppler frequency of the received signal, the signal correlation becomes low, so the values of all elements of X ₁ can be expressed as v.) Parallel correlation without using a compression detection technique. In general, a receiver with a group uses only the correlation matrix Ψ and simply finds the element with the largest absolute value in the L correlator outputs X ₁ and indexes the element.

Find it.

Hereinafter, information necessary for signal acquisition from a measurement value of a GNSS or a spread spectrum signal obtained by a compression sensing technique according to the present invention (for GPS, a received signal in case of a mobile communication system having a code phase value, a Doppler frequency value of a received signal, and a spread spectrum method) The algorithm of the decoding module 330 that obtains the code phase value of &quot;

First, in the first embodiment of FIG. 3, each row of the matrix A (total M rows) with respect to the Doppler frequency (expressed by the index F) to be searched is assigned to each correlator 311, 312, and 313 of the parallel correlator 310. By correlating with the sampled signal that is input and received, M correlation outputs Y (1 ~ M | F) are obtained, then the Doppler frequency and the approximate code phase value are obtained from Y ₁ (⊂Y) and Y ₂ ( ⊂Y) is the correct code phase. Firstly, the Fast Walsh Hadamard Transform (FWHT) function 331 is the correlation output Y of M parallel correlators 310 (measured with A obtained using Ψ using the Doppler frequency f _F value to search). 1]) to the receiving input to perform a M ₁ gae FWHT (M ₁ point-FWHT) to that of the Y _1, M ₁ results obtain a value as a result takes the absolute value to a value M ₁ of Z ₁ (F, wi Get) (Wherein Z ₁ (F, wi) represents the index to the index 1≤wi≤M ₁ F and M ₁ of FWHT output indicating a Doppler frequency of searching.) The outputs M ₁ Z ₁ (F, wi) (1≤wi≤M ₁₎ is that the first decision function (332) a first threshold (TH ₁₎ is compared with the first threshold value becomes equal to or greater than the output (TH ₁₎ Z ₁ (F, wi) preset in To judge. If the first difference is determined to go to board formal (Z ₁ (F, wi)> TH ₁₎ Z ₁ (F, wi), the Doppler frequency control function 335. If not satisfying the function 332, the Doppler frequency index Is increased by 1 to obtain parallel correlator 310 output Y (1 ~ M | F) for another Doppler frequency signal. If there exists Z ₁ (F, wi) that satisfies the _first decision formula (Z ₁ (F, wi)> TH ₁ ) of the first decision function 332, the FWHT output index wi is designated as Wi. do. The detailed detection function 333 has three inputs: (1) Y ₂ from the FWHT function 331, (2) Wi from the first decision function 332, and (3) from the signal generator 320.

Get input. The detailed detection function 333 first calculates a set Iw having all (maximum β) column indices of Φ ₁ having Wi as the Walsh code index. Here, the set Iw may be expressed as in Equation 4.

[Equation 4]

At this time, the set Iw has a maximum maximum β values between [1, L].

And, the detailed detection function 333 is Y ₂ and

(We perform correlation between k and take the absolute value of the correlation results to get Z ₂ (F, k). (In some cases, we found Nw (> 1) Wis under the influence of noise.) In this case, Iw is a set having all (maximum Nw × β) column indices of Φ ₁ having all Wi as Walsh code indices. The output is passed to the second decision function 334 to find k that satisfies the _second decision formula (Z ₂ (F, k)> TH ₂ ) with respect to the preset second threshold value TH ₂ . If Z ₂ (F, k) is found to satisfy the second decision, the second decision function 334 uses the index k to estimate the code phase of the currently received signal.

Doppler frequency estimate of the actual received signal

Will print If Z ₂ (F, k) that satisfies the second decision equation is not found in the second decision function 334, the Doppler frequency control function 335 is performed to change the Doppler frequency to be searched to change the new Doppler frequency. Repeat the above signal search process.

The first embodiment of the present invention with respect to FIG. 3 obtains a correlation output Y of magnitude [M × 1] every time for all Doppler frequencies, but the determination is performed using only Y _{1 in} the first determination function 332. Y ₂ is input to the detailed detection function 333 only when the first determination is satisfied. Therefore, in most cases Y ₂ is not used and the Doppler frequency

Is used only if is nearly equal to the actual Doppler frequency received.

In order to reduce such unnecessary measurement, Figure 4 shows a signal search unit 400 according to a second embodiment of the present invention. Functions 401, 402, 403, 422 and 423 of FIG. 4 perform the same functions as the functions 301, 302, 303, 322 and 323 of FIG.

First, generation of the first compression detection matrix A ₁ starts by receiving the following information. Set Stage to 1, and set Ψ and Φ ₁ to [L × L] and [M ₁ × L], respectively, and specify the Doppler frequency index value to determine the Doppler frequency to be searched. Start with 1). In addition, in order to specify the PN code information to be generated, the PN code information (PRN number) of the GNSS is received in the case of CDMA or WCDMA mobile communication, and receives the PN code information of the base station to be found. The Φ ₁ generator 424 generates a measurement matrix Φ _{1 in} which each column of the Walsh matrix WH _{M1 having} the size of [M ₁ × M ₁ ] is repeated β times in succession. Make. Accordingly, the output of the A ₁ generator 425 may be expressed as Equation 5 as the product of the correlation matrix Ψ and Φ ₁ output from the Ψ generator 423 as the _first compression detection matrix A ₁ (where Ψ). Is a correlation matrix, which is made according to the expression in Equation 1).

[Equation 5]

A ₁ = Φ ₁ Ψ

Each row of the first compression detection matrix has a size of [1 × L] as shown in FIG. 3 and is input to each correlator 411, 412, 413 of the first parallel correlator 410, respectively. It is correlated with the sampled received signal (sample signal magnitude = [1 × L]). The output Y ₁ of each correlator 411, 412, 413 is the output of the first parallel correlator 410 for any Doppler frequency (index F) to be searched, and is expressed as Y ₁ (1 to M ₁ | F). Has a size of [M ₁ × 1]. FWHT function 431 is equal to _{Y 1 (1 ~ M 1 |} F) in the first embodiment of Figure 3 receives the M ₁ perform gae -FWHT (M ₁ -point Fast Walsh Hadamard Transform), and the M Z ₁ (F, wi) is obtained by taking the absolute value of M ₁ output of ₁ -FWHT. (Wherein Z ₁ (F, wi) represents an index wi (1≤wi≤M ₁₎ for index F and M ₁ of FWHT output indicating a Doppler frequency of searching.) The results M ₁ Z ₁ (F, wi) (1≤wi≤M ₁₎ is the first decision function (432) is input to a preset first threshold (TH ₁₎ and the first threshold (TH ₁₎ or more Z ₁ (F compared, wi Is determined. If the first proceeds to the first chapan formal (Z ₁ (F, wi)> TH ₁₎ Z ₁ (F, wi), the Doppler frequency control function 435. If not satisfying the difference determination function 432 doppler The frequency index is increased by 1 to repeat the signal measurement and determination (perform functions 420, 410, 431, 432) for the other Doppler frequency signals. If there exists Z ₁ (F, wi) that satisfies the first decision formula (Z ₁ (F, wi)> TH ₁ ) of the first decision function 432, the index wi is designated as Wi and the current Searched Doppler Frequency Estimates (

) And Wi to block 451. In block 451, information for estimating a detailed code phase value and some basic information for generating a second compression detection matrix A ₃ are passed to block 450. (In some cases, due to the influence of noise, Nw (> 1) Wis may be found, where Iw is an index of all (up to Nw × β) columns of Φ ₁ having all Wis as Walsh code indices.) Information transmitted from the block 451 to the block 450 is the current searched Doppler frequency estimate (the result of the first decision function 432).

Information about the size of column indices (Iw), A ₃ of all (maximum β) columns having Wi as the Walsh code index in the measurement matrix Φ ₁ -M ₃ and L, and PN PRN information of the GNSS for code generation (PN code generation information to be searched by the terminal in case of CDMA and WCDMA mobile communication). The Iw is made as shown in Equation 4. Also, in block 451 the variable 'Stage' is assigned a value of 2 (Stage = 2). Based on the information input from the block 451, the A ₃ generation block 450 first generates a PN code from the PN generator 452 (function 422 is the same as function 322). The generated PN code and the estimated Doppler frequency estimate (

) Is input to the generator 453 so that the Doppler frequency

Create a correlation matrix Ψ at. When the total number of elements of Iw input from the block 451 is N _I , Iw is input to the Φ ₃ generator 454 and the measurement matrix Φ ₃ has a size of [M ₃ × L] (where M ₃ As x = 2 ^X , x is a smallest positive integer such that 2 ^X ≥ N _I. The measurement matrix Φ ₃ consists of zero vectors ( 0 ) of all columns except the column whose element value of Iw is an index, and the rest of the columns are different Walsh codes of length M ₃ . Have with heat. Therefore, the Φ _{3 3} m-th row and

Th column (

) Is expressed as

[Equation 6]

Here, WH _M3 is a Walsh-Hadamard matrix having a size of [M ₃ × M ₃ ].

E.g,

When Wi is 2, Iw = {9, 10, 11, 12, 13, 14, 15, 16}, and Φ ₃ is as follows.

here,

Represents the nth column of WH _M3 .

Finally, the A ₃ generator 455 makes the second compression detection matrix A _{3 the product} of the measurement matrix Φ ₃ and the correlation matrix Ψ (A ₃ = Φ ₃ Ψ). All rows of the second compression detection matrix A ₃ are input to the second parallel correlator 440 to perform correlation between each row of A ₃ and the received sample signal and as a result of the parallel correlation. Doppler frequency currently searching

And phase of received signal

The compression detection measurement Y ₃ , assuming an approximate estimate of in the Iw interval, is obtained. (Wherein Y ₃ has a size [M ₃ × 1] and may be expressed as Y ₃ (1 to M ₃ | F, Iw). Also, the second parallel correlator 440 may be the first parallel correlator 440). 410). The output Y ₃ is input again to the FWHT function 431, and the FWHT function 431 performs M ₃ -FWHT (M ₂ -point Fast Walsh Hadamard Transform) and the FWHT function. M ₃ Z ₃ (F, k) is obtained by taking the absolute value of the M ₃ output results of (431). (Wherein Z ₃ (F, k) represents the index of the index F 1≤k≤M ₃ and M ₃ of FWHT output indicating a Doppler frequency of searching.) The results M ₃ Z ₃ (F, k) is the a second threshold previously set in the second function determination (434) (TH ₃₎ that is the second threshold value (TH ₃₎ above are compared to the results to determine if the Z ₃ (F, k). Ten thousand and one second control proceeds to the second chapan formal _{(Z 3 (F, k)} > TH 3) Z 3 when there is no (F, k), the Doppler frequency adjustment function (435) that satisfies the difference determination function 434 doppler The frequency index is increased by 1 to repeat the signal measurement and determination on another Doppler frequency signal (perform functions 420, 410, 431, 432). If there exists Z ₃ (F, k) that satisfies the second determination formula (Z ₃ (F, k)> TH ₃ ) of the second determination function 434, the FWHT output index k is currently received. Code phase estimate of the signal

And the currently searched Doppler frequency estimate (

)and

Is output as a final signal acquisition result.

In the case of signals such as Galileo in Europe, in GPS, X ₁ shows a triangular shape where the code phase of the signal fits, but since the BOC (Binary Offset Carrier) method is used, the correlation result is not + It takes the form of a sinc function where the value and the -value intersect. 5 shows a correlation result between a general GPS and a Galileo signal. In FIG. 5, the horizontal axis is a relative code phase difference between a spreading code of a received signal and a spreading code generated internally by a receiver, and a value thereof is expressed in units of chips. Although the signal acquisition technique using the compression detection technique of GPS and spread spectrum signal proposed in the present invention can be used as it is in the general GNSS signal, but if you want to acquire the signal of the Galileo navigation satellite using the most common BOC (1,1) All odd rows of all measurement matrices Φ (ie, Φ ₁ , Φ _2, and Φ _3, etc. as shown in FIGS. 3 and 4) with 2N (= L) rows made in units of 0.5 chips (T _C ) multiply -1 by columns or all even columns. For example, when β = 8 and L = 2046 as an implementation example of FIG. 3, M ₁ = 256, and Φ ₁ for Galileo signal acquisition is obtained as follows.

or

In the embodiment of Φ ₃ illustrated in the description of FIG. 4, the present invention is modified as follows.

or

Becomes That is, it is generated by multiplying -1 by an even or odd column of a measurement matrix having L (= 2N) columns in units of 0.5 chips (T _C ). Thus, for Galileo signals using BOC (1,1), all odd columns or all even columns of all measurement matrices Φ (Φ ₁ , Φ ₂ and Φ _3, etc.) in the implementations of FIGS. even columns).

As such, according to two embodiments of the present invention, a separate communication device is not required as in the conventional A-GPS, and signals can be searched and acquired several times faster than the receiver technology of the conventional GNSS and spread spectrum communication systems. Since the hardware of the receiver for searching and signal acquisition is greatly consumed, the optimal receiver can be implemented with minimum time and hardware cost.

Methods according to an embodiment of the present invention may be implemented in the form of program instructions that can be executed by various computer means and recorded in a computer readable medium. The computer readable medium may include program instructions, data files, data structures, etc. alone or in combination. Program instructions recorded on the media may be those specially designed and constructed for the purposes of the present invention, or they may be of the kind well-known and available to those having skill in the computer software arts. In addition, the above-described file system can be recorded in a computer-readable recording medium.

Another embodiment of a compression sensing technique can be described as follows.

6 illustrates a signal search unit of a conventional GNSS receiver using a correlator. In addition, FIG. 6 illustrates a serial search method as a basic example of a method of using a correlator. Since the overall structure of the two-dimensional compression correlation technique proposed in the present invention is similar to the general signal searcher illustrated in FIG. 6, the description of FIG.

First, the GNSS signal received from the antenna is converted into an intermediate frequency (IF) signal through a radio frequency (RF) processing block. In FIG. 6, the received signal which has been converted to IF is described as an input of the signal search unit. The signal received is amplitude A, phase

Is the code signal c (t) of satellite s with the center frequency f ₁ and the code phase delay t ₀ . As shown, AWGN (Additive White Guassian Noise, white) represented by n _I (t) and n _Q (t) are respectively included in the received signals r _I (t) and r _Q (t) input to the I and Q channels. Noise) is included. The received signals r _I (t) and r _Q (t) are specific frequencies generated by the receiver in block 204 in the first multipliers 201 and 202.

Multiplied by the Carrier signal and the 90-degree phase-shifted Carrier signal, respectively

Is a value known to the receiver). The output of the first multiplier generates I _IF [nTs] and Q _IF [nTs] through band pass filters (BPF: 211 and 212) and samplers (ADC: 621 and 622), respectively. Where Ts is the interval between samples and n is the time index at each sampling point. The I _IF [nTs] and Q _IF [nTs] are again code signals c of a specific satellite s having a specific code phase delay (or code phase code phase) k _t generated by the receiver in the quadratic multipliers 631 and 632. multiplied by (t) The outputs of the quadratic multipliers 631 and 632 are represented by I [nTs] and Q [nTs], respectively, and N temporal consecutive outputs are accumulated in the accumulators 641 and 642, respectively, to generate Z _I and Z _Q , respectively. Occurs. The Z _I and Z _Q are squared in the squarers 651 and 652, respectively, and the values are added in the summer 661 to determine a decision variable.

Make The determination variable Z is a detection threshold in the primary discriminator 662.

Is compared to

If larger, it is determined that the GNSS signal is found, and the index k _f and the code phase delay value k _t representing the Doppler frequency value at this time are output. The detection thresholds are Coherent Correaltion Length and Non-Coherent Accumulation Length, Probability of Detection, Probability of False Alarm, and Doppler Frequency Hypothesis Interval. Doppler frequency search step). If Z is

If it is smaller, the current hypothesis (the Doppler frequency hypothesis represented by k _f and the code phase delay hypothesis represented by k _t ) is judged to be wrong, and the signal search is performed in the new hypothesis by increasing k _f or k _t by one. . This signal search is typically

For the Doppler frequency hypothesis interval and the code phase delay hypothesis interval 1≤k _t ≤K _T. (For GPS receivers using GPS L1 C / A signals, they usually have a correlation length of 1 msec. Since sampling is performed twice per chip and the C / A signal period is 1023, K _F = 5KHz, Δf = 500Hz, K _T = 2046).

In general, the GNSS receiver is implemented by verifying the code phase delay hypothesis of all signals within the set Doppler frequency hypothesis. Therefore, the Z value in the primary discriminator 662

If it is determined to be smaller, the secondary discriminator 671 determines whether the code phase delay hypothesis currently tested is the last hypothesis. That is, k _t = K _T if the case whether or not to determine the, otherwise (k _t ≠ K _T), the following code-phase (delay) of one or more increases the k _t value at block 634 to see test the hypothesis is increased k _t value In block 633, a PRN code signal of a GNSS satellite having a current code phase delay value is generated. (Note that the initial Doppler frequency and code phase delay hypotheses are k _f = 1 and k _t = 1, as specified in block 605.) Thus, the signal generated at the current time t = nTs is C, as shown. ^s [nk _t ]. If it is determined in the secondary discriminator that k _t = K _T , it initializes to k _t = 0 in block 272 and proceeds to block 634. At the same time, if it is determined that k _t = K _T in the secondary discriminator, it proceeds to the tertiary discriminator 681 to determine whether k _f = K _F , and if k _f ≠ K _F , block 635 Proceed to increase the value of k _f by one more to start searching for signals in the new Doppler frequency hypothesis. If k _f = K _F , initialize block _f 682 to k _f = 0 and proceed to block 635. Block 603 is a phase shifter that causes the internally generated Carrier signal generated at block 604 to be phase shifted by 90 degrees.

As described in FIG. 6, the conventional GNSS signal search unit performs signal correlation for each individual hypothesis representing one Doppler frequency and one code phase delay, and thus, 21 Doppler frequencies and 2046 codes. If phase delay is possible, all 43,000 hypotheses must be tested individually to verify the detection variables in each hypothesis.

7 illustrates a 1 ^st stage signal searcher according to a specific embodiment of the present invention. Unlike the conventional GNSS receiver, which detects only one Doppler frequency and code phase delay hypothesis pair each time, as shown in FIG. 7, the primary signal searcher correlates multiple Doppler frequency hypotheses and multiple code phase delay hypotheses once. A signal searcher using a two-dimensional compression correlator that can be searched by In the present invention, the Doppler frequency and the code phase found according to the implementation method of FIG. 7 provide a method of additionally performing the secondary signal search unit of FIG. 8 from an approximate value. In the present invention, the first signal search unit first finds the Doppler frequency and the code phase delay of the received signal within the reduced hypothesis range by using these two sequential (first and second) signal searching methods, and then, more specifically and in detail. By detecting and finding the correct Doppler frequency and code phase delay, the overall GNSS signal acquisition time is greatly reduced. In the following description, while describing the implementation of FIG. 3, the description of the same block will be simplified as much as possible because a number of blocks have the same function as that of FIG. 6.

First, the signals r _I (t) and r _Q (t) received through the I and Q channels are the same as those of the input signal of FIG. 6, and the functions of the first multipliers 701 and 702 and the phase shifter 703 are respectively shown in FIG. The first multipliers 601 and 602 and the phase shifter 603 shown in FIG. The receiver internally generated carrier signal, which is multiplied with the signal received by being input to the first multipliers 701 and 702, is a combined carrier signal unlike in FIG. 2. Blocks 711 and 712 in FIG. 7 represent the band pass filters BPF 611 and 612 and the samplers ADC 621 and 622 introduced in FIG. Thus, the outputs of blocks 711 and 712 are I _IF [nTs] and Q _IF [nTs], and I _IF [nTs] and Q _IF [nTs] are again complex codes generated by the receiver in quadratic multipliers (731, 732). It is multiplied by the combined code phase signal c (t) of a particular satellite s with a combined code phase delay. The outputs of the quadratic multipliers 731 and 732 are represented by I [nTs] and Q [nTs] (or I and Q), respectively. N accumulators 741 and 742 output N temporally consecutive quadratic multipliers. Accumulate to generate Z _I and Z _Q respectively. The and are squared in the squarers 751 and 752, respectively, and summed in the first-order summer 761 to determine the decision variable.

Make

The determination variable Z is the 1 ^st detection threshold in the primary discriminator 762.

Is compared to

If larger, it is determined that the GNSS signal is found, and the index k _{f, 1} representing the detected Doppler frequency value at this time and the value k _t representing the approximate code phase delay are output (this output is shown in FIG. 8). It is input to the secondary signal search unit and decomposed into detailed hypotheses, and becomes a reference for more detailed signal search.) The first detection threshold (Detection Threshold)

An appropriate value is determined by various variables such as the detection threshold of FIG. 6. If the Z value is

If it is smaller, it is judged that the current hypothesis is wrong and the signal search in the new complex hypothesis is performed by increasing k _f or k _t one step further. The Z value in the primary discriminator 762

If it is determined to be smaller, it is determined whether the composite code phase delay hypothesis currently tested in the secondary discriminator 771 includes the last code phase hypothesis. In other words,

If not, the PRN signal generator 773 generates a PRN code signal of the GNSS satellite which increases the k _t value by K _t further and blocks the increased k _t value as a code phase delay value at block 772. Let's do it. The output of the PRN signal generator 773 and its output are again [1, 2,... , Pt signals delayed in time by K _t -1] Ts are output to the second summer 775. In the case of a signal having Binary Phase Shift Keying (BPSK), such as GPS, the PRN signals are all added with the same sign so that the output cc (t) of the quadratic summer 775 is c (nk _t ) + c (nk _t -1) + c (nk _t- 2) + .. + c (nk _t -K _t + 1). Alternatively, signals such as Galileo's BOC (n, n) are added with the sign changed for each time delay value such that the output cc (t) of the quadratic summer 775 is c (nk _t ) -c (nk _t). -1) + c (nk _t -2)-.. + (-1) ^Kt-1 c (nk _t -K _t +1). Accordingly, the output of the quadratic summer 775 is a composite signal in which a plurality of PRN signals having different successive code phase delays are summed. If, in the secondary discriminator 771

If it is determined to be, in block 774 k _t = 1 is initialized and proceeds to the PRN signal generator 773. In addition, at the same time proceed to the third discriminator (781)

Determine whether or not. if

If, then proceed to blocks 7821, 7822, and 7823, the frequency index of the composite signal.

Increase the values of K by all K _f and proceed to the fourth order discriminator (7841, 7842, 7843), respectively.

Determine if the values are less than K _F. If there is a frequency index with a value greater than K _F , no Carrier signal with that Doppler frequency is generated. Therefore, as shown in the figure, when the frequency index is smaller than K _F , the process proceeds to block 7181 or block 7272 or block 7535 and the frequency index inputted from the Carrier Generator (

Etc.) to generate Carrier signals. Carrier signals generated in the block 7785 or block 7702 or block 7755 are inputted to the third-order summer 704 without increasing or decreasing the output carrier signal of the block 3851 as shown in the block 7786 and the block 7703. The signal is phase-decreased by 90 degrees in the primary phase converter 7802 and input to the tertiary summer 704. The carrier signal is then phase-converted in a 180-degree phase-reduced manner and input to the tertiary summer 704. . Therefore, the output signal of the Carrier Block 7853 is the phase difference in K _f-phase converter (7863) is reduced by 90 * _(f K -1) is input to the third summer 704. The As a result, the output of the cubic summer is mathematically

It can be expressed as. For reference, the initial code phase delay and Doppler frequency values of the primary signal search unit shown in FIG. 7 follow the values of blocks 774 and blocks 7831 to 7333. For reference, since the primary signal searcher of FIG. 7 may have values of K _t ≥ 3 and K _f ≥ 3, at least 9 times faster searching is possible than the signal searcher of FIG. 6.

The reason for using the phase shifters 7786 and 7863 for the outputs of the carrier generators 7785 to 7535 is that the carrier signals generated by the carrier generators 7785 to 7855 are accumulated in the accumulators 741 and 742. This is to compensate for the average phase change that occurs. In the present invention, since the correlation length of Δf = 500 Hz and 1 msec is taken as an example, neighboring Carrier signals generated by the Carrier generators 7785 to 7855 having a relative Doppler frequency difference of Δf are included in the accumulator 741. A phase increase of 90 ^° occurs in the output signal accumulated at 742. The phase shifters 7802 and 7863 are used to compensate for this and to maximize the size of Z _I and Z _Q. Therefore, if the product of the relative Doppler frequency deviation Δf of the neighboring Carrier signals generated by the Carrier generators 7785 to 7855 and the temporal cumulative length of the accumulators 741 and 742 is different from the example of the present invention (500 Hz and 1 msec, respectively) Unless otherwise 0.5, the phase increase and decrease in the phase shifters 7786 and 7863 must also be appropriately varied.

8, there is shown a second signal search section ^{((2 nd Stage Signal Searcher)} ) presented in this invention. The basic structure diagram of the secondary signal searcher is very similar to the implementation structure of the signal searcher of the general GNSS receiver described with reference to FIG. 8. For example, blocks 801, 802, 803, 804, 811, 812, 821, 822, 831, 832, 833, 841, 842, 851, 852, 861, and 862 of FIG. 8 are blocks 601, 602, and 603 of FIG. 6, respectively. , 604, 611, 612, 621, 622, 631, 632, 633, 641, 642, 651, 652, 661, 662, respectively, the description thereof is omitted.

The secondary signal search unit shown in FIG. 8 receives the outputs k _{f, 1} and k _{t of the} primary signal search unit of FIG.

And

Replace with

and

Performs signal search in the Doppler frequency and code phase delay hypothesis region indicated by. At this time, the input

and

The hypothesis region represented by is expressed by the frequency index k _f and the code phase delay index k _t .

And

As shown in block 805, the initial Doppler frequency and code phase delay hypothesis index are respectively

Wow

It is designated as. Block 804 is a given initial Doppler frequency hypothesis (

Carrier signal is generated and supplied to the first multiplier 801, the same Carrier signal is passed through the phase converter 803 90 degrees phase increase is supplied to the first multiplier 802. Block 833 is a given initial code phase delay hypothesis (

A PRN code signal c ^s [nk _t ] having a code phase delay according to C 1) is generated and supplied to the second multipliers 831 and 832. The determination variable Z thus obtained is the same value as the detection threshold shown in FIG.

This is because the secondary signal search unit (Fig. 8) has the same correlation scheme as the signal search unit shown in FIG. That is, since the secondary signal search unit has the same implementation as the signal search unit of the general GNSS receiver, the search performance is the same, and thus the same detection threshold may be used. (The difference between the secondary signal searcher of FIG. 8 and the signal searcher of FIG. 2 is that only the secondary signal searcher of FIG. 8 searches for a signal in a given narrow two-dimensional hypothesis region.) In the primary discriminator 862

If it is determined as, the currently verified Doppler frequency and code phase delay values (k _f and k _t ) are output as the final search result.

If it is determined to be, the process proceeds to block 871 to determine whether the current code phase delay k _t is the maximum value. therefore,

In case of, proceed to block 834 to increase k _t by 1 and generate a PRN code signal corresponding to the increased value in block 833. if

If is, proceed to block 872

Initialize k _t and proceed to tertiary discriminator 881 to determine if the current Doppler frequency k _f is the maximum of the given hypothesis range. therefore,

In case of, proceed to block 882 to increase k _f by 1 and generate a carrier signal corresponding to the increased k _f value in the oscillator 804. if,

In the case of, since the search in all hypothesis areas is completed, the search ends.

9 shows a brief operation plan of the primary signal search unit and the secondary signal search unit proposed in the present invention. First, in block 901, the primary signal search unit shown in FIG. 7 is performed to obtain approximate Doppler frequency and code phase delay information k _{f, 1} and k _t . Thereafter, in block 902, a hypothesis region (for example, to perform a detailed search from the inputted general information)

And

) Is transferred to block 903. In block 903, the second signal search unit shown in FIG. 8 performs the final Doppler frequency and code phase delay information (k _f and k _t ).

Compression hypothesis can be used for signal acquisition as well as signal tracking. 10 shows an example of a code signal tracking device using a compression correlator. This code phase tracker is the result of applying blocks 773 and 775 of FIG. 7 to the code phase tracker.

 After the output of the Early Compressed Code generation block 1001 is multiplied and integrated with the received signal (1005), the value determined by the Code Loop Discriminator 1004 is the pre-time code. The generator 1011 enters an input to generate a pre-time code signal (a code signal that is advanced by a specific phase than a currently estimated code estimate value). After the output of the Late Compressed Code Generation block 1002 is also multiplied and integrated with the received signal (1006), the value determined by the code loop discriminator 1004 is input to the rear-view code generator 1012. And generates a back-view code signal (a code signal delayed by a certain phase from the currently received code estimate value). The current code generation block 1003 generates a current code phase estimation value of the received signal, which is also multiplied by the received signal and integrated (1007) and output as a result value of the tracker.

In block 1001, n is an odd number of two

(1 ≦ i ≦ n) is a nonzero real or imaginary number and the delay element Di (1 ≦ i ≦ n) may be 1 chip. N is an odd number of 2

(1 ≦ i ≦ n) is a nonzero real or imaginary number and the delay element Di (1 ≦ i ≦ n) may be 1 chip.

 The pre-time compression code generation unit 1001, for example, linearly combines each of the signals delayed from the early code sequence generator 1011 by multiplying and adding 1031 each specific coefficient 1021. do. When PP is the output of 1003 blocks, PE is the output of 1001 blocks (output of 1031), and D = Di = 1chip, the equation is as follows.

Equation 7

The back-view compression code generation unit 1002 also includes, for example, each signal that is delayed from the back-code sequence generator 1012 is multiplied by each specific coefficient 722 and summed (1032). do. When PP is the output of 1003 blocks, PL is the output of 1002 blocks (output of 1032), and D = Di = 1chip, the equation is as follows.

Equation 8

When the output of the pre-time code generator 1011 is Pe and the output of the current-time code generator 1003 is PP from equations (7) and (8), the relation is as follows.

Equation 9

In addition, when the output of the rear-view code generator 1012 is Pl and the output of the current-time code generator 1003 is PP, the relational expression is as follows.

Equation 10

As a preferred example of the linear combination proposed in the present invention n is 3 in 1001 block

,

,

Di (1≤i≤n) is 1 chip. Also, in a 1002 block, n is 3

,

,

Di (1≤i≤n) is 1 chip.

When the code tracking device using the compression correlator is applied as shown in FIG. 11, the traceable linear section is wider than the conventional E-L correlator, so that it is easy to track in a high-traffic situation.

As described above, according to an exemplary embodiment of the present invention, since the signal search and acquisition can be performed several times faster than the receiver technology of the existing GNSS and spread spectrum communication system, the hardware of the receiver for signal search and signal acquisition is greatly consumed. The optimal receiver can be implemented with minimum time and hardware cost.

Methods according to an embodiment of the present invention may be implemented in the form of program instructions that can be executed by various computer means and recorded in a computer readable medium. The computer readable medium may include program instructions, data files, data structures, etc. alone or in combination. Program instructions recorded on the media may be those specially designed and constructed for the purposes of the present invention, or they may be of the kind well-known and available to those having skill in the computer software arts. In addition, the above-described file system can be recorded in a computer-readable recording medium.

As described above, the present invention has been described by way of limited embodiments and drawings, but the present invention is not limited to the above embodiments, and those skilled in the art to which the present invention pertains various modifications and variations from such descriptions. This is possible.

Therefore, the scope of the present invention should not be limited to the described embodiments, but should be determined not only by the claims below but also by the equivalents of the claims.

The present invention is applied to Global Navigation Satellite System (GNSS) and Spread Spectrum signal receivers.

Claims (22)

1. A signal search unit for searching for at least one received signal of a global navigation satellite system (GNSS) positioning signal or spread spectrum signal,

   The signal search unit,

   A signal generator for generating a compression sensing matrix having a matrix size of [M × L] using a correlation matrix having a matrix size of [L × L] and a measurement matrix having a matrix size of [M × L];

   A parallel correlator having M correlators configured in parallel to perform correlation between the received signal and the compression sensing matrix; And

   And a decoding module for obtaining a code phase and a Doppler frequency index of the received signal using the compression correlation result of the parallel correlator.
2. The method of claim 1,

   The signal generator,

   And generating the compression sensing matrix by matrix multiplication of the correlation matrix and the measurement matrix.
3. The method of claim 1,

   Each row of the correlation matrix is

   And a spreading code having a code delay of a predetermined time unit and the same as the code signal corresponding to the received signal.
4. The method of claim 1,

   The parallel correlator,

   And performing correlation between the M rows constituting the compression sensing matrix and the received signal to generate the compression correlation result having a matrix size of [M × 1].
5. The method of claim 7, wherein

   Each row of the compression correlation result is

   And a linear sum of each of the L columns forming the measurement matrix and the received signal as a coefficient.
6. The method of claim 7, wherein

   The decoding module,

   Computing an SNR that is a ratio of the signal found by the code phase based on the M compression coefficients and other signal components for all Doppler frequency indices, and then obtaining and obtaining the Doppler frequency index having the largest SNR. GNSS and spread spectrum signal acquisition device, characterized in that to obtain a code phase corresponding to one Doppler index.
7. A signal search unit for searching for at least one received signal of the GNSS positioning signal or spread spectrum signal,

   The signal search unit is a multi-stage signal search unit, and the GNSS and spread spectrum signal acquisition device that searches for decreasing the compression degree of the compression hypothesis as the stage increases
8. The method of claim 7, wherein

   A GNSS that searches for a compressed hypothesis that includes a plurality of individual frequency hypotheses and a plurality of individual code phase delay hypotheses in at least one of the stages of the signal searcher to obtain a reduced initial frequency range and an initial code phase delay range than the previous stage. And spread signal acquisition device
9. The method of claim 7,

   Compression hypothesis is a GNSS and spread signal acquisition device, characterized in that by generating a linear combination of a plurality of selected individual frequency hypothesis and a plurality of individual code phase delay hypothesis
10. The method of claim 7,

    Compression hypothesis is a GNSS and spread signal acquisition device, characterized in that by generating a linear combination of a plurality of consecutive individual frequency hypothesis and a plurality of individual code phase delay hypothesis
11. A signal search unit for searching for at least one received signal of the GNSS positioning signal or spread spectrum signal,

    The signal search unit GNSS and spread spectrum signal acquisition device using a compression hypothesis
12. The method of claim 11,

    GNSS and Spread Spectrum Acquisition Device for Approximate Signal Acquisition with Compressed Hypothesis
13. The method of claim 11

    Compression hypothesis is a GNSS and spread signal acquisition device, characterized in that by generating a linear combination of a plurality of selected individual frequency hypothesis and a plurality of individual code phase delay hypothesis
14. The method of claim 11,

    Compression hypothesis is a GNSS and spread signal acquisition device, characterized in that by generating a linear combination of a plurality of consecutive individual frequency hypothesis and a plurality of individual code phase delay hypothesis
15. A signal tracking unit for tracking at least one received signal of the GNSS positioning signal or spread spectrum signal,

    The signal tracking unit is a GNSS and spread spectrum tracking device using a pre-time compressed code correlator and a post-view compressed code correlator
16. The method of claim 15,

    Compression code is a GNSS and spread spectrum tracking device characterized in that to generate a linear combination of a plurality of consecutive individual code phase signals
17. A method of obtaining a code phase and a Doppler frequency index of a received signal by finding at least one received signal of a GNSS positioning signal or a spread spectrum signal,

    Generating a compression sensing matrix having a matrix size of [M × L] using a correlation matrix having a matrix size of [L × L] and a measurement matrix having a matrix size of [M × L];

    Performing correlation between the received signal and the compression sensing matrix through M correlators configured in parallel; And

    And obtaining a code phase and a Doppler frequency index of the received signal from the correlation result.
18. The method of claim 17,

    Generating the compression sensing matrix,

    And generating the compression sensing matrix by matrix multiplication of the correlation matrix and the measurement matrix.
19. The method of claim 17,

    Wherein each row of the correlation matrix is a spreading code having a code delay of a predetermined time unit and the same as a code signal corresponding to the received signal.
20. The method of claim 17,

    Performing correlation between the received signal and the compression detection matrix,

    And performing correlation between the M rows constituting the compression sensing matrix and the received signal to generate the correlation result having a matrix size of [M × 1].
21. The method of claim 20,

    Each row of the correlation result is

    And a linear sum of each of the L columns forming the measurement matrix and the received signal as a coefficient.
22. The method of claim 20,

    Acquiring a code phase and a Doppler frequency index of the received signal,

    Computing an SNR, which is the ratio of the signal found by the code phase based on the M correlation results and other signal components for all Doppler frequency indices, obtaining the Doppler frequency index with the largest SNR and obtaining the obtained And obtaining a code phase corresponding to the Doppler index.

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