KR20120076723A - Phase current controller for driving multi-phase bldc motor - Google Patents

Abstract

PURPOSE: A phase current control apparatus for driving a multiphase BLDC motor is provided to prevent malfunction due to noise and current ripple increase from an irregular switching frequency by driving with a fixed switching frequency. CONSTITUTION: A current detector(200) detects a current supplied to each phase of a seven-phase BLDC motor. A position detector(300) detects a rotation position of a rotor comprising the seven-phase BLDC motor. A speed controller(500) controls a speed of the rotor of the seven-phase BLDC motor. A pulse width modulator(700) outputs a pulse width control signal according to the number of pulses to a seven-phase inverter(800). The seven-phase inverter supplies voltage to each phase of the seven-phase BLDC motor.

Description

Phase current controller for driving multi-phase BLDC motor

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a brushless DC (BLDC) motor, and more particularly to a technology for driving a multiphase BLDC motor (Y connection).

BLDC (Brushless DC) motor is a brushless DC motor that has the maximum life span, which is a deadly weak point of general DC motors. In addition, it refers to a controlled electric motor capable of controlling speed, torque, and distanced. Recently, multiphase BLDC motors with high output characteristics per unit area have been used in various industrial fields, ships, aerospace and military fields. As the drive control algorithm of the multiphase BLDC motor, analog control such as hysteresis control and minimum switching time controller (MSTC) control is mainly applied. However, in analog control, the torque pulsation of the motor is large due to the irregular switching pattern, and the unbalance of the current due to the constant unbalance of the motor is inevitable. In addition, since the instantaneous current value is continuously sampled and switched, the current sampling failure due to noise is likely to cause fatal damage to the motor. In order to make up for this drawback, attempts to reduce the effects of noise by applying filters to inverters have been actively conducted. However, as the cost of filter design adds up, the overall production cost increases and remains difficult to resolve due to current unbalance and large torque pulsation analog control.

An object of the present invention is to provide a technical solution capable of driving a polyphase BLDC motor (Y connection) with a digital control algorithm.

According to an aspect of the present invention, a phase current control apparatus for driving a multiphase BLDC motor includes a speed controller configured to calculate and output a command current to zero a difference between a command speed and a polyphase BLDC motor speed; Current controllers for calculating and outputting command voltages for zeroing the difference between the output command current and the current flowing in each phase of the multiphase BLDC motor, and the command voltage corresponding to each phase of the BLDC motor depends on the output command voltages. A pulse width modulator for outputting a pulse width control signal to be applied, and a multi-phase inverter for interrupting switching elements connected to each phase of the multiphase BLDC motor according to the pulse width control signal and applying a corresponding command voltage to each phase of the BLDC motor. do.

The current controller preferably compensates the estimated back electromotive force of the BLDC motor to the output, and may limit the integral value of the current control integrator included therein according to the output limit value of the current controller.

Furthermore, the phase current control apparatus for driving a multiphase BLDC motor further includes a conduction control unit for controlling the multiphase inverter to conduct all phases of the multiphase BLDC motor or to conduct all phases except one phase.

The present invention can make full use of the advantages of analog control, and can compensate for the disadvantages of fatal malfunction due to an increase in motor torque pulsation, current pulsation increase, noise, etc. due to an irregular switching frequency according to the analog control. That is, the digital control according to the present invention can be easily applied to a variety of existing control algorithms, and it is easy to prepare for noise because it is driven at a fixed switching frequency. This means that the filter design is easy. It also reduces torque pulsation. Furthermore, the present invention reduces the unbalance of the phase current flowing through each winding according to the unbalance of the R and L values generated in the manufacturing process of the motor.

1 is a driving circuit diagram of a seven-phase BLDC motor.
2 is a block diagram of a multiphase BLDC motor drive system.
3 is a conceptual diagram of a current controller of an electric motor.
4 is a conceptual diagram of a current controller that considers counter electromotive force as disturbance.
5 is a frequency response characteristic diagram of an open loop transfer function.
6 is a frequency response characteristic diagram of a closed loop transfer function.
7 is an illustration of a proportional integral current controller.
8 is an exemplary diagram for limiting an integral value in the case of a proportional gain controller.
9 illustrates an anti-wingup controller.
10 is a block diagram of a phase current control device for driving a multi-phase BLDC motor according to an embodiment of the present invention.

BRIEF DESCRIPTION OF THE DRAWINGS The foregoing and further aspects of the present invention will become more apparent from the following detailed description of preferred embodiments with reference to the accompanying drawings. Hereinafter, the present invention will be described in detail to enable those skilled in the art to easily understand and reproduce the present invention.

1 is a driving circuit diagram of a seven-phase BLDC motor as an example of a multiphase BLDC motor.

As shown, the driving circuit of the multiphase BLDC motor is largely composed of a DC voltage unit, an inverter unit for converting the DC voltage into the driving power form of the BLDC motor, and a motor. In addition, such a motor is equivalent to a back electromotive force generated in proportion to the rotational speed of the motor and an inductance component and resistance inside the motor. In the driving circuit shown in FIG. 1, if KVL (Kirchhoff's Voltage Law) is applied to an electric circuit including an inverter output voltage V, a motor counter electromotive force component, a magnetic inductance of each phase, and mutual inductance and resistance component, the circuit equation of the driving system is as follows. It can be expressed by Equation 1.

[Equation 1]

V _x : Inverter output voltage (x = a, b, c, d, e, f, g)

L _{x :} Magnetic inductance of each phase (x = a, b, c, d, e, f, g)

M _αβ : each mutual inductance (α, β = a, b, c, d, e, fg)

i _x : current of each phase (x = a, b, c, d, e, f, g)

e _x : back EMF of each phase (x = a, b, c, d, e, f, g)

v _no = Voltage between GND and neutral

In this case, assuming that the counter electromotive force and current of each phase are ideal waveforms, the mutual inductance of any one phase is affected by the other phase to obtain the same effect, so that the voltage equation of each phase is expressed by Equation 2 as follows. do.

[Equation 2]

Where x = a, b, c, d, e, f, g.

L _x -M _x = L _x R _x = R In this case, it is assumed that the effects of the magnetic inductance and the mutual inductance of each phase are in equilibrium. As a formula, it can be expressed as follows.

&Quot; (3) "

V _a = (Ls + R) i _a + e _a + v _no

&Quot; (4) "

V _b = (Ls + R) i _b + e _b + v _no

[Equation 5]

V _c = (Ls + R) i _c + e _c + v _no

&Quot; (6) "

V _d = (Ls + R) i _d + e _d + v _no

[Equation 7]

V _e = (Ls + R) i _e + e _e + v _no

[Equation 8]

V _f = (Ls + R) i _f + e _f + v _no

[Equation 9]

V _g = (Ls + R) i _g + e _g + v _no

In Equation 3 to Equation 9, L means inductance of each phase equivalent, and R means resistance of each phase equivalent.

In addition, since the total current becomes zero at the neutral point n of the electric motor, the relationship as shown in Equation 10 is established.

[Equation 10]

i _a + i _b + i _c + i _d + i _e + i _f + i _g = 0

By substituting Equations 3 to 9 into Equation 10, Equation 11 can be obtained.

&Quot; (11) "

If the loss is ignored, the electrical output power is the same as the mechanical output, so if the waveform of the counter electromotive force and the phase current is ideal, the output relation is modeled as shown in Equation 12 according to the relationship between the output torque and the rotational speed.

[Equation 12]

T _e : Output torque

P _e : electrical output power

ω _m : motor angular velocity

E _s : Maximum value of back EMF in each phase

I _s : Maximum value of each phase current

In this case, since the electrical output of the three-phase BLDC motor of the same rating is represented by Equation 2, it can be seen that according to Equation 12, the multi-phase BLDC motor shows hundreds of percent increase in output compared to the BLDC motor having the same rating. . The equation of motion of the motor is modeled by the output torque at this time.

[Equation 13]

J _m : moment of inertia

B _m : Viscous friction coefficient

T _L : Load torque

When the block diagram of the motor is drawn by the modeling equations of Equation 1 to Equation 13, the equation of motion of the speed of the multiphase motor is the same as that of the three-phase motor, and the electrical circuit is controlled by a constant. You can see that only the modeling changes.

In general, in order to implement high performance control of a multi-phase BLDC motor, a current controller and a speed controller that perform perfect operation are basically required. First, the basic operation of the current controller will be described and the gain selection method of the current controller will be described. Looking at the operation of the motor from the viewpoint of the current controller, it can be concluded that the current of the motor can be controlled by controlling the voltage input to the terminal. Thus, the general current control algorithm is configured to feed back the terminal current and compare it with the current command and apply a corresponding voltage to the motor as shown in FIG. In this case, the function in FIG. 3 serves as a controller for generating a voltage from the difference between the voltage command and the actual current. The controller generates a voltage output by proportional to the current difference as shown in Equation (14). It consists of terms that generate a voltage from the integral of the current difference.

[Equation 14]

Where K _pc is proportional gain and K _ic is integral gain.

On the other hand, in the aspect of the current controller, the counter electromotive force of the motor serves as a kind of disturbance to the current controller, the configuration diagram in which the current control target system and the current controller is simplified in the electric RL circuit at this time is shown in FIG. have. The transfer function is calculated based on FIG. 4 and is represented by Equation 15.

[Equation 15]

Equation 15 may be represented by Equation 16 as follows.

[Equation 16]

The numerator of the transfer function is the same as (17) and has one defined zero.

[Equation 17]

In addition, the denominator of the transfer function can be expressed by the pole as shown in Equation (18).

&Quot; (18) "

Eventually the poles of the system are determined by the gain of the current controller, which determines the characteristics of the system. Where the pole is located on the s-plane determines the stability, damping ratio, and response speed of the system.

On the other hand, there are a number of ways to calculate the gain of the current controller depending on the overall configuration of the motor control system. For example, a control gain can be obtained based on a gain selection method using a frequency response curve. First, as part of selecting the control gain, a time constant of the current control system is introduced to convert the control function of Equation 14 into Equation 19 as follows.

&Quot; (19) "

When selecting the gain, analyze the frequency characteristic at this time, assuming that the controller consists only of the proportional gain K _pc proportional to the magnitude of the difference, and then again use the frequency characteristic of the controller including the integral term from the actual proportional integral ( PI) Select the current controller gain. Therefore, we first consider the open loop characteristics of the current control system when the controller is configured only with proportional gain. Considering the case where there is no current feedback in FIG. 3, an open loop transfer function such as Equation 20 can be obtained.

&Quot; (20) "

From this, the closed loop transfer function of the control system with current feedback is obtained from Equation 21 below.

&Quot; (21) "

In this case, assuming that the current controller is configured in the rotational coordinate system, the frequency can be set to zero since all physical quantities are given as direct current components without frequency components in a steady state. Therefore, the transfer function at this time is equal to (22).

[Equation 22]

Equation 22 means that even in the steady state, the actual current does not exactly match the current command, that is, the result is not 1. Of course, assuming that the proportional gain is sufficiently large compared to the fixed resistor, the gain may be close to 1, but the inverter system operating according to the voltage command, which is generally the output of the current controller, is practically impossible because the output voltage is limited. Therefore, the integral term must be added to the controller to remove the steady-state difference and to perform the stable current control. When using a general PI controller with an integral term, the voltage and current relations are shown in Equation 23.

&Quot; (23) "

Using this, the open loop transfer function of the current control system can be obtained from Equation (24).

&Quot; (24) "

At this time, to simplify the equation, the time constant of the controller is selected to be the same as the time constant of the control target system.

[Equation 25]

By doing so, the transfer function of equation (24) is simply described as in equation (26).

[Equation 26]

The frequency response curve at this time is shown in FIG. In this response curve, the crossover frequency in the open loop is given by Equation 27.

[Equation 27]

At this time, the upper limit of the constant gain of the system in the closed loop frequency response curve becomes this cross-angle frequency (the frequency at which the gain is reduced by 3 dB). That is, when the closed loop transfer function is obtained from Equation 26, it is described as Equation 28. And the frequency response curve for this is shown in FIG.

[Equation 28]

As can be seen from FIG. 6, the current controller operates with gain 1 up to a frequency band corresponding to the cross-angle frequency of the open loop, and from above, the phase converges to 90 ° while the gain gradually decreases. Therefore, when selecting the gain of the current controller, it is very important to know these characteristics in advance and select the cross-angle frequency in the open loop transfer function to an appropriate value for the system. At this time, increasing the cross-angle frequency increases the current controller's performance. However, selecting too large a frequency can cause the system to diverge due to external factors such as noise and propagation delay of signals excluded from the frequency response analysis. to be.

In particular, this constraint is mainly determined by the current sampling period of the digital controller, which is usually determined by the switching frequency of the inverter. Experimental results show that in systems with a switching frequency of 10kHz, cross-angle frequencies can be selected up to about 6000rad / sec. From the analysis of the current controller using the above frequency response curve, the gain of the current controller can be selected as follows.

[Equation 29]

&Quot; (30) "

을 전류 제어기의 출력에 전향보상(Feed-forward Compensation)함으로써, 도 7의 블록에서 역기전력 변동에 의한 영향을 아예 상쇄시킬 수도 있다. 이와 같이 역기전력의 영향을 배제하고 비례 적분 제어기의 영점(zero)이 시스템의 극점(pole)을 상쇄하도록 설계하면, 전류 제어 기 폐루프 응답 특성을 1차 지연요소와 같이 되도록 할 수 있다.On the other hand, when the inertia of the motor is large and the change in the back EMF is sufficiently slow compared to the responsiveness of the current controller, the current controller can be designed for the R _s L _s load of the armature circuit as shown in FIG. If you can estimate the back EMF by measuring the speed, etc.

By forward-compensating the output of the current controller, the influence of the back EMF fluctuation in the block of FIG. 7 may be canceled at all. In this way, if the zero of the proportional integral controller cancels out the pole of the system by eliminating the effect of back EMF, the current control closed-loop response characteristic can be made as the first delay element.

When the proportional gain and the integral gain are selected as in Equation 29 and Equation 30, the transfer function between the output current and the current command value is as in Equation 31.

Equation 31

,

,

In Equation 31, the frequency bandwidth of the current controller is given by ω _c . Therefore, when the frequency band of the desired current controller is determined, and the gain of the PI controller is determined using Equation 29 and Equation 30, the current controller without over-shoot is given as given by Equation 31. Can be designed.

그 값이 제한되지 않고 제어기의 제한폭(상한, 하한)을 넘어 쌓이게(wind-up) 된다. 이 경우, 제어기 입력의 부호가 반전(reverse)될 경우에도 쌓여있는 적분기의 내부 적분 값으로 인하여 제어기 출력이 입력에 대해 제대로 반응하지 않거나 그 값이 느려지는(sluggish) 현상이 발생한다. 이러한 현상을 방지하게 위해서는 적분기 내부의 값을 제어기 출력의 제한 값에 따라 적절히 제한할(anti-windup) 필요가 있다.On the other hand, the physical inputs and outputs all have an upper limit value and a lower limit value due to physical limitations. The power amplifier is simply implemented using a current control limiter because the maximum and minimum values of the voltage that can be output by the input AC voltage or the input DC voltage are limited. However, if a current control integrator is included, the integral value of the integrator is

Its value is not limited and will wind up beyond the controller's limit (upper limit, lower limit). In this case, even when the sign of the controller input is reversed, the controller output does not respond properly to the input or sluggish due to the accumulated internal integral value of the integrator. To prevent this phenomenon, it is necessary to appropriately anti-windup the value inside the integrator according to the limit value of the controller output.

In the case of an analog controller using an operational operational amplifier, the output of the integrator is automatically limited by the magnitude of the power supply (+ Vcc, -Vcc) of the analog circuit. If you want to limit the controller output to some value within the supply voltage, you can limit the integrator's capacitor voltage (integral value) by a simple circuit using a zener diode.

은 전향(Feed forward) 보상되어 전류 제어 특성을 향상시킬 수 있다. 도 9에서 K_a는 Anti-windup 제어기의 이득이며, 통상

정도로 설정하는데, 요구되는 제어 특성에 따라서

을 기준으로 하여

정도에서 가변될 수 있다.In case of Digital Signal Processor (DSP) using Fixed Point Operation, the output of the integrator is compared with the controller output by the saturation function of the calculated variable and proper scaling technique. You can limit the value. However, when implementing a controller using a DSP using floating point operation or implementing a computer simulation program, an appropriate method of limiting the output of the integrator must be used. In the case of the PI controller, the limit of the integrator output can be implemented through various techniques as shown in FIG. 8, but FIG. 9 shows the best control characteristics. Estimated Back EMF in Each Controller Configuration

Feed forward compensation can improve current control characteristics. In Figure 9 K _a is the gain of the anti-windup controller, typically

Set to a degree, depending on the required control characteristics

On the basis of

Can vary in degree.

10 is a block diagram of a phase current control device for driving a multi-phase BLDC motor according to an embodiment of the present invention. Hereinafter, it will be described on the assumption that it is a 7-phase BLDC motor.

The seven-phase BLDC motor 100 rotates the rotor by varying the electromagnetic shape in a non-contact manner to the rotor provided with the stator as a permanent magnet. The current detector 200 is configured to detect a current applied to each phase of the seven-phase BLDC motor 100. The current detector 200 may be composed of a plurality of current sensors attached to the winding of each phase. The position detector 300 detects the rotational position of the rotor constituting the 7-phase BLDC motor 100. In one embodiment, the position detector 300 is a hall sensor. The hall sensor is attached to the 7-phase BLDC motor 100, and outputs a certain amount of current according to the rotation of the rotor constituting the 7-phase BLDC motor 100. The hall sensor is used to detect the rotational position of the rotor constituting the 7-phase BLDC motor 100, and thus may be replaced by an encoder using a photodiode or a magnetoresistive element. The speed measuring unit 400 receives the rotation position detection signal of the rotor from the position detector 300 and calculates the actual rotation speed ω of the rotor.

)을 전향 보상(Feed-forward)하는 기능과 안티 와이드 업(Anti-wideup) 기능 중 적어도 하나를 포함할 수 있다.The speed controller 500 is a PI controller for controlling the rotor speed of the 7-phase BLDC motor 100, and the command speed ω ^* and the actual rotational speed of the rotor constituting the 7-phase BLDC motor 100 (ω). Calculate and output the current (torque) value needed to make the difference of) zero. The current controller 600 is composed of a plurality. That is, if the motor is N phase, there are a total of N current controllers. Since the electric motor is seven phases, seven current controllers 600 are configured. Each current controller 600 is a PI controller for controlling the current flowing in each phase of the 7-phase BLDC motor 100, and each of the DC command current I ^* and the 7-phase BLDC motor 100, which are outputs of the speed controller, is controlled. The voltage to be applied to the seven-phase BLDC motor 100 is calculated and output in order to make the difference between the phase currents I _x flowing in the phases (A-G phase) zero. The command voltages Va _a ^* to V _g ^* of each phase calculated by the current controller 600 are input to the pulse width modulator 700. This command voltage (V _a ^* ~ V _g ^* ) determines the duty of the pulse width modulation (PWM). That is, the on / off time of the switching elements constituting the seven-phase inverter 800 is determined. On the other hand, the current controller 600 is a counter electromotive force (

) May include at least one of a function of forward-forwarding and an anti-wideup function.

Pulse width modulation unit 700 calculates the number of pulses corresponding to the input reference voltage _{^{(V a * ~ V g *}} ) and outputting a pulse width control signal corresponding to the number of the calculated pulse to the 7-phase inverter (800) . The pulse width control signal is an inverter drive control signal for controlling the switching elements of the seven-phase inverter 800, so that the command voltages Va _a ^* to V _g ^* are supplied to each phase of the seven-phase BLDC motor 100. This is a control signal. The seven-phase inverter 800 is composed of a plurality of switching elements. The switching elements constituting the seven-phase inverter 800 operate on / off according to the pulse width control signal output from the pulse width modulator 700. Since each of the switching elements are reference voltage _{^{(V a * ~ V g *}} ) on / off according to the pulse width control signal corresponding to each of the switching on / off time of each switching element may be different.

Accordingly, in each phase of the seven-phase BLDC motor 100 by the PWM duty corresponding to each command voltage (V _a ^* to V _g ^* ), the command voltages (V _a , V _b , V _c , V _d , V _e , V _f , V _g ) are applied, and this command voltage (V _a , V _b , V _c , V _d , V _e , V _f , V _g ) is the respective reference voltage (V ^* ) determined in the current controllers 600. ) Can have different values. When each command voltage is applied to each phase of the 7-phase BLDC motor 100, each winding of the 7-phase BLDC motor 100 has the same or different amount of current (I _a , I _b , I _c , I _d). , I _e , I _f , I _g ) will flow.

According to a further aspect of the present invention, the phase current control device further includes a conduction control unit 900. The conduction control unit 900 performs full phase conduction or n-1 phase conduction control. The full-phase conduction method is a method of conducting the full phase of the 7-phase BLDC motor 100. In addition, the n-1 phase conduction method is a method of non-conducting one phase of the 7-phase BLDC motor 100 and conducting all other phases. Specifically, in the case of the full-phase conduction method, the conduction control unit 900 has all the switching elements of the 7-phase inverter 800 pulsed by the pulse width modulator 700 so that a command voltage is applied to all phases of the 7-phase BLDC motor 100. Activate on / off according to the width control signal. In the n-1 phase conduction method, the conduction control unit 900 deactivates any one switching element of the 7-phase inverter 800 such that a command voltage is not applied to any one of the 7-phase BLDC motors 100.

According to a further aspect of the present invention, the phase current control device further includes an electric motor protector 1000. For example, when the rated current of the motor is 10 [A], when the current of several hundreds [A] flows, the motor burns out. Also, for example, an electric motor capable of rotating up to 1000 RPM may be mechanically dangerous if it is rotated at a high speed such as 2000 RPM or 5000 RPM. Therefore, to prevent this, the motor protection unit 1000 is added to the phase current control device. In one embodiment, the motor protection unit 1000 receives the speed value measured from the speed measuring unit 400, and deactivates and controls the seven-phase inverter 800 when the input speed value is greater than or equal to a preset speed threshold. . In one embodiment, the motor protection unit 1000 compares the phase current value detected by the current detector 200 with a preset current threshold, and if the comparison results in a higher phase current than the current threshold, deactivates the seven-phase inverter 800. To control.

Meanwhile, among the configurations of the phase current control device described above, the speed measuring unit 400, the speed controller 500, the current controller 600, the pulse width modulator 700, the conduction control unit 900, and the motor protection unit 1000 are included. The configurations can be implemented with a controller. In one embodiment, these configurations are implemented as a single microprocessor.

So far I looked at the center of the preferred embodiment for the present invention. It will be understood by those skilled in the art that various changes in form and details may be made therein without departing from the spirit and scope of the invention as defined by the appended claims. Therefore, the disclosed embodiments should be considered in an illustrative rather than a restrictive sense. The scope of the present invention is shown in the claims rather than the foregoing description, and all differences within the scope will be construed as being included in the present invention.

100: 7-phase motor 200: current detector
300: position detection unit 400: speed measurement unit
500: speed controller 600: current controller
700: pulse width modulator 800: 7-phase inverter
900: conduction control unit 1000: motor protection unit

Claims (8)

A speed controller for calculating and outputting a command current for zeroing the difference between the command speed and the speed of the polyphase BLDC motor;
Current controllers for calculating and outputting command voltages for zeroing a difference between the outputted command current and a current flowing in each phase of the polyphase BLDC motor;
A pulse width modulator for outputting a pulse width control signal such that a command voltage corresponding to each phase of the BLDC motor is applied according to the output command voltages; And
A multiphase inverter for interrupting switching elements connected to each phase of the polyphase BLDC motor according to the pulse width control signal and applying a corresponding command voltage to each phase of the BLDC motor;
Phase current control device for driving a multi-phase BLDC motor comprising a. The method of claim 1,
A current detector for detecting a current flowing in each phase of the BLDC motor;
Phase current control device for driving a multi-phase BLDC motor, characterized in that it further comprises. The method of claim 1,
And the current controllers forward-compensate the estimated back EMF of the BLDC motor to the output. The method of claim 1,
And the current controllers limit the integral value of the current control integrator included therein according to the output limit value of the current controller. The method of claim 1,
A conduction controller controlling the multiphase inverter to conduct all phases of the multiphase BLDC motor or to conduct all phases except one phase;
Phase current control device for driving a multi-phase BLDC motor, characterized in that it further comprises. The method of claim 1,
A motor protection unit which acquires a current value flowing in each phase of the BLDC motor and controls the switching elements of the multi-phase inverter to be turned off when the obtained current value is greater than or equal to a threshold;
Phase current control device for driving a multi-phase BLDC motor, characterized in that it further comprises. The method of claim 1,
A motor protection unit for turning off switching elements of the polyphase inverter when the speed of the BLDC motor is greater than or equal to a threshold;
Phase current control device for driving a multi-phase BLDC motor, characterized in that it further comprises. The method according to any one of claims 1 to 7,
The phase current control device for driving a multi-phase BLDC motor, characterized in that the multi-phase BLDC motor is a 7-phase BLDC motor.

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