KR20080020276A - Dc-to-dc conversion apparatus adopting sllc series resonant converter - Google Patents

Abstract

A DC-to-DC conversion apparatus adopting SLLC(Secondary Logical Link Control) series resonant converter is provided to obtain a high voltage gain characteristic by adding a parallel inductor to a secondary side of a transformer. A series resonant converter circuit includes a plurality of main switching elements(Q1~Q4) and a transformer(TR). The transformer has a primary winding and a secondary winding to generate induction voltage by current interruption of the main switching elements. A resonant device is connected to the secondary winding of the transformer in series. The transformer does not have a gap. The resonant device includes a capacitor and a primary leakage inductance and a secondary leakage inductance of the transformer. A first terminal of the capacitor is connected to the secondary winding of the transformer and a second terminal thereof is connected to an output side.

Description

SLC serial resonant converter circuit and DC converter using this circuit {DC-to-DC conversion apparatus adopting SLLC series resonant converter}

1 is a schematic circuit diagram of a conventional voltage full bridge DC converter.

2 is an operation waveform diagram of a voltage DC converter as shown in FIG. 1.

3 is a schematic circuit diagram of a current-type DC converter.

4 is an operation waveform diagram of the current-type DC converter as shown in FIG.

5 is a circuit diagram of a conventional LLC series resonant converter to which a series resonant capacitor is applied on a primary side.

Fig. 6 is a circuit diagram of an embodiment of the SLLC series resonant converter of the present invention to which a series resonant capacitor and a parallel inductor L ₂ are applied on the secondary side.

7 is an operational waveform diagram of an embodiment of the SLLC series resonant converter of the present invention.

8 to 12 are operation mode explanatory diagrams for explaining the operation of one embodiment of the SLLC series resonant converter of the present invention.

13 is an equivalent circuit diagram of FIG. 6.

14 is a circuit diagram showing another embodiment of the SLLC series resonant converter of the present invention.

15 is an equivalent circuit diagram of FIG. 14.

16 is a voltage gain characteristic curve for FIG.

FIG. 17 is an experimental waveform diagram (50V /) between the primary terminal voltage V _ab and the current I _T1 , the secondary resonant current I _D , and the parallel inductor voltage V _{L2 in the} embodiment shown in FIG. 14. div., 50 A / div., 500 V / div., 2.5 A / div., 4 μs / div).

18 is a prototype photograph of the embodiment shown in FIG. 14.

19 shows another embodiment of the SLLC series resonant converter of the present invention.

20 is a circuit diagram of the modified embodiment of FIG.

21 is an equivalent circuit of FIG.

22 is an input / output voltage gain characteristic curve of the circuit shown in FIGS. 19 and 20. FIG.

Fig. 23 is a circuit diagram showing a main circuit of a boost type push-pull DC converter as another embodiment of the SLLC series resonant converter of the present invention.

24 is a circuit diagram showing a main circuit of a boost type half bridge DC converter as another embodiment of the SLLC series resonant converter of the present invention.

Fig. 25 is a circuit diagram of an application example when using a lamp load in the present invention.

The present invention relates to a direct current conversion device, that is, a DC / DC converter. More specifically, the present invention relates to a SLLC series resonant converter circuit and a direct current converter using the same as a main circuit.

In the high frequency step-up DC converter that converts a low input voltage to a high output voltage, a conventional converter main circuit can be generally divided into a voltage-fed converter and a current-fed converter.

FIG. 1 illustrates a circuit of a voltage full bridge DC converter, and FIG. 2 shows an operating waveform diagram of the voltage full bridge DC converter. [Reference: Vlako Vlatkovic, Juan A. Sabate, Raymond B. Redley, Fred C Lee, Bo H Cho "Small-Signal Analysis of the Phase-Shifted PWM Converter", IEEE Trans. Power Electron. vol. 7, no, pp. 128-135, 1992; Lai, J.S "A High-Performance V6 Converter for Fuel Cell Power Conditioning System" Vehicle Power and Propulsion, 2005 IEEE Conference, sept 2005, pp 624-630].

Voltage converters are generally stable to primary bridge main switching devices (S1, S2, S3, S4) by applying a phase-shift PWM method in a medium-to-high voltage voltage full bridge DC converter. Since the switching loss can be reduced by providing the zero voltage switching operation, it is widely applied as a boost type DC converter. However, the voltage-type full bridge DC converter requires a high transformer turn ratio for boosting, resulting in a duty loss (ΔD) due to an increase in the transformer leakage inductance.

,

이다.At this time, the output voltage

,

to be.

Since the circulating current flowing during the duty loss period DELTA D does not transfer energy to the secondary side but only flows through the primary bridge main switching element, only the conduction loss increases, resulting in a decrease in efficiency characteristics.

3 is a circuit diagram of a current-type DC converter, and FIG. 4 shows operation waveforms of the current-type DC converter. [Note: L. Zhu, K. Wang, F.C. Lee, J. Lai, "Design Considerations of Start-up Process for Active-Clamp Isolated Full-Bridge Boost Converter", VPEC Seminar, 1999]

The current converter attaches a boost inductor (L) to the input terminal to accumulate energy in the boost inductor (L) during the short-circuit period, and converts the sum of the voltage accumulated in the input voltage (Vin) and the boost inductor (L) during the transfer period. It is applied to the primary side to transfer energy. At this time, since the current flowing through the boost inductor L is a predetermined DC current having a small ripple current, it has a desirable feature as a DC DC-DC booster supplying a low input voltage and a large current.

However, in FIG. 3 and FIG. 4, the current flowing through the boost inductor L causes parasitic capacitance and parasitic vibration in the switching element when the main switching elements S1, S2, S3, and S4 turn off, and is switched. Increase the voltage stress of the device. To solve this problem, an RCD snubber circuit is attached to both ends of the DC connection of the switching element so that the current path flowing through the boost inductor through the snubber diode and the snubber capacitor is applied to the switching element when the switching element is turned off. The surge voltage can be clamped.

However, the application of the RC or RCD snubber circuit consumes the energy charged in the snubber capacitor through the snubber resistor during discharge, so the higher the switching frequency, the greater the switching loss, thereby reducing the energy conversion efficiency.

In consideration of the characteristics and problems of the conventional main circuit described above, both the boost mode and the step-down mode can be operated, and a wide range of input / output voltage control can be performed in a relatively narrow frequency control range. A step-up DC converter using an LLC series resonant converter has been proposed, which has several advantages such as soft switching of switching elements. This converter circuit is shown in FIG. [Note: Bo. Yang, F.C. Lee, A.J. Zhang, "LLC Resonant converter for Front End DC / DC Conversion," IEEE-APEC'2002, Vol. 2, pp. 1108-1112, 2002; Guisong Huang, Alpha J. Zhang, Yilei Gy, "LLC Series Resonant DC-TO-DC Converter", Patent No .: US 6,344,9779 B1, Feb. 5, 2002]

The LLC series resonant converter was developed to further increase power density and efficiency characteristics, and uses a transformer integrating an inductor and a transformer into one transformer. However, since the LLC series resonant converter of FIG. 5 is operated by stepping up a voltage from a low input voltage to a high output voltage, a large current flowing to the primary side as long as a series resonant capacitor, which is one of the resonance means of the LLC series resonant converter, is used on the primary side. Due to the deterioration of the series resonant capacitor may cause a change in characteristics, there are a number of problems to use in the primary side due to the difficulty of high integration due to the increase in the size of the capacitor and the increase in the application cost.

The present invention has been made to solve the above problems, an object of the present invention is to provide a SLLC series resonant converter circuit that can reduce the size and cost of the resonator means by placing the resonator means of the LLC series resonant converter on the secondary side of the transformer and It is to provide a DC converter adopting this.

Another object of the present invention is to provide a high voltage gain characteristic by adding a parallel inductor on the secondary side of a transformer to allow current for zero voltage switching of a primary juice switching element of an LLC series resonant converter. It is to provide a circuit and a DC converter employing the same.

Summary of the Invention

In order to achieve the above object, the present invention discloses a SLLC series resonant converter circuit of a novel configuration. Hereinafter, in order to distinguish the LLC series resonant circuit used in the converter circuit of the present invention from the conventional one, it will be referred to as a SLLC (Secondary LLC) series resonant circuit.

The SLLC series resonant converter circuit according to the first aspect of the converter circuit includes a plurality of juice switching elements and a transformer having a primary winding and a secondary winding to generate an induced voltage by the current interruption of the juice switching element. It includes a resonating means connected in series with the secondary winding of the transformer.

The resonator means includes a capacitor having a first end connected to one end of the secondary winding of the transformer and a second end connected to an output side, and resonating with the primary leakage inductance and the secondary leakage inductance of the transformer. .

In addition, the resonator means preferably further comprises a parallel inductor connected to the second end of the capacitor.

In addition, the resonating means may further include a series inductor connected between the output terminal and the connection portion and the second end of the capacitor and the parallel inductor.

The SLLC series resonant converter circuit according to the second aspect of the converter circuit according to the present invention has a plurality of juice switching elements and a primary winding and a secondary winding to generate an induced voltage by the current interruption of the juice switching elements. It includes a transformer, comprising a resonating means connected in series to the secondary side winding of the transformer, in series with the resonating means, characterized in that for use by connecting a LCTR (Loosely Coupled Transformer) transformer having a gap.

The resonating means includes a capacitor having a first end connected to one end of the secondary side winding of the transformer and a second end connected to an output side.

)및 누설인덕턴스(

)와 함께 공진작용을 한다.In this converter circuit, the capacitor is a magnetizing inductance of the LCTR (

) And leakage inductance (

Resonance with

The SLLC series resonant converter circuit described above can be employed in various types of DC converters, that is, DC / DC converters for converting the direct current of the first voltage input to the input terminal into the direct current of the second voltage and outputting the output to the output terminal. Examples of the DC converter that can adopt the SLLC series resonant converter circuit include full bridge, half bridge, and push-pull types.

As described above, in the DC converter adopting the SLLC series resonant converter circuit of the present invention, a DC biasing phenomenon may occur due to the DC voltage input to the input terminal. Therefore, in order to prevent this, a blocking capacitor is placed on the primary winding of the transformer of the converter circuit. It is preferable to connect further.

Referring to the configuration of the present invention described above more visually as follows.

,

,

,

)의 영전압 스위칭을 위한 전류를 흘리기 위해 2차측에 병렬인덕터(L₂)를 추가하여 SLLC 직렬공진 컨버터의 특성을 얻도록 하였다. 이와 같이 병렬인덕터(L₂)의 추가에 의해, 이 병렬인덕터(L₂) 값에 따라 높은 전압 이득특성을 갖게 되어 변압기의 적은 턴수비에도 높은 출력전압을 얻을 수 있고 넓은 입력전압 범위와 모든 부하 범위에서 사용 가능한 장점이 있다. 따라서 본 발명에서 제안한 SLLC 직렬공진 컨버터는 변압기(TR)의 1, 2차측 누설인덕턴스 및 변압기(TR) 2차측에 설치한 커패시터(C_S)와의 직렬공진과 병렬인덕터(L₂)의 추가로 도 5의 종래 LLC 직렬공진 컨버터에서 발생되는 문제점을 해결할 수 있다. As shown in FIG. 6, the resonating means of the SLLC series resonant converter is positioned on the secondary side of the transformer TR to configure the main circuit, thereby reducing the size and cost of the series resonant capacitor C _S. In order to make magnetizing inductance as large as possible in general transformers, no gap is placed on the transformer TR, and the primary leakage inductance L _l1 and the secondary leakage inductance L _l2 are connected in series. The winding method used as is applied. In particular, the bridge juice switching element on the primary side (

,

,

,

The parallel inductor (L ₂ ) was added to the secondary side to obtain the characteristics of the SLLC series resonant converter. Thus to obtain a high output voltage by the addition of the parallel inductor (L _2), the parallel inductor (L ₂₎ is given a high gain characteristic according to the value in the low-turn defense of the transformer and a wide input voltage range, and all the load There are advantages available in the range. Therefore, the SLLC series resonant converter proposed in the present invention further shows a series resonance and a parallel inductor (L ₂ ) between the primary and secondary leakage inductance of the transformer (TR) and the capacitor (C _S ) installed on the secondary side of the transformer (TR). 5 can solve the problems occurring in the conventional LLC series resonant converter.

Description of the operation of the DC converter according to an embodiment of the present invention

)가 1보다 낮은 불연속모드(Discontinuous Conduction Mode) 영역에서 동작했을 때의 본 발명의 직렬공진 컨버터의 동작파형이다. 이 동작파형들은 변압기 1차측 단자전압

, 변압기 2차측 전류

, 병렬인덕터 전류

, 2차측 공진전류

, 병렬인덕터 양단전압

를 나타내고 있으며 반주기의 동작모드를 5가지 모드로 나누었다. 각 모드의 동작은 도 8~12에 나타내었다. The operation mode of the boost type DC / DC converter according to the embodiment of the DC converter applying the SLLC series resonant converter proposed by the present invention will be described by time. 7 shows the normalized frequency (

) Is an operating waveform of the series resonant converter of the present invention when operating in a discontinuous conduction mode region lower than one. These operating waveforms are the primary voltage of the transformer

Transformer secondary current

Parallel inductor current

, Secondary side resonant current

, Voltage across parallel inductor

The half cycle operation mode is divided into five modes. The operation of each mode is shown in Figs.

) : 도 8에서, 모드 1은 주스위칭소자

,

가 턴오프되고 2차측의 병렬인덕터 전류

가 1차측으로 유도(Reflected)되어 흐르는 변압기 전류 I_T1에 의해 주스위칭소자

,

,

,

의 출력커패시터가 충·방전을 끝마치는 직후의 동작시점이다.

시점 이후 일정하게 흐르는 병렬인덕터 전류

가 변압기 1차측으로 유도되어 주스위칭소자

에 접속된 다이오드(바디다이오드)와 변압기 1차측을 통해 부전류(Negative Current)가 흐른다. 따라서 변압기 1차측 단자 전압은 극성이 변하여 입력전압

이 인가되고 변압기 2차측 전압

도 극성이 변한다. 이때 공진전류

가 정류다이오드

를 도통하여 흐르므로 병렬인덕터 L₂에 출력전압 V_o이 인가되고 병렬인덕터 전류

가 부(Negative)의 피크값에서 선형적으로 일정 기울기로 상승하며 흐른다. t₁ 시점에서 주스위칭소자

의 바디다이오드를 통해 흐르는 부전류 때문에 주스위칭소자

가 영전압 스위칭(Zero Voltage Switching) 동작 상태에서 턴온(Turn-on)하게 된다. Mode 1 (

In Figure 8, mode 1 is the juice switching device

,

Is turned off and the parallel inductor current on the secondary side

Switching element due to transformer current I _T1 flowing through which is induced to the primary side

,

,

,

This is the point of operation right after the output capacitor of has finished charging and discharging.

Parallel Inductor Current Flows Constantly After Time

Is switched to the transformer primary and the juice switching element

Negative current flows through the diode (body diode) connected to the transformer and the primary side of the transformer. Therefore, the voltage of the primary terminal of the transformer is changed in polarity,

Is applied to the transformer secondary voltage

The polarity also changes. Resonant current

Rectification Diode

Flows through and the output voltage V _o is applied to the parallel inductor L ₂ and the parallel inductor current

Flows linearly with a constant slope from the negative peak value. Juice switching element at time t ₁

Due to the negative current flowing through the body diode

Turn-on during zero voltage switching operation.

) : 도 9에서, t₁ 시점에서 주스위칭소자

는 영전압 스 위칭으로 턴온(Turn-on)하고 주스위칭소자를 통해 입력전압(V_in)이 변압기에 인가되므로 변압기 1차측 전류

이 정방향으로 흐르기 시작한다. 모드 1과 모드 2에서 직렬공진은 변압기(TR) 1차측 누설인덕턴스

과 2차측 누설인덕턴스

, 그리고 2차측 직렬공진 커패시터

에 의해 이루어지게 되며, 정류다이오드

는 계속 도통하고 있으며 부하측으로 흐르는 공진전류

는 점차 상승하고 있다. 이때 병렬인덕터 L₂는 직렬공진용 변압기(TR)의 1, 2차측 누설인덕턴스

,

에 비해 큰 값의 인덕턴스를 가지므로 직렬공진특성에 큰 영향을 주지 않는다. 이 모드가 끝나는 시점은 병렬인덕터 전류

가 정(Positive)방향으로 바뀌는 시점 t₂이다. Mode 2 (

9: juice switching device at time t ₁

Is turned on with zero voltage switching and the input voltage (V _in ) is applied to the transformer through the juice switching device, so the transformer primary current

This begins to flow in the forward direction. In mode 1 and mode 2, the series resonance causes the primary leakage inductance of the transformer (TR).

And secondary leakage inductance

, And secondary-side series resonant capacitor

Rectified diode

Continues to conduct and the resonant current flowing to the load side

Is gradually rising. At this time, the parallel inductor L ₂ is the leakage inductance of the primary and secondary side of the series resonance transformer (TR).

,

Since it has a large value of inductance compared to, it does not affect the series resonance characteristics. At the end of this mode, the parallel inductor current

Is the time t ₂ when the positive direction changes.

) : 도 10에서, 모드 3은 모드 2와 같은 특성을 나타내고 있다. 다만 병렬인덕터 전류

가 선형적인 일정 기울기를 가지고 정(Positive) 전류로서 흐르며, 이 모드가 끝나는 시점은 병렬인덕터 전류

와 변압기(TR) 2차측 전류

가 같고, 부하측에 더 이상 공진전류가 흐르지 않으며 직렬공진이 끝나는 시점이다.Mode 3 (

In Fig. 10, mode 3 shows the same characteristics as mode 2. Parallel inductor current

Flows as a positive current with a constant linear slope, and the end of this mode is the parallel inductor current.

And transformer (TR) secondary side current

Is the same, resonant current no longer flows to the load side and it is the point when series resonance ends.

) : 도 11에서, 모드 4는

시점에서 공진이 끝나는 시점으로, 2차측 공진전류

는 흐르지 않고 변압기(TR) 2차측 정류다이오드

는 출력전압(V_o)으로 역바이어스 되어 있다. 이 모드 동안에 주스위칭소자

는 턴온되어 있으므로 입력전압(V_in)이 변압기에 인가되어 있고 이 모드동작에서 공진수단은 변압기(TR) 1차측 누설인덕턴스

과 2차측 누설인덕턴스

그리고 2차측 직렬공진 커패시터

및 병렬인덕터

에 의해 이루어지게 된다. 이 모드는 주스위칭소자

가 턴오프시 끝나게 된다.Mode 4 (

In Figure 11, mode 4 is

Secondary resonance current at the end of resonance at that point

Rectifier diode for transformer (TR) secondary

Is reverse biased to output voltage (V _o ). Juice switching element during this mode

Since is turned on, the input voltage (V _in ) is applied to the transformer and in this mode of operation, the resonator means the primary leakage inductance of the transformer (TR).

And secondary leakage inductance

And secondary series resonant capacitor

And parallel inductors

Will be done by This mode is juice switching device

Ends at turn-off.

) : 도 12에서, 모드 5는

시점에서 주스위칭소자

가 턴오프될 때 시작한다. 일정하게(Constant) 흐르는 병렬인덕터 전류

가 1차측으로 유도(Reflected)되어 흐르는 변압기 1차측 전류

에 의해 주스위칭소자

의 출력 커패시터가 순간적으로 충·방전을 개시하여 끝마친 후 주 스위칭소자

의 바디다이오드를 통해 흐르며 이때 변압기(TR)의 1차측 단자전압 극성(V_ab)이 바뀌게 된다. Mode 5 (

In Figure 12, mode 5 is

Juice switching device

Starts when is turned off. Constant Flowing Inductor Current

Primary current flows through which is reflected to primary side

By juice switching element

The main switching element after the output capacitor of the capacitor starts charging and discharging momentarily

Flow through the body diode of the primary terminal voltage polarity (V _ab ) of the transformer (TR) is changed.

) : 다음 반주기에 대한 동작모드는 모드 1 이후의 동작모드와 같으므로 설명을 생략한다. 위의 동작모드설명과 제 9도의 동작파형에서 볼 수 있듯이 모든 주스위칭소자

는 턴온/오프시 영전압에서 스위칭됨을 확인할 수 있다. 즉, 주스위칭소자

턴온/오프시의 영전압 스위칭은 병렬인덕터

에 저장된 에너지에 의해 흐르는 전류가 1차측으로 유도(Reflected)되어 흐르는 전류

에 의해 결정되므로 부하에 관계없이 영전압 스위칭을 이룰 수 있다. 또한 변압기 2차측 공진전류

는 불연속으로 흐르기 때문에 출력다이오드

,

,

,

의 역회복 특성에 따른 스위칭 손실을 저감할 수 있다. 위의 사항들을 고려할 때 본 발명에서 제안된 SLLC 승압형 컨버터는 낮은 전압정격의 스위칭소자 사용과 스위칭손실 저감 등을 통하여 효율을 개선할 수 있는 특징을 갖는다. Mode 6 (

): The operation mode for the next half cycle is the same as the operation mode after Mode 1, and thus description is omitted. As shown in the operation mode description above and in the operating waveforms of FIG.

It can be seen that the switch at zero voltage when turned on / off. That is, juice switching device

Zero voltage switching at turn on / off

Current flowing through the current flowing by the energy stored in the primary side

It is determined by, so zero voltage switching can be achieved regardless of the load. Also, transformer secondary side resonant current

Output diode because it flows discontinuously

,

,

,

The switching loss due to the reverse recovery characteristic can be reduced. Considering the above, the SLLC boost converter proposed in the present invention has the characteristics of improving efficiency through the use of a low voltage rating switching element and reducing switching losses.

I / O voltage gain characteristics of the circuit according to the present invention

An equivalent circuit of the secondary side SLLC series resonant converter shown in FIG. 6 is shown in FIG. 13 to obtain the input / output voltage gain characteristics of the circuit proposed in the present invention.

In addition, as shown in Figure 14 for the experimental implementation of the proposed circuit and to reduce the transformer loss in the main circuit configuration circuit using two transformers, the transformer primary side is connected in parallel to the primary current of the transformer 1 It was reduced to / 2, and the secondary side of the transformer was connected in series so that the required output voltage was obtained even at a low turn ratio. The equivalent circuit for this is shown in FIG.

는 주스위칭소자

의 스위칭 동작에 의해서 얻어진 1차측 구형파 단자전압을 권선비를 고려하여 2차측으로 반영한 등가전원이며,

와

(도 13),

(도 15),

(도 13),

(도 15)는 2차측 공진을 위한 직렬 커패시터와 턴수비(

)를 고려하여 2차측으로 반영한 변압기(TR) 1차측 누설인덕턴스와 2차측 누설인덕턴스이다. 그리고

는 변압기 2차측에 추가한 병렬인덕터이고,

는 부하저항, 정류다이오드 및 커패시터 필터를 등가화한 등가 AC부하저항이다. In the equivalent circuit of FIG. 13 and FIG. 15

A main switching element

Equivalent power source that reflects the primary square wave terminal voltage obtained by the switching operation to the secondary side in consideration of the winding ratio.

Wow

(FIG. 13),

(FIG. 15),

(FIG. 13),

Figure 15 shows a series capacitor for the secondary side resonance Turn-rate

) Is the primary leakage inductance and secondary leakage inductance of the transformer (TR) reflected to the secondary side. And

Is the parallel inductor added to the transformer secondary,

Equivalent AC load resistance equals load resistance, rectifier diode and capacitor filter.

는 변압기 등가모델을 이용하여 변압기(TR) 2차측에서 권선 비(

)를 고려하여 바라본 1차측 누설인덕턴스와 2차측 누설인덕턴스 및 2차측 자화인덕턴스의 관계로 등가누설인덕턴스를 얻을 수 있다. 하지만 이러한 등가누설인덕턴스

는 2차측 자화인덕턴스가 권선비를 고려한 1차측 누설인덕턴스보다 매우 크므로 (

), 해석의 용이성을 위해 식 (2-2) 및 식 (2-4)와 같이 간략화된 등가 누설인덕턴스로 나타내었다. In the equivalent circuit of FIG. 13 and FIG. 15 Of formula (2-1) and formula (2-3)

Is the ratio of winding ratio on the secondary side of transformer (TR)

Equivalent leakage inductance can be obtained in relation to primary leakage inductance, secondary leakage inductance and secondary magnetization inductance. But these equivalent leakage inductance

Since the secondary magnetization inductance is much larger than the primary leakage inductance considering the turns ratio,

For simplicity, the simplified equivalent leakage inductance is shown in equations (2-2) and (2-4).

(1)

(One)

(2-1)

(2-1)

(2-2)

(2-2)

(2-3)

(2-3)

(2-4)

(2-4)

과 개방일 때의 주파수, 즉 코너 주파수(corner frequency)

를 식(3)과 식(4)에 정의하였다. In the present invention, in the equivalent circuit of Figs. 13 and 15, which is equivalent to Figs. 6 and 14, respectively, the frequency when the equivalent AC load resistance Req is a short circuit, that is, the resonance frequency

And frequency when open, ie corner frequency

Is defined in equations (3) and (4).

(3)

(3)

(4)

(4)

은 공진 주파수

과 스위칭 주파수

의 비이고, A는 병렬인덕턴스

와 등가누설인덕턴스

의 비이다. 그리고 Q는 Quality Factor이다. Normalized Resonance Frequency

Silver resonant frequency

Switching frequency

Is the ratio of and A is the parallel inductance

Equivalent Leakage Inductance

Is rain. And Q is the quality factor.

(5)

(5)

(6)

(6)

(7)

(7)

The voltage gain (M) characteristics for the input and output obtained based on the impedance relationship and the above equations (1) to (7) were obtained by equation (8).

(8)

(8)

(식(8))으로부터 알 수 있듯이 전압이득은 등가누설인덕턴스인 직렬공진인덕턴스

와 2차측 병렬인덕턴스

의 비(A) 및 부하 Quality factor(Q)와 규준화된 주파수

값에 의존한다. 이러한 경우 규준화된 공진 주파수

및 Q 변화에 대해 전압이득(M) 변화를 알아보기 위해 식(8)을 이용하여 시뮬레이션한 결과를 도 16에 나타내었다. Voltage gain characteristic

As can be seen from equation (8), the voltage gain is the series resonance inductance, which is equivalent leakage inductance.

And secondary parallel inductance

Ratio (A) and load quality factor (Q) and normalized frequency

Depends on the value Normalized resonant frequency in this case

And the results of the simulation using the equation (8) to find the change in the voltage gain (M) against the Q change is shown in FIG.

와 변압기 2차측 병렬인덕턴스

의 비인 A가 0.3일 때 규준화된 주파수

과 Q 변화에 대한 전압이득 특성 곡선이다. 등가누 설인덕턴스와 병렬인덕턴스 비(A)가 클수록 즉 병렬 인덕턴스가 작아질수록 전압이득이 높아지며 스위칭 동작을 하기 위한 규준화된 주파수의 스위칭 범위가 협소해진다. 이러한 협소한 스위칭 주파수 범위와 높은 전압이득 특성은 병렬인덕턴스 값이 감소하여야 하므로, 병렬인덕터에 흐르는 전류를 증가시킬 뿐만 아니라 도통 손실을 증가시키는 원인이 된다. 따라서 중부하에서도 주어진 입력 전압변동 범위에 대한 1차측 스위칭소자가 영전압 스위칭을 이룰 수 있는 병렬 인덕턴스

값과 병렬인덕터 전류를 구하기 위해 A와 Q 값을 적절하게 사용해야 한다. Figure 16 shows equivalent leakage inductance

And transformer secondary parallel inductance

Normalized frequency when A = 0.3

Voltage gain characteristic curves for and Q changes. The larger the equivalent leakage inductance and the parallel inductance ratio (A), that is, the smaller the parallel inductance, the higher the voltage gain and the narrower the switching range of the normalized frequency for switching operation. Such a narrow switching frequency range and high voltage gain characteristics cause the parallel inductance value to be reduced, which not only increases the current flowing through the parallel inductor, but also increases the conduction loss. Therefore, even in heavy loads, the parallel inductance of the primary switching element can achieve zero voltage switching for a given input voltage range.

The A and Q values should be used appropriately to find the value and parallel inductor current.

이 1보다 낮은 주파수에서 사용하고 있으며, 이러한 경우 주스위칭소자의 영전압 스위칭(ZVS) 뿐만 아니라 변압기 2차측 정류 다이오드의 영전류 스위칭(ZCS: Zero Current Switching)을 얻을 수 있고, 입출력전압 이득특성도 1보다 큰 승압모드에서 동작된다.In addition, the voltage booster type DC converter applying the transformer secondary side LLC series resonant converter proposed in the present invention has a frequency in which the switching operation region is normalized among ZVS (Zero Voltage Switching) regions as shown in FIG. 16.

It is used at a frequency lower than 1, and in this case, not only the zero voltage switching (ZVS) of the juice switching element but also the zero current switching (ZCS) of the secondary rectifier diode of the transformer can be obtained, and the input / output voltage gain characteristics are also obtained. Operates in boost mode greater than one.

이 1보다 높은 주파수 영역에서는 입출력전압이득특성이 1보다 적은 강압모드로 동작되고 주회로 스위치는 ZVS에서 동작된다. Also, normalized frequency

In the frequency region higher than 1, the input / output voltage gain characteristic is operated in the step-down mode with less than 1, and the main circuit switch is operated in ZVS.

Actual application example with specific specification

Input voltage 22 to 30 VDC Output voltage 400 VDC Output current 2.5 A Juice switching frequency 70 kHz to 100 kHz Series resonance frequency 100 kHz Juice Switching Devices (Connect 2 in parallel) IRF 3077PbF (75V, 210A, R _DS = 3.3mΩ)

)Output rectifier diode

) DESI 12-06A (600V, 14A, V _F = 1.5V)

Table 1 is a specific application example of the step-up DC converter applying LLC series resonant converter according to the present invention, Figure 17 is a primary terminal voltage (V _ab ) and the current of the step-up DC converter shown in Figure 14, an embodiment of the present invention Experimental waveform diagram of (I _T1 ) and secondary resonant current (I _D ) and parallel inductor voltage (V _L2 ) (50V / div., 50A / div., 500V / div., 2.5A / div., 4us / div ). In this application, 1kW class is designed to boost low input voltage (22 ~ 30VDC) to high output voltage (400VDC). As shown in the parameters shown in Table 1, the experimental conditions and the applied device for the converter device according to the present invention were applied.

17A, 17B, and 17C are experimental waveforms of a transformer primary terminal voltage and current, a secondary load current, and parallel inductor voltages for 300 W, 600 W, and 1 kW for a constant output voltage. As shown in Figs. 17A to 17C, in the converter device of the present invention, the terminal current from the heavy load (1kW) to the light load 300W is the ground current (Lagging Current) with respect to the terminal voltage (V _ab ). It can be seen that the flow as, and the zero voltage switching of the juice switching element was confirmed for all load changes. In addition, the experimental results from (a) to (c) show that the frequency change (82.39 kHz to 88.19 kHz) of the waveform operates within the same switching frequency range as the simulation switching frequency range (100 kHz to 70 kHz) of FIG. 16. Also, because the transformer primary side resonant current flows discontinuously, the reverse recovery loss of the transformer secondary side rectifier diode can be reduced. Fig. 18 shows a prototype photograph of the boost type DC converter of the present invention designed for 1 kW class.

Another embodiment

와 출력부(정류다이오드단) 사이에 직렬인덕터

를 둠으로서 더 높은 DC 이득특성 및 더 좁은 주파수 제어범위에서 출력전압을 제어할 수 있는 특징을 갖도록 한 응용 실시예를 나타낸다. 도 19의 주회로에 대한 도 21의 등가회로를 이용하여 전압이득(M) 특성식 (

)을 식(9)와 같이 구하였다. 19 shows a parallel inductor as another embodiment of the present invention.

Series inductor between the output and rectifier diode stage

An application embodiment is shown to have a feature of controlling the output voltage at a higher DC gain characteristic and a narrower frequency control range. Voltage gain (M) characteristic equation using the equivalent circuit of FIG. 21 for the main circuit of FIG.

) Was obtained as in Equation (9).

(9)

(9)

은 스위칭주파수

와 공진주파수

의 비로써 규준화된 주파수를 나타낸 것이며,

는 변압기의 병렬인덕터

와 식 (2-2)에 나타낸 변압기(TR) 1, 2차측 누설인덕턴스

의 비율을 나타낸 것이고,

는 병렬인덕터

와 직렬인덕터

의 비율을 나타낸 것이다. 그리고

는 부하 Quality factor를 나타낸 것이다. As you can see in equation (9)

Switching frequency

And resonant frequency

Represents the normalized frequency as the ratio of

Is the parallel inductor of the transformer

And transformer (TR) primary and secondary leakage inductance shown in Equation (2-2)

Represents the ratio of

Is a parallel inductor

With serial inductor

It shows the ratio of. And

Denotes the load quality factor.

Another embodiment

)를 삽입하는 것 대신에 갭(Gap)이 있는 변압기(LCTR: Loosely Coupled Transformer)를 적용하여 변압기 자화인덕턴스

및 누설인덕턴스

를 공진수단로 이용함으로 더 높은 DC이득특성을 얻을 수 있는 특징을 갖는 응용 실시예를 제안한다. 이 회로에 대한 전압이득(M) 특성식 (

)은 식(9)와 같다. 식 (9)에서 볼 수 있듯이

은 스위칭주파수

와 공진주파수

의 비로써 규준화된 주파수를 나타낸 것이며, 이때

는 변압기의 1차측 자화인덕턴스

과 1차측 누설인덕턴스

의 비율을 나타낸 것이고,

는 1차측 자화인덕턴스

과 1차측에서 권선비를 고려하여 바라본 2차측 누설인덕턴스

의 비율을 나타낸 것이다. 그리고

는 부하 Quality factor를 나타낸 것이다. In addition, the basic circuit proposed in the present invention and the circuit in FIG. 19 are neglected by minimizing the primary and secondary leakage inductances of the transformer TR as in FIG.

Transformer magnetization inductance by applying a loosely coupled transformer (LCTR) instead of inserting

And leakage inductance

Proposes an application embodiment having a feature that can obtain a higher DC gain characteristics by using a resonator means. Voltage gain (M) characteristic equation for this circuit (

) Is the same as (9). As you can see in equation (9)

Switching frequency

And resonant frequency

Which represents the normalized frequency as the ratio of

Is the magnetizing inductance of the primary side of the transformer

And primary leakage inductance

Represents the ratio of

Is the primary magnetization inductance

Leakage inductance on the secondary side with consideration of turns ratio

It shows the ratio of. And

Denotes the load quality factor.

일 때 도 19의 인덕터

또는 도 20의 변압기(LCTR)의 2차측 누설인덕턴스

를 고려할 경우 각각 식(10)과 식(11)로 나타낼 수 있다.From the voltage gain characteristic of equation (9)

Inductor of FIG. 19 when

Or secondary leakage inductance of transformer LCTR of FIG.

In consideration of this, it can be represented by equation (10) and equation (11), respectively.

(10)

10

(11)

(11)

또는 도 20의 변압기(LCTR)의 2차측 누설인덕턴스

에 따라 DC 전압이득이 높아지는 것을 확인할 수 있다. 또한 식 (9)의 전압이득특성식을 이용한 도 20의 시뮬레이션 결과는 A=0.1, B=0.2인 경우 규준화된 주파수와 Q 변화에 대한 전압이득 특성곡선을 나타내고 있다(도 22 참조). 이러한 특성곡선에서 볼 수 있듯이 규준화된 주파수(

)가 1인 곳에서 높은 전압이득 특성을 갖게 되어 변압기(TR) 2차측의 적은 턴수에도 높은 출력 전압을 얻을 수 있으며 보다 좁은 주파수 제어범위에 따라 무부하 및 경부하시 일정출력전압 제어를 위한 주파수 제어가 용이하며 과부하시에도 주스위칭소자의 영전압 스위칭을 이룰 수 있는 특성을 가지고 있다. The inductor of FIG. 19 from the voltage gain characteristic formulas of Equations (10) and (11)

Or secondary leakage inductance of transformer LCTR of FIG.

As a result, the DC voltage gain increases. In addition, the simulation result of FIG. 20 using the voltage gain characteristic equation of Equation (9) shows the voltage gain characteristic curve for the normalized frequency and Q change when A = 0.1 and B = 0.2 (see FIG. 22). As you can see from this characteristic curve,

) Has a high voltage gain characteristic at 1), so a high output voltage can be obtained even at a small number of turns on the secondary side of the transformer (TR), and frequency control for constant output voltage control at no load and light load is possible according to a narrower frequency control range. It is easy and has a characteristic that can achieve zero voltage switching of juice switching element even under overload.

In all the circuits disclosed above, there may be a blocking capacitor on the primary side to prevent DC biasing in series with the transformer primary terminal.

In the above, a specific embodiment of the DC converter using the SLLC series resonant converter of the present invention has been described. However, the technical scope of the present invention is not limited only to the above-described embodiment. For example, the circuit proposed in the present invention may be applied to a push-pull converter and a half bridge converter as shown in FIGS. 23 and 24. As shown in Fig. 25, the converter according to the present invention can be applied to an AC load such as a lamp load which has a high voltage gain characteristic under no load and requires a low gain characteristic under load. In addition, the converter circuit proposed in the present invention can be applied to a step-up DC converter as well as a step-up DC converter. The technical scope of the present invention is to be determined by reasonable interpretation of the appended claims.

The DC converter according to the present invention can operate in both the boost mode and the step-down mode, can control a wide range of input / output voltage in a relatively narrow frequency control range, and softly switch all the switching elements in the switching operation region. DC-DC converter adopting SLLC series resonant converter with several advantages.

)에 저장된 에너지에 의한 전류에 의해 결정되므로 부하에 관계없이 영전압 스위칭을 이룰 수 있다. 또한 변압기 2차측 공진전류는 불연속적으로 흐르기 때문에 출력다이오드(

,

,

,

) 역회복 특성에 따른 스위칭 손실을 저감할 수 있다. In the present invention, as shown in Figure 6 by placing the resonating means of the series resonator converter on the transformer secondary side to configure the main circuit can reduce the size and the cost of the capacitor (C _S ) for the series resonance. In addition, zero voltage switching at the time of turning on / off the juice switching element is a parallel inductor.

The zero voltage switching can be achieved regardless of the load because it is determined by the current by the energy stored in the circuit. Also, because the transformer secondary side resonant current flows discontinuously,

,

,

,

) The switching loss due to the reverse recovery characteristic can be reduced.

Claims (10)

In a series resonant converter circuit comprising a plurality of juice switching elements and a transformer having a primary winding and a secondary winding to generate an induced voltage by the current interruption of the juice switching element, Resonating means connected in series with the secondary winding of the transformer, And said transformer does not have a gap. 2. The series resonant converter circuit according to claim 1, wherein the resonator means comprises a capacitor having a first end connected to one end of a secondary side winding of the transformer and a second end connected to an output side. 3. The series resonant converter circuit according to claim 2, wherein said resonator means comprises a primary leakage inductance and a secondary leakage inductance of said transformer. 3. The series resonant converter circuit of claim 2, wherein the resonator means further comprises a parallel inductor connected to the second end of the capacitor. 3. The series resonant converter circuit according to claim 2, wherein said resonating means further comprises a series inductor connected between a second end of said capacitor and a connection with said parallel inductor and an output end. In a series resonant converter circuit comprising a plurality of juice switching elements and a transformer having a primary winding and a secondary winding to generate an induced voltage by the current interruption of the juice switching element, Resonator means connected in series to the secondary winding of the transformer, And a loosely coupled transformer (LCTR) transformer having a gap in series with the resonance means. 7. The series resonant converter circuit according to claim 6, wherein the resonator means comprises a capacitor having a first end connected to one end of a secondary winding of the transformer and a second end connected to an output side. 8. The magnetic resonance inductance of claim 7, wherein the resonator means

) And leakage inductance (

Serial resonance converter circuit comprising a). A DC conversion device employing the series resonant converter circuit according to any one of claims 1 to 8 as a main circuit, and converts a direct current of the first voltage input to the input terminal into a direct current of the second voltage and outputs it to the output terminal. 10. The DC converter as set forth in claim 9, further comprising a blocking capacitor connected to a primary side winding of a transformer of the converter circuit to prevent DC biasing due to a voltage input to the input terminal.

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