KR100510435B1 - Over current protection circuit - Google Patents

Abstract

The present invention relates to an overcurrent protection circuit, which is connected between a current sensing terminal for sensing a current flowing in a secondary side of a power supply and an output voltage terminal for outputting an output voltage, and between the current sensing terminal and the output voltage terminal. An overcurrent protection circuit for a power supply having a choke coil having a very small equivalent resistance for the function of an LC filter of the power supply, the overcurrent protection circuit comprising: a reference voltage from an external source; A temperature variable offset generator including an input signal generator for outputting a first voltage having a temperature coefficient equal to a coefficient, a voltage-current converter for inputting the first voltage and converting the first voltage into a current signal, and the output voltage; A terminal is connected to the inverting input terminal and the current sensing terminal Connected to a non-inverting input terminal and an output of the temperature variable offset generator is applied between the output voltage terminal and the inverting input terminal and overcurrent protection according to a signal input to the inverting input terminal and the non-inverting input terminal. By providing a comparator for generating and outputting a signal, it has the advantage of reducing unnecessary power loss due to voltage drop across the equivalent resistance of the choke coil, which operates stably at temperature changes and acts as a current sensing resistor.

Description

Over current protection circuit

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an overcurrent protection circuit of a power supply, and in particular, by setting a simple and accurate offset voltage, an overcurrent of a power supply configured to reduce costs and perform a stable function under conditions such as process dispersion, temperature change, and the like. It relates to a protection circuit.

SMPS, a power supply, has a number of built-in protection circuits such as over current protection (OCP) and over voltage protection (OVP) to ensure reliable operation of the system. For example, in SMPS which is a power supply device of a personal computer (PC), various outputs, such as + 3.3V, + 5V, + 12V, are designed on the secondary side of SMPS. Since performing individual functions such as over current protection (OCP) and over voltage protection (OVP) for each output can give more accurate results than protection in the overall concept, It is a common trend to perform separate over current protection (OCP) and over voltage protection (OVP) for typical + 3.3V / + 5V / + 12V outputs.

1 shows a circuit diagram of an overcurrent protection circuit of a conventional power supply device together with a secondary side of the power supply device.

Referring to FIG. 1, an overcurrent protection circuit of a conventional power supply includes resistance elements 112 and 114 and a comparator 116. Reference numeral 100 denotes a secondary side of the power supply device, and reference numeral 110 denotes an overcurrent protection circuit. The terminal 101 represents a terminal for sensing the current IS flowing to the secondary side of the power supply, and the terminal 102 represents a terminal for outputting the output voltage Vo of the power supply. Reference numeral 108 denotes a choke coil connected between the terminals 101 and 102 and having a very small equivalent resistance Rs for the function of the LC filter of the power supply.

One end of the resistance element 112 is connected to the terminal 101.

The resistance element 114 is connected between the other terminal of the resistance element 112 and the ground terminal GND.

The comparator 116 inputs a signal applied to the other terminal of the resistor element 112 to the non-inverting input terminal, inputs an output voltage Vo output from the terminal 102 to the inverting input terminal, and compares the signal. The signal is output as an overcurrent signal OUT.

The operation of the overcurrent protection circuit of the conventional power supply apparatus will be described with reference to FIG. 1.

Terminal 101 is a current sensing terminal, and terminal 102 is an output voltage Vo terminal. Between the terminals 101 and 102 is connected a choke coil 108 with a very small equivalent resistance Rs for the function of an LC (inductor-capacitor) filter. Here, the equivalent resistance Rs of the choke coil 108 serves as a current sensing resistor of the secondary side 100 of the current supply device. When an excessive amount of current flows in the output terminal of the secondary side 100, that is, the terminal 102, the voltage difference between both ends of the current sensing resistor exceeds a set value. At this time, the comparator 116 detects this and outputs the overcurrent signal OUT to a high ('H') level, whereby the main power is turned off by the signal. The overcurrent protection circuit protects the system.

Here, the terminals 101 and 102 of the secondary side 100 of the power supply device should have a potential of basically the same value. For this purpose, the smaller the value of the equivalent resistance Rs of the choke coil 108 is, the better. In addition, the voltage difference across the equivalent resistance (Rs) should be designed as small as possible. In a conventional circuit, a comparator 116 connects the resistors 112 and 114 to the terminal 101, and uses the voltage divided by the resistance ratio as one input and the output voltage Vo of the power supply as the other input. The overcurrent protection function was performed by the structure of. In other words, it can be said that the voltage divided by the resistance ratio has the same effect as the offset voltage of the comparator 116.

However, the biggest problem of the overcurrent protection circuit of the conventional voltage supply device is the resistance elements 112 and 114. As mentioned above, the voltage across the equivalent resistor (Rs), which acts as a current sensing resistor, should be about the same, and the closer the offset voltage is to near zero, the better. Therefore, the resistance ratios of the resistors 112 and 114 have a very large value such as 1:50 or 1: 100. However, in actual integrated circuit design, it is difficult to accurately match such a resistance value or a resistance ratio, and a large number of cells must be used to match this through matching of resistance values. Therefore, the burden due to the increase in chip size and the increase in costs is inevitable. In addition, in the overcurrent protection circuit of the conventional power supply apparatus, when the output voltage Vo changes, the sensing current sensed through the terminal 101 also changes accordingly.

Accordingly, an object of the present invention is an overcurrent protection circuit which is configured to simply and accurately set an offset voltage in an overcurrent protection circuit of a power supply, and to perform a stable function with respect to power supply voltage, temperature, process dispersion, etc. in integrated circuit design. To provide.

In order to achieve the above object the overcurrent protection circuit according to the present invention

A current sensing terminal for sensing a current flowing to the secondary side of the power supply, an output voltage terminal for outputting an output voltage, and connected between the current sensing terminal and the output voltage terminal and connected to the LC filter of the power supply. An overcurrent protection circuit for a power supply having a choke coil having a very small equivalent resistance for its function, the overcurrent protection circuit comprising: a first input having a temperature coefficient equal to a temperature coefficient of an equivalent resistance of the choke coil by inputting a reference voltage from an external source; A temperature variable offset generator including an input signal generator for outputting a voltage and a voltage-current converter for inputting the first voltage and converting the first voltage into a current signal; And the output voltage terminal is connected to an inverting input terminal, the current sensing terminal is connected to a non-inverting input terminal, and an output of the temperature variable offset generator is applied between the output voltage terminal and the inverting input terminal. And a comparator for generating and outputting an overcurrent protection signal according to a signal input to the inverting input terminal and the non-inverting input terminal.

Next, a preferred embodiment of the present invention will be described in detail with reference to the accompanying drawings.

2 shows an overcurrent protection circuit of a power supply according to an embodiment of the present invention together with the secondary side of the power supply.

2, the overcurrent protection circuit of the power supply apparatus according to the embodiment of the present invention includes a temperature variable offset generator 222 and a comparator 224. Reference numeral 210 denotes a secondary side of the power supply device, and reference numeral 220 denotes an overcurrent protection circuit according to an embodiment of the present invention. Terminal 201 represents a current sensing terminal for sensing current flowing to the secondary side of the power supply, and terminal 202 represents an output voltage terminal for outputting the output voltage Vo of the power supply. Reference numeral 208 denotes a choke coil connected between the terminals 201 and 202 and having a very small equivalent resistance Rs for the function of the LC filter of the power supply.

The variable temperature offset generator 222 has a temperature coefficient equal to the equivalent resistance Rs of the choke coil 208 and generates an offset voltage Voffset that is stable to conditions such as process dispersion and varies with temperature.

The comparator 224 has a terminal 202 of the secondary side 210 of the power supply connected to an inverting input terminal, and a terminal 201 of the secondary side 210 of the power supply connected to a non-inverting input terminal. An offset voltage Voffset, which is connected and is generated from the temperature variable offset generator 222, is applied between the terminal 202 and the inverting input terminal, and is applied to a signal input to the inverting input terminal and the non-inverting input terminal. Therefore, the overcurrent protection signal OUT is generated and output.

3 is a circuit diagram of a circuit according to a specific embodiment of the temperature variable offset generator 222 and the comparator 224 in FIG. 2.

Referring to FIG. 3, in FIG. 2, the temperature variable offset generator 222 includes an input signal generator 310 and a voltage-current converter 320.

The input signal generator 310 inputs the reference voltage VREF from the outside to output the voltage VIN having the same temperature coefficient as that of the equivalent resistance Rs of the choke coil 208.

The input signal generator 310 is composed of resistance elements 311, 312, 313, a transistor 314, and a diode 315.

One end of the resistor 311 is connected to a terminal for inputting the reference voltage VREF.

The resistance element 312 is connected between the other terminal of the resistance element 311 and the ground terminal GND.

The transistor 314 is an N-type bipolar transistor in which a collector terminal is connected to a terminal for inputting a reference voltage VREF and gated by another terminal of the resistance element 311.

In the diode 315, an anode terminal is connected to an emitter terminal of the transistor 314, and a cathode terminal is connected to an output terminal 316 for outputting an input voltage VIN.

The resistance element 313 is connected between the output terminal 316 and the ground terminal GND.

The voltage-current converter 320 inputs the voltage VIN output from the input signal generator 310 and converts the voltage VIN into a current signal having a corresponding value.

The voltage-current converter 320 includes current mirror circuits 322 and 324, a resistor 326, an amplifier 328, and a transistor 330.

The current mirror 322 is composed of transistors Q1 and Q2.

The transistor Q1 is a P-type bipolar transistor whose emitter terminal is connected to the power supply terminal VCC and gated by the collector terminal.

Transistor Q2 is a P-type bipolar transistor whose emitter terminal is connected to power supply terminal VCC and gated by the collector terminal of transistor Q1.

The current mirror circuit 322 supplies the current of the collector terminal of the transistor Q2 based on the current output from the collector terminal of the transistor Q1.

The current mirror circuit 324 is composed of transistors Q3 and Q4.

The transistor Q3 is a P-type bipolar transistor whose emitter terminal is connected to the collector terminal of the transistor Q2 and is gated by the collector complex.

The transistor Q4 is a P-type bipolar transistor whose emitter terminal is connected to the collector terminal of the transistor Q1 and is gated by the collector terminal of the transistor Q3.

The current mirror circuit 324 is controlled by the current supplied from the current mirror circuit 322 to supply the current of the collector terminal of the transistor Q4 based on the current output from the collector terminal of the transistor Q3.

The resistor 326 has one terminal connected to the ground terminal GND.

The amplifier 328 has a non-inverting input terminal connected to the output terminal 316 of the input signal generator 310, and an inverting input terminal connected to the other terminal of the resistance element 326.

The transistor 330 is an N type in which a collector terminal is connected to the collector terminal of the transistor Q3, and an emitter terminal is connected to the other terminal of the resistor 326, and is gated by the output terminal of the amplifier 328. Of bipolar transistors.

Referring to FIG. 3, in FIG. 2, the comparator 224 is configured as a comparator circuit 340 and a driver 360.

The comparison circuit is composed of transistors 342 to 350, current mirror circuit 352, resistance element 354, and current source 356.

One terminal of the current source 356 is connected to the power supply terminal VCC.

The current mirror circuit 352 is composed of transistors Q5 and Q6.

The transistor Q5 is an N-type bipolar transistor connected to the ground terminal GND by the emitter terminal and gated by the collector terminal of the transistor Q6.

Transistor Q6 is an N-type bipolar transistor connected to the ground terminal GND with an emitter terminal and gated by a collector terminal.

The current mirror circuit 352 supplies the current of the collector terminal of the transistor Q5 with reference to the current input to the collector terminal of the transistor Q6.

The transistor 342 has an emitter terminal connected to the collector terminal of the transistor Q1 of the voltage-current converter 320, and is gated by the collector terminal of the transistor Q3 of the voltage-current converter 320. P-type bipolar transistor.

The transistor 344 has an emitter terminal connected to the other terminal of the current source 356, a collector terminal connected to the collector terminal of the transistor Q5 of the current mirror circuit 352, and a voltage-current converter 320. It is a P-type bipolar transistor gated by the collector terminal of transistor Q4.

The transistor 346 has an emitter terminal connected to the other terminal of the current source 356, a collector terminal connected to the collector terminal of the transistor Q6 of the current mirror circuit 352, and a collector terminal of the transistor 342. It is a P-type bipolar transistor gated.

One terminal of the resistor 354 is connected to the collector terminal of the transistor 342.

The transistor 348 has an emitter terminal connected to the collector terminal of the transistor Q4 of the voltage-current converter 320, a collector terminal connected to the ground terminal GND, and the power supply secondary side 210. It is a P-type bipolar transistor gated by the terminal 201 of.

Transistor 350 has an emitter terminal connected to another terminal of resistor element 354, a collector terminal connected to ground terminal GND, and is gated by terminal 202 of power supply secondary side 210. P-type bipolar transistor.

The driver 360 inputs a signal output from the comparison circuit 340, drives the same, and outputs the signal as an overcurrent protection signal OUT.

The driver 360 is composed of a resistance element 362 and a transistor 364.

The resistance element 362 is connected between the power supply terminal VCC and the output terminal 363 for outputting the overcurrent protection signal OUT.

The transistor 364 has a collector terminal connected to an output terminal 363 for outputting an overcurrent protection signal OUT, an emitter terminal connected to a ground terminal GND, and constitutes a comparison circuit 340. It is an N-type bipolar transistor gated by the collector terminal of the transistor 344.

The operation of the overcurrent protection circuit according to the embodiment of the present invention will be described with reference to FIGS. 2 and 3 as follows.

Twice the base-emitter voltage Vbe, i.e., 2 Vbe, to match the temperature coefficient of the equivalent resistance Rs of the choke coil 208 at a stable reference voltage VREF under conditions such as supply voltage, process dispersion, and temperature. The voltage VIN dropped by the amount becomes the input of the voltage-current converter 320. The purpose of using the voltage dropped through 2 Vbe is to equally match the temperature coefficient of the equivalent resistance Rs of the choke coil 208 in this section. The reference voltage VREF is stable to temperature and divided by the resistor element 311 and the resistor 312 so that the voltage VREF1 dropped by the resistor element 311 from the reference voltage VREF is always at a constant resistance ratio. As always, the same value is maintained for temperature changes. Accordingly, the voltage VIN is a value dropped by 2Vbe using the N-type bipolar transistor 314 and the diode 315 to the voltage VREF1, VREF1-2Vbe. Since the base-emitter voltage Vbe has a temperature coefficient of about -2 mV / ° C, the voltage VIN, on the contrary, has a temperature coefficient of +4 mV / ° C. The exact temperature coefficient of the transistor 314 can be obtained through simulation considering the temperature parameter. Therefore, after extracting the temperature coefficient of the equivalent resistance Rs of the choke coil 208 used in the system, the temperature coefficient of the voltage VIN may be adjusted to this.

The temperature change rate Ratio (VIN) of the voltage VIN may be expressed by Equation 1 below. Here, R1 and R2 represent resistance values of the resistance element 311 and the resistance element 312, respectively.

The amplifier 328 constituting the voltage-current converter 320 serves as a buffer having a gain of 1. The input of the amplifier 328 is composed of a voltage VIN. Since the input and output of the buffer are always the same, the voltage VX of the upper node of the resistor element 326 is always equal to the voltage VIN and has the same temperature coefficient Ratio (VX) as the temperature coefficient Ratio (VIN). . The reference current IREF flowing through the resistance element 326 by the voltage VX is determined as in Equation 2 below. Here, RX represents the resistance value of the resistance element 326.

The setting of the voltage VX is most important in order to make the temperature coefficient of the equivalent resistance Rs of the choke coil 208 equal to the temperature coefficient of the voltage VIN. If the temperature coefficient of the equivalent resistance Rs of the choke coil 208 is, for example, 0.5% / ° C, then the temperature coefficient of the voltage VX having the same value as the voltage VIN is equal to, for example, 0.5% / ° C. do. As can be seen from Equation 2, the voltage VIN dropped by 2 Vbe from the voltage VREF1 is input to the voltage-current converter 320, and the temperature coefficient of the equivalent resistance Rs of the choke coil 208 is At 0.5% / ° C, the value of the voltage VX in consideration of the temperature coefficient is 0.8V, as shown in Equation 3 below.

By adjusting the resistance ratios of the resistance elements 311 and 312 to determine the value of the voltage VX, the value of the voltage VX can be adjusted to 0.8 V, and the temperature coefficient of the equivalent resistance Rs of the choke coil 208 is determined. Even if is different, the value of the voltage (VX) can be obtained by using the relationship shown in equation (3).

The current mirror circuit 322 constituting the voltage-current converter 320 is added to reduce the early effect of the P-type bipolar transistor to minimize current distortion caused by the current mirror circuit 324. to be. The transistors Q1 to Q4 are arranged so that the diodes are staggered from each other so that the base currents are compensated for each other.

Since the current I1 flowing through the collector terminal of the transistor Q4 and the current I2 flowing through the transistor 342 are the same as the reference current IREF flowing through the resistance element 326, the transistor 344 and the transistor 346. The base-emitter voltage Vbe (on) at the turn-on state of Δ is the same. At this time, the voltage difference across the resistor element 354 connected between the base terminal of the transistor 346, which is the inverting input terminal of the comparison circuit 340, and the emitter terminal of the transistor 350, has an offset voltage Voffset. do. As shown in Equation 4 below, the offset voltage Voffset is represented by the product of the voltage VIN and the resistance ratio of the resistance elements 326 and 354. Therefore, the offset voltage Voffset is stable to the power supply voltage, process dispersion, and the like and changes only in accordance with the temperature coefficient of the voltage VIN. Therefore, it changes in the same manner as the temperature coefficient of the equivalent resistance Rs of the choke coil 208, and stable offset voltage Voffset can be obtained under other conditions. Here, Ros represents a resistance value of the resistance element 354.

The inverting and non-inverting inputs of the comparing circuit 340 are the voltages drawn from the choke coil 208 both terminals 201, 202 of the secondary side 210 of the power supply. The voltage Vo of the terminal 202 is basically always smaller than the voltage of the terminal 201. However, the offset voltage Voffset is applied to the inverting input terminal of the comparison circuit 340 to which the voltage Vo of the terminal 202 is artificially input. Therefore, when excessive current flows in the equivalent resistance Rs of the choke coil 208 of the secondary side 210 of the power supply device and the potential difference between the both ends of the equivalent resistance Rs becomes larger than the offset voltage Voffset, the driving unit ( The overcurrent protection signal OUT, which is the output of 360, is at a high ('H') level. This signal causes the main power to be turned off, so that the overcurrent protection circuit protects the system.

4 illustrates the results of the operation simulation of FIG. 3. Here, for example, the value of the voltage Vo is 5V, and the offset voltage Voffset is set to 50mV. If the conventional circuit is used to satisfy this condition, a resistance ratio of 1: 1000 is required. This resistance ratio is very difficult to accurately match inside the chip, and considering the temperature coefficient of the equivalent resistance of the choke coil, it is difficult to meet the 50% specification margin.

As shown in Fig. 4, in the simulation verification, the simulation was conducted under conditions corresponding to the bad conditions that are higher than the conditions allowed by the process rather than the normal conditions. The temperature coefficient of the choke coil equivalent resistance is 0.5% / ° C, the variation rate of resistance (ΔRs) is -20%, and four conditions of various temperatures, -25 ° C, 25 ° C, 75 ° C, and 125 ° C. Outputting the results 402, 404, 406, and 408, the offset voltages (Voffsets) of 23.7, 50, 74, and 98.6 mV were obtained, respectively. That is, the offset voltage (Voffset) was able to obtain a result of 50mV at room temperature (25 ℃), the temperature change rate of the offset voltage (Voffset) was 0.49% / ℃ almost the same as the temperature coefficient of the equivalent resistance.

5 is a circuit diagram of a circuit according to another specific embodiment of the temperature variable offset generator 222 in FIG. 2.

Referring to FIG. 5, in FIG. 2, a circuit according to another exemplary embodiment of the temperature variable offset generator 222 includes an input signal generator 510 and a voltage-current converter 520.

The input signal generator 510 inputs a reference voltage VREF from the outside to output a voltage VIN having the same temperature coefficient as that of the equivalent resistance Rs of the choke coil 208.

The input signal generator 310 includes resistance elements 511, 512, 513, transistors 514, and diodes D1 to Dn-1.

One end of the resistance element 511 is connected to a terminal for inputting the reference voltage VREF.

The resistance element 512 is connected between the other terminal of the resistance element 511 and the ground terminal GND.

The transistor 514 is an N-type bipolar transistor in which a collector terminal is connected to a terminal for inputting a reference voltage VREF and gated by another terminal of the resistance element 511.

The diodes D1-Dn-1 are connected in series with each other between the emitter terminal of the transistor 514 and the output terminal 516 for outputting the input voltage VIN.

The resistance element 513 is connected between the output terminal 516 and the ground terminal GND.

Since the voltage-current converter 520 may be configured in the same manner as the voltage-current converter 320 of FIG. 3, a detailed description thereof will be omitted.

As shown in Figs. 2 and 5, the base-emitter voltage is set to match the temperature coefficient of the equivalent resistance Rs of the choke coil 208 at a stable reference voltage VREF under conditions such as power supply voltage, process dispersion, and temperature. An input voltage of the voltage-to-current converter 320 is n times Vbe, that is, the voltage VIN dropped by nVbe. The purpose of using the voltage dropped through nVbe is to equally match the temperature coefficient of the equivalent resistance Rs of the choke coil 208 in this section. The reference voltage VREF is stable to temperature and is divided by the resistor 511 and the resistor 512 so that the voltage VREF2 dropped by the resistor 511 from the reference voltage VREF is always at a constant resistance ratio. As always, the same value is maintained for temperature changes. Therefore, the voltage VIN is a value dropped by nVbe using the N-type bipolar transistor 514 and the diodes D1 to Dn-1 to the voltage VREF2, and VREF1 to nVbe. Since the base-emitter voltage Vbe has a temperature coefficient of about -2 mV / ° C, the voltage VIN, on the contrary, has a temperature coefficient of +2 nmV / ° C. The exact temperature coefficient of the transistor 514 can be obtained through simulation considering the temperature parameter. Therefore, after extracting the temperature coefficient of the equivalent resistance Rs of the choke coil 208 used in the system, the temperature coefficient of the voltage VIN may be adjusted to this.

The temperature change rate Ratio (VIN) of the voltage VIN may be expressed by Equation 5 below. Here, R1 and R2 represent resistance values of the resistance element 511 and the resistance element 512, respectively.

The amplifier 528 constituting the voltage-current converter 520 serves as a buffer having a gain of 1. The input of the amplifier 528 is comprised of the voltage VIN. Since the input and output of the buffer are always the same, the voltage VX at the top node of the resistor 526 is always equal to the voltage VIN and has the same temperature coefficient Ratio (VX) as the temperature coefficient Ratio (VIN). . The reference current IREF flowing through the resistance element 526 by the voltage VX is determined as shown in Equation 6 below. Here, RX represents the resistance value of the resistance element 526.

The setting of the voltage VX is most important in order to make the temperature coefficient of the equivalent resistance Rs of the choke coil 208 equal to the temperature coefficient of the voltage VIN. If the temperature coefficient of the equivalent resistance Rs of the choke coil 208 is, for example, 0.5% / ° C, then the temperature coefficient of the voltage VX having the same value as the voltage VIN is equal to, for example, 0.5% / ° C. do. As can be seen from Equation 6, the voltage VIN dropped by nVbe from the voltage VREF1 is input to the voltage-current converter 520, and the temperature coefficient of the equivalent resistance Rs of the choke coil 208 is At 0.5% / ° C, the value of the voltage VX in consideration of the temperature coefficient is 0.4 nV as shown in Equation 7 below. DELTA Rs denotes the temperature coefficient of the equivalent resistance Rs of the choke coil 208.

By adjusting the resistance ratio of the resistors 511 and 512 and the number of diodes D1 to Dn-1 to determine the value of the voltage VX, the value of the voltage VX can be adjusted to 0.4 nV, and the choke coil Even if the temperature coefficient of the equivalent resistance Rs of (208) is different, the value of the voltage VX can be obtained by using the relational expression shown in Equation (7).

Referring to the circuit of the comparator 224 illustrated in FIG. 3, the overcurrent protection circuit including the variable temperature offset generator 222 of FIG. 5 is a voltage obtained by subtracting the offset voltage Voffset as shown in equation (4). VIN) and the resistance ratio of the resistance elements 526 and 354. Therefore, the offset voltage Voffset is stable to the power supply voltage, process dispersion, and the like and changes only in accordance with the temperature coefficient of the voltage VIN. Therefore, it changes in the same manner as the temperature coefficient of the equivalent resistance Rs of the choke coil 208, and stable offset voltage Voffset can be obtained under other conditions.

As described above, the overcurrent protection circuit according to the embodiment of the present invention has a voltage having the same temperature coefficient as the equivalent resistance Rs in consideration of the temperature characteristic of the equivalent resistance Rs of the choke coil 208 unlike the conventional circuit. The offset voltage Voffset is obtained by using the and resistance ratios. Therefore, it has fundamentally eliminated the cause of error due to temperature, and has the advantage that it can sufficiently cope with the specifications of the system with strict standards. In addition, since the offset voltage Voffset can be set to a very small value, for example, several tens of mV, an unnecessary power loss due to the voltage drop across the equivalent resistance of the choke coil acting as a current sensing resistor has an advantage.

According to the present invention, in consideration of the temperature characteristic of the equivalent resistance of the choke coil, the offset voltage is obtained using a voltage and a resistance ratio having the same temperature coefficient as the equivalent resistance. Therefore, it has fundamentally eliminated the cause of error due to temperature and has the advantage that it can sufficiently cope with the specification of the system with strict standards. In addition, since the offset voltage can be set to a very small value, for example, several tens of mV, it has the effect of reducing unnecessary power loss due to the voltage drop across the equivalent resistance of the choke coil serving as the current sensing resistor.

1 is a block diagram of an overcurrent protection circuit of a conventional power supply.

2 is a block diagram of an overcurrent protection circuit of a power supply according to an embodiment of the present invention.

3 is a circuit diagram of a circuit according to a specific embodiment of an overcurrent protection circuit of a power supply according to an embodiment of the present invention.

4 is a simulation result of the operation of FIG. 3.

5 is a circuit diagram of a circuit according to a specific embodiment of an overcurrent protection circuit of a power supply according to another embodiment of the present invention.

* Detailed description of the signs in the drawings

VCC: power terminal, GND: ground terminal,

VREF: reference voltage, OUT: overcurrent protection signal,

VIN: input voltage, Q1 to Q6: transistors,

Rs: equivalent resistance, Vo: output voltage of the power supply,

Voffset: Offset voltage.

Claims (15)

A current sensing terminal for sensing a current flowing to the secondary side of the power supply, an output voltage terminal for outputting an output voltage, and connected between the current sensing terminal and the output voltage terminal and connected to the LC filter of the power supply. In the overcurrent protection circuit for the power supply having a choke coil having a very small equivalent resistance for its function, An input signal generator for inputting a reference voltage from an external source and outputting a first voltage having a temperature coefficient equal to a temperature coefficient of an equivalent resistance of the choke coil, and inputting the first voltage and converting the first voltage into a current signal to output the first voltage; A variable temperature offset generator including a voltage-current converter; And The output voltage terminal is connected to an inverting input terminal, the current sensing terminal is connected to a non-inverting input terminal, and an output of the temperature variable offset generator is applied between the output voltage terminal and the inverting input terminal And an comparing unit configured to generate and output an overcurrent protection signal according to a signal input to the inverting input terminal and the non-inverting input terminal. The method of claim 1, wherein the input signal generator A first resistor element having one terminal connected to the terminal for inputting the reference voltage; A second resistance element connected between the other terminal of the first resistance element and a ground terminal; A first N-type bipolar transistor connected to a terminal for inputting the reference voltage and gated by another terminal of the first resistance element; A diode having an anode terminal connected to an emitter terminal of the first N-type bipolar transistor; And An overcurrent protection circuit comprising a third resistance element connected between the cathode terminal of the diode and the ground terminal; The method of claim 2, wherein the voltage-current converter A current mirror circuit connected between a first terminal and a second terminal and supplying a current of the second terminal based on a current output from the first terminal; A fourth resistance element having one terminal connected to the ground terminal; An amplifier having a non-inverting input terminal connected to the cathode terminal of the diode and an inverting input terminal connected to the other terminal of the fourth resistance element; And a second N-type bipolar transistor connected between the first terminal and the other terminal of the fourth resistor, and gated by an output terminal of the amplifier. 4. The circuit of claim 3, wherein the current mirror circuit A first current mirror circuit connected between a third terminal and a fourth terminal and supplying a current of the third terminal based on a current output from the fourth terminal; And And a second current mirror circuit which is controlled by a current supplied from the first current mirror circuit and supplies a current of the second terminal based on a current output from the first terminal. . The circuit of claim 4, wherein the first current mirror circuit comprises: A third P-type bipolar transistor having an emitter terminal connected to the power supply terminal, a collector terminal connected to the third terminal, and gated by the fourth terminal; And And a fourth P-type bipolar transistor connected to an emitter terminal to said power supply terminal, to a collector terminal to said fourth terminal, and gated by said fourth terminal. 6. The method of claim 5, wherein the second current mirror circuit A fifth P-type bipolar transistor having an emitter terminal connected to the third terminal, a collector terminal connected to the first terminal, and gated by the first terminal; And And a sixth P-type bipolar transistor connected to an emitter terminal to said fourth terminal, to a collector terminal to said second terminal, and gated by said first terminal. The method of claim 6, wherein the comparison unit A current source having one terminal connected to the power supply terminal; A third current mirror circuit connected between a fifth terminal and a sixth terminal and supplying a current of the fifth terminal based on a current input to the sixth terminal; A seventh P-type bipolar transistor connected to the fourth terminal and gated by the first terminal; An eighth P-type bipolar transistor having an emitter terminal connected to the other terminal of the current source, a collector terminal connected to the fifth terminal, and gated by the second terminal; A ninth P-type bipolar transistor having an emitter terminal connected to the other terminal of the current source and a collector terminal connected to the sixth terminal, and gated by another terminal of the seventh P-type bipolar transistor; A fifth resistor element having one terminal connected to the other terminal of the seventh P-type bipolar transistor; A tenth P-type bipolar transistor having an emitter terminal connected to the fourth terminal and a collector terminal connected to the ground terminal and gated by a first terminal of the power supply device; An eleventh P-type bipolar transistor having an emitter terminal connected to the other terminal of the fifth resistor element, a collector terminal connected to the ground terminal, and gated by the second terminal of the current supply device; A sixth resistor element connected between the power supply terminal and the output terminal; And And an twelfth N-type bipolar transistor connected to the output terminal, to the ground terminal, to an emitter terminal, and to a gated by the fifth terminal. 8. The circuit of claim 7, wherein the third current mirror circuit is A thirteenth N-type bipolar transistor having a collector terminal connected to the fifth terminal and an emitter terminal connected to the ground terminal and gated by the sixth terminal; And And a fourteenth N-type bipolar transistor connected to the sixth terminal and to an emitter terminal connected to the ground terminal and gated by the sixth terminal. The method of claim 1, wherein the input signal generator A first resistor element having one terminal connected to the terminal for inputting the reference voltage; A second resistance element connected between the other terminal of the first resistance element and a ground terminal; A first N-type bipolar transistor connected to a terminal for inputting the reference voltage and gated by another terminal of the first resistance element; First to n-th diodes connected in series between another terminal of the first N-type bipolar transistor and an input signal output terminal; An overcurrent protection circuit comprising a third resistor element connected between the input signal output terminal and the ground terminal; The method of claim 9, wherein the voltage-current converter A current mirror circuit connected between a first terminal and a second terminal and supplying a current of the second terminal based on a current output from the first terminal; A fourth resistance element having one terminal connected to the ground terminal; An amplifier having a non-inverting input terminal connected to the input signal output terminal and an inverting input terminal connected to the other terminal of the fourth resistance element; And a second N-type bipolar transistor connected to the first terminal, the emitter terminal connected to the other terminal of the fourth resistor, and gated by the output terminal of the amplifier. Overcurrent protection circuit. The circuit of claim 10, wherein the current mirror circuit comprises: A first current mirror circuit connected between a third terminal and a fourth terminal and supplying a current of the third terminal based on a current output from the fourth terminal; And And a second current mirror circuit which is controlled by a current supplied from the first current mirror circuit and supplies a current of the second terminal based on a current output from the first terminal. . 12. The circuit of claim 11 wherein the first current mirror circuit comprises: A third P-type bipolar transistor having an emitter terminal connected to the power supply terminal, a collector terminal connected to the third terminal, and gated by the fourth terminal; And And a fourth P-type bipolar transistor connected to an emitter terminal to said power supply terminal, to a collector terminal to said fourth terminal, and gated by said fourth terminal. The circuit of claim 12, wherein the second current mirror circuit comprises: A fifth P-type bipolar transistor having an emitter terminal connected to the third terminal, a collector terminal connected to the first terminal, and gated by the first terminal; And And a sixth P-type bipolar transistor connected to an emitter terminal to the fourth terminal and to a collector terminal to the second terminal and gated by the first terminal. The method of claim 13, wherein the comparison unit A current source having one terminal connected to the power supply terminal; A third current mirror circuit connected between a fifth terminal and a sixth terminal and supplying a current of the fifth terminal based on a current input to the sixth terminal; A seventh P-type bipolar transistor connected to the fourth terminal and gated by the first terminal; An eighth P-type bipolar transistor having an emitter terminal connected to the other terminal of the current source, a collector terminal connected to the fifth terminal, and gated by the second terminal; A ninth P-type bipolar transistor having an emitter terminal connected to the other terminal of the current source and a collector terminal connected to the sixth terminal, and gated by another terminal of the seventh P-type bipolar transistor; A fifth resistor element having one terminal connected to the other terminal of the seventh P-type bipolar transistor; A tenth P-type bipolar transistor having an emitter terminal connected to the fourth terminal and a collector terminal connected to the ground terminal and gated by a first terminal of the power supply device; An eleventh P-type bipolar transistor having an emitter terminal connected to the other terminal of the fifth resistor element, a collector terminal connected to the ground terminal, and gated by the second terminal of the current supply device; A sixth resistor element connected between the power supply terminal and the output terminal; And And an twelfth N-type bipolar transistor connected to the output terminal, to the ground terminal, to an emitter terminal, and to a gated by the fifth terminal. 15. The circuit of claim 14, wherein the third current mirror circuit is A thirteenth N-type bipolar transistor having a collector terminal connected to the fifth terminal and an emitter terminal connected to the ground terminal and gated by the sixth terminal; And And a fourteenth N-type bipolar transistor connected to the sixth terminal and to an emitter terminal connected to the ground terminal and gated by the sixth terminal.