JPWO2020070814A1 - Power converter control device and feedback control device - Google Patents

Abstract

The feedback control device (20) amplifies the deviation (e) between the command signal (r) and the output (y) of the control target (26), and calculates the operation amount (u) to be input to the control target (26). A control unit (24) is provided. The control unit (24) includes a first amplifier (241) that resonates at the frequency of the command signal (r) and a second amplifier (243) that performs proportional calculation or proportional integration calculation. When the transfer function of the first amplifier (241) is represented by K1 (s) and the transfer function of the second amplifier (243) is represented by K2 (s), the transfer function of the control unit (24) is K2 (s). ) (1-K1 (s)). Further, the first amplifier (241) has a negative phase response to the frequency, and the gain is monotonically reduced in the high frequency region than the frequency of the command signal.

Description

The present invention relates to a control device and a feedback control device of a power converter that converts AC power output from an AC power source into DC power.

There is a pulse width modulation (PWM) converter as a power converter connected to an AC power source and mutually converting AC power and DC power. The role of the PWM converter is roughly divided into two. One is the control of DC voltage, and the other is the control of AC current. Hereinafter, the PWM converter is simply referred to as a "converter".

There are roughly two methods in which a converter controls alternating current. One is a method in which the feedback value of the alternating current is separated into the active current and the reactive current of the DC amount by the rotational coordinate conversion based on the phase of the power supply voltage, and these are made to follow the command value of the DC amount. The other is a method in which the feedback value of the alternating current is directly followed by the sinusoidal current command value. Hereinafter, the former is referred to as "DC-ACR (Direct Current-Automatic Currant Regulator)", and the latter is referred to as "AC-ACR (Alternate Current-Automatic Currant Regulator)". Generally, in the AC-ACR method, a proportional controller (hereinafter referred to as "P controller") or a proportional integral controller (hereinafter referred to as "PI controller") is used. The following Patent Document 1 discloses an AC-ACR type control device.

Japanese Unexamined Patent Publication No. 11-32486

The DC-ACR method has an advantage that the active current and the reactive current can be controlled without steady-state deviation. However, in the DC-ACR system, when the power supply is single-phase alternating current, it is necessary to prepare a signal having a phase difference of 90 ° with respect to the original alternating current in order to define a rotating instantaneous current vector. .. This preparatory process causes a delay in the output of the signal. Therefore, the DC-ACR method has a drawback that the control response cannot be designed to be high. Further, the DC-ACR method has a drawback that the step-like disturbance superimposed on the alternating current cannot be suppressed.

On the other hand, the AC-ACR method has a high control response and is easy to design, and has high suppression performance against step-like disturbance. However, since the AC-ACR method uses a P controller or a PI controller, there is a problem that a steady deviation remains with respect to the sine wave command value.

The present invention has been made in view of the above, and an object of the present invention is to obtain a power converter control device and a feedback control device capable of reducing a steady deviation with respect to a sine wave command value in the AC-ACR method. To do.

In order to solve the above-mentioned problems and achieve the object, the feedback control device according to the present invention includes a control unit that amplifies the deviation between the command signal and the output of the control target and calculates the operation amount to be input to the control target. .. The control unit includes a first amplifier that resonates at the frequency of the command signal and a second amplifier that performs proportional calculation or proportional integration calculation. When the transfer function of the first amplifier is represented by K1 (s) and the transfer function of the second amplifier is represented by K2 (s), the transfer function of the control unit is K2 (s) (1-K1 (s)). It is represented by. Further, in the first amplifier, the phase response to the frequency of the command signal is negative, and the gain is monotonically reduced in the high frequency region than the frequency of the command signal.

According to the present invention, it is possible to reduce the steady-state deviation with respect to the sine wave command value while maintaining the suppression performance against step-like disturbance.

Configuration diagram of the drive system including the control device of the power converter according to the first embodiment. A block diagram showing the configuration of the control calculation unit shown in FIG. A block diagram showing a detailed configuration of the voltage control unit shown in FIG. A vector diagram illustrating the operation of the converter during power running according to the first embodiment. A vector diagram illustrating the operation of the converter during regeneration according to the first embodiment. A block diagram showing a current control system including a current control unit according to the first embodiment. A vector diagram illustrating an operation during power running when the operating state of the converter of the first embodiment is not a power factor of 1. A block diagram showing a detailed configuration of the current control unit according to the first embodiment. Bode diagram showing the frequency characteristics of the first amplifier according to the first embodiment configured by the secondary lag system. Bode diagram for explaining the effect when the current control unit according to the first embodiment is used. Block diagram showing a configuration example of the current controller according to Comparative Example 1. Bode diagram showing the frequency characteristics of the bandpass filter according to Comparative Example 1. Bode diagram showing the frequency characteristics of the open-loop transfer function of the current control system according to Comparative Example 1. Block diagram showing a configuration example of the current controller according to Comparative Example 2. Bode diagram showing the frequency characteristics of the high-pass filter according to Comparative Example 2. Bode diagram showing the frequency characteristics of the open-loop transfer function of the current control system according to Comparative Example 2. A block diagram showing an example of a hardware configuration that realizes a calculation function in the control calculation unit of the first embodiment. A block diagram showing another example of a hardware configuration that realizes an arithmetic function in the control arithmetic unit of the first embodiment. The figure which shows the 1st modification of the current control part by Embodiment 1. The figure which shows the 2nd modification of the current control part by Embodiment 1. The figure which shows the 3rd modification of the current control part by Embodiment 1. A block diagram showing a configuration of a feedback control device according to a fourth embodiment.

The power converter control device and the feedback control device according to the embodiment of the present invention will be described in detail with reference to the accompanying drawings. The present invention is not limited to the following embodiments.

Embodiment 1.
FIG. 1 is a configuration diagram of a drive system 100 including a control device (hereinafter, simply referred to as “control device”) 10 of the power converter according to the first embodiment. The converter 2 is a power converter that converts AC power output from AC power supply 1 into DC power. The load 3 is a controlled object controlled by the converter 2. Seen from the converter 2, the side of the AC power supply 1 is called the "AC side", and the side of the load 3 is called the "DC side". When the control device 10 configures the feedback control device described later, the output of the main part on the AC side is fed back to the control device 10.

The DC power converted by the converter 2 is supplied to the load 3. The load 3 is a concept including an inverter (not shown) that converts DC power output from the converter 2 into AC power, and an actuator driven by the AC power of the inverter. An example of an actuator is a motor.

The converter 2 is connected to the AC power supply 1 via the impedance element 4. The impedance element 4 includes an impedance element of electrical wiring (not shown) that connects the AC power supply 1 and the converter 2, an internal impedance element (not shown) of the AC power supply 1.

The control device 10 has a control calculation unit 10a that generates a PWM signal. The control calculation unit 10a controls the converter 2 by generating a PWM signal for PWM control of the converter 2. Depending on the relationship between the output of the AC power supply 1 and the output of the load 3, the AC power output by the AC power supply 1 may be output to the load 3, or the DC power output by the load 3 is output to the AC power supply 1. In some cases. That is, the converter 2 has a function of mutually converting the AC power output by the AC power supply 1 and the DC power output by the load 3. There are many publicly known documents regarding PWM control related to the converter 2, and detailed description thereof will be omitted here.

FIG. 2 is a block diagram showing a configuration of the control calculation unit 10a shown in FIG. The configuration shown in FIG. 2 is a diagram for explaining an outline of converter control in the present embodiment.

As shown in FIG. 2, the control calculation unit 10a includes a phase calculation unit 11, subtractors 12, 14 and 16, a voltage control unit 13, a current control unit 15, and a switching command generation unit 17. The operation of each part will be described below.

The phase calculation unit 11 generates the power supply voltage phase θ based on the power supply voltage vs. The power supply voltage vs. is the output voltage of the AC power supply 1. The power supply voltage phase θ is a reference phase when the current command value is * is generated. The power supply voltage phase θ is input to the voltage control unit 13.

The subtractor 12 generates a DC voltage deviation ΔEd, which is a deviation between the DC voltage command value Ed * and the DC voltage Ed. The DC voltage Ed is a detected value of the actual DC voltage applied to the load 3.

The voltage control unit 13 generates the current command value is * based on the DC voltage deviation ΔEd and the power supply voltage phase θ. The current command value is * is a target value of the current to be passed to the AC side of the converter 2. The current command value is * is input to the subtractor 14.

The subtractor 14 generates a current control deviation Δis, which is a deviation between the current command value is * and the alternating current is. The alternating current is is a detected value of the actual alternating current flowing in and out of the converter 2. The current control deviation Δis is input to the current control unit 15.

The current control unit 15 generates a voltage compensation amount Δvc based on the current control deviation Δis. The details of the voltage compensation amount Δvc will be described later. The voltage compensation amount Δvc is input to the subtractor 16.

The subtractor 16 generates an AC voltage command vc * obtained by subtracting the voltage compensation amount Δvc from the feedforward voltage vFF, which will be described in detail later. The AC voltage command vc * is a target value of the AC voltage to be output to the AC side of the converter 2. The details of the feedforward voltage vFF will be described later.

The switching command generation unit 17 generates a switching command sw * based on the AC voltage command vc *. The switching command sw * is a PWM signal for PWM control of the converter 2.

In the subtractor 16 of FIG. 2, the voltage compensation amount Δvc is subtracted from the feedforward voltage vFF. In the subtractor 16, the voltage compensation amount Δvc is subtracted rather than added because the direction from the AC power supply 1 to the converter 2 is defined as the positive AC current is. That is, the polarity of the voltage compensation amount Δvc can be determined according to the polarity of the alternating current is.

FIG. 3 is a block diagram showing a detailed configuration of the voltage control unit 13 shown in FIG. As shown in FIG. 3, the voltage control unit 13 includes a unit sine wave generation unit 13a, a PI controller 13b (denoted as “PI” in FIG. 3), and a multiplier 13c.

As described above, the subtractor 12 generates the DC voltage deviation ΔEd. The PI controller 13b amplifies the DC voltage deviation ΔEd. The output of the PI controller 13b is a DC amount corresponding to the amplitude of the current command value is *. In FIG. 3, this DC amount is expressed as | is * |.

The unit sine wave generation unit 13a generates a unit sine wave sin θ having the same phase as the power supply voltage vs and an amplitude of 1. The multiplier 13c multiplies the DC amount | is * | by the unit sine wave sinθ to generate the current command value is *. The current command value is * is the output of the voltage control unit 13. The current command value is * is the amount of alternating current that oscillates at the frequency of the power supply voltage vs.

Next, the operation of the converter 2 according to the first embodiment will be described. First, in FIG. 1, the circuit equation of the following equation holds between the power supply voltage vs. the AC voltage vc and the AC current is.

In the above equation (1), "r" is a resistance component in the impedance element 4, and "x" is a reactance component in the impedance element 4.

Further, when the above equation (1) is expressed in a vector diagram, it becomes as shown in FIGS. 4 and 5. FIG. 4 is a vector diagram illustrating the operation of the converter 2 during power running according to the first embodiment. FIG. 5 is a vector diagram illustrating the operation of the converter 2 at the time of regeneration according to the first embodiment. Both FIGS. 4 and 5 show an operating state with a power factor of 1 in which the power supply voltage vs. the alternating current is does not have a component orthogonal to each other.

In the operating state with a power factor of 1, during power running, the power supply voltage vs. the alternating current is are in phase, as shown in FIG. Therefore, the flow of electric power in the drive system 100 is directed from the AC power supply 1 to the DC side of the converter 2. On the other hand, in the operating state with a power factor of 1, during regeneration, the power supply voltage vs. the alternating current is are in opposite phases, as shown in FIG. Therefore, the flow of electric power in the drive system 100 is directed from the DC side of the converter 2 to the AC power supply 1.

The operating principle of the converter 2 is to control the voltage across the impedance element 4 by controlling the amplitude and phase of the AC voltage vc, and to control the AC current is to a desired value. According to the above equation (1), once the desired AC current is is determined, the required AC voltage vc can be calculated by using the power supply voltage vs. the resistance component r and the reactance component x of the impedance element 4. This required AC voltage vc is called "feedforward voltage" and is represented by "vFF" as described above. The feedforward voltage vFF is expressed by the following equation.

In the above equation (2), there is a method of using the feedback value of the alternating current is instead of the current command value is *, but the present embodiment is not limited to either of them.

Here, in the example of FIG. 2, the AC voltage command vc * to the converter 2 is obtained by subtracting the voltage compensation amount Δvc from the feedforward voltage vFF. That is, most of the AC voltage command vc * is a component of the feed forward voltage vFF, and the current control unit 15 is affected by the voltage compensation amount Δvc that works to suppress the influence of unknown disturbance, parameter error of the impedance element 4, and the like. The output will be adjusted.

Next, the current control system according to the first embodiment will be described. FIG. 6 is a block diagram showing a current control system including the current control unit 15 according to the first embodiment. The meanings of the symbols shown in FIG. 6 are as follows.

R (s): Command signal (corresponds to the current command value is *)
Y (s): Output (corresponding to alternating current is)
E (s): Control deviation (corresponding to current control deviation Δis)
K (s): Transfer function of current control unit 15 P (s): Transfer function of control target 52 (corresponding to impedance element 4)

The transfer function of the control target 52 can be represented by a first-order lag system of P (s) = 1 / (R + sL), where R is the resistance component of the impedance element 4 and L is the inductance component. "S" is a Laplace operator. In general, the inductance component L is sufficiently larger than the resistance component R, so the resistance component R may be ignored and P (s) = 1 / sL may be set.

Here, in order to understand the operation of the current control system, it is assumed that the controller of the current control unit 15 is a P controller, and K (s) = Kc. At this time, the transfer function from the command signal R (s) to the control deviation E (s) is expressed by the following equation.

Further, the input to the current control system is assumed to be a sine wave input, and r (t) = sinωt is set. r (t) is a time function of the input signal to the current control system. At this time, since the Laplace transform R (s) of the input signal r (t) is given by R (s) = ω / (s ² + ω ² ), the control deviation E (s) is expressed by the following equation. ..

The following equation is obtained by modifying the above equation (4).

Further, when the time waveform is obtained by inverse Laplace transform of the above equation (5), the following equation is obtained.

In the above equation (6), the third term is an exponential function of monotonically decreasing, and converges to zero with the passage of time. On the other hand, the first term and the second term do not become zero and continue to vibrate at the vibration frequency. Although the calculation is omitted, the vibration term of the control deviation does not disappear even if the controller is controlled by PI. That is, the control deviation remains in the current control system.

Further, while the input signal r (t) is a sine wave, the time function e (t) of the control deviation E (s) also includes a cosine wave. That is, since r (t) and e (t) have different phases, r (t) and y (t) also have different phases. This means that the phase between the current command value is * and the alternating current is is out of phase, and the power factor is not controlled as intended.

It is expected that the feedforward voltage vFF described above reduces the current control deviation Δis, which is the control deviation of the current control system. However, compensation by feedforward voltage vFF (referred to here as “feedforward compensation”) is not robust against unknown disturbances or parameter fluctuations of the controlled object 52. That is, the feedforward compensation is not sufficiently effective when there is an unknown disturbance or a parameter fluctuation in the controlled object 52.

When the current control deviation Δis occurs in the current control system, the problem is that the power factor does not reach the desired value. In the control calculation unit 10a, the amount of effective current required to control the DC voltage Ed to be constant is automatically achieved without deviation by the function of the voltage control unit 13. Therefore, the influence of the current control deviation Δis is mainly reflected in the reactive current amount or the power factor.

Next, the operation of the converter 2 when the operating state is not the power factor 1 will be described. FIG. 7 is a vector diagram illustrating the operation during power running when the operating state of the converter 2 of the first embodiment is not the power factor 1. Here, as shown in FIG. 7, a case where the phase of the alternating current is is delayed with respect to the power supply voltage vs. is taken as an example.

In FIG. 7, the point A is the end point of the vector of the AC voltage vc'that generates the AC current is, and the point C is the end point of the vector of the AC voltage vc that generates the current command value is *. From the vector diagram of FIG. 7, it can be seen that a desired AC current can be achieved if the AC voltage output of the converter 2 can be changed from vc'to vc. That is, it can be considered that the difference between the AC voltage vc'that generates the AC current is and the AC voltage vc that generates the current command value is * is the voltage compensation amount Δvc to be output by the current control unit 15.

Next, in FIG. 7, attention is paid to the two vectors constituting the voltage compensation amount Δvc. One of the two vectors is rΔis, which is in phase with the current control deviation Δis, and the other of the two vectors is jxΔis, which is 90 degrees ahead of the current control deviation Δis. Here, in the current control unit according to the conventional method, the amplifier is configured by P control or PI control. Therefore, the current control unit according to the conventional method can output only signals having the same phase or the lag phase as the current control deviation Δis. The detailed operation of the current control unit by the conventional method will be described later.

FIG. 8 is a block diagram showing a detailed configuration of the current control unit 15 according to the first embodiment. In FIG. 8, the same or equivalent components as those in FIG. 2 are indicated by the same reference numerals and symbols.

As described above, the current control unit 15 is a control unit that amplifies the current control deviation Δis, which is the deviation between the current command value is * and the alternating current is, and outputs the voltage compensation amount Δvc. As shown in FIG. 8, the current control unit 15 includes a first amplifier 151, a subtractor 152, and a second amplifier 153. K1 (s) is the transfer function of the first amplifier 151, and K2 (s) is the transfer function of the second amplifier 153.

The first amplifier 151 amplifies the current control deviation Δis. The subtractor 152 produces a difference between the current control deviation Δis and the output of the first amplifier 151. The second amplifier 153 amplifies the output of the subtractor 152, that is, the difference between the current control deviation Δis and the output of the first amplifier 151.

The second amplifier 153 is a P controller or a PI controller, like the amplifier of the current control unit according to the conventional method. On the other hand, the first amplifier 151 is a controller that satisfies the following conditions.

-Resonance characteristics are obtained at the frequency of the power supply voltage vs.
-Has a negative phase response characteristic at a frequency of power supply voltage vs.
-The gain is monotonically reduced in the high frequency range from the frequency of the power supply voltage vs.

At this time, the voltage compensation amount Δvc, which is the output of the current control unit 15 according to the first embodiment, is expressed by the following equation.

Here, assuming that the second amplifier 153 is a P controller, the first term of the above equation (7) is a signal having the same phase as the current control deviation Δis. On the other hand, the second term of the above equation (7) becomes a leading phase with respect to the current control deviation Δis when a negative sign is included. Therefore, according to the above equation (7), the first term operates so as to compensate the voltage corresponding to the vector BA of FIG. 7, and the second term compensates the voltage corresponding to the vector CB of FIG. Operate. Even when the second amplifier 153 is used as the PI controller, the same relationship holds if the time constant of the integrator used for the PI controller is sufficiently larger than the period of the power supply voltage vs. the period.

Further, in general, the reactance component x is sufficiently larger than the resistance component r. Therefore, the voltage compensation amount Δvc in FIG. 7 is dominated by the component (vector CB) in the lead phase rather than the component (vector BA) having the same phase as the current control deviation Δis. Therefore, even in the equation (7), it is necessary to make the second term function predominantly. This is achieved by designing the first amplifier 151 to have resonance characteristics at a frequency of power supply voltage vs. frequency.

Further, since the first amplifier 151 has a cutoff characteristic in a frequency region higher than the frequency of the power supply voltage vs., the change in the gain characteristic and the phase characteristic in the frequency region can be reduced in the open loop transfer function of the current control system. In other words, it is possible to suppress deterioration of the stability of the current control system as compared with the case where the current control system is configured only by the second amplifier 153. The stability of the current control system when the first amplifier 151 is provided will be described later in comparison with another known method.

Here, again, attention is paid to the phase difference δ between the current control deviation Δis in FIG. 7 and the voltage compensation amount Δvc. The phase difference δ represents the impedance angle, and δ = tan ^-1 (x / r). As described above, since the reactance component x is sufficiently larger than the resistance component r, the voltage compensation amount Δvc has a relationship of approximately 90 degrees advance with respect to the current control deviation Δis. Therefore, the conditions of the first amplifier 151 described above may be changed as follows.

-Resonates at a frequency of power supply voltage vs.
-The phase response is -90 degrees at the frequency of the power supply voltage vs.
-The gain is monotonically reduced in the high frequency range from the frequency of the power supply voltage vs.

In order to satisfy the above conditions, it is easy to configure the first amplifier 151 with a second-order lag system. The second-order lag system can be expressed by the transfer function of the following equation, where the proportional gain is α, the natural frequency is ωr, and the attenuation coefficient is ζ.

FIG. 9 is a Bode diagram showing the frequency characteristics of the first amplifier 151 according to the first embodiment configured by the secondary lag system. FIG. 9 shows the frequency characteristics of the second-order lag system according to the first embodiment when, for example, the proportional gain is 1 and the natural frequency is 120π [rad / s] (in the case of 60 [Hz]). In the second-order lag system according to the example of FIG. 9, at the natural frequency of 60 [Hz], the gain becomes maximum as shown in the upper part, and the phase response becomes "-90 degrees" as shown in the lower part. .. Further, as shown in the upper part, the gain is monotonically decreased in the high frequency region than the natural frequency. Therefore, if the natural frequency is matched with the frequency of the power supply voltage vs., it resonates at the frequency of the power supply voltage vs. the phase response is "-90 degrees" at the frequency of the power supply voltage vs. The three conditions that the gain is monotonically decreasing are achieved at the same time.

FIG. 10 is a Bode diagram for explaining the effect when the current control unit 15 according to the first embodiment is used. In FIG. 10, in both the upper and lower stages, the solid line is the frequency characteristic of the open-loop transfer function of the current control system according to the first embodiment in which the current control unit 15 is configured as shown in FIG. 8, and the broken line is the current control unit 15. Is the frequency characteristic of the open-loop transfer function of the current control system when only the second amplifier 153 is used. The broken line is shown as a reference characteristic for explaining the difference between the current control unit 15 of the first embodiment and the current control unit according to two comparative examples described later. Further, in FIG. 10, a dead time of 500 [μs] is assumed.

In FIG. 10, the open-loop transfer function of the current control system according to the first embodiment has a maximum gain at a natural frequency of 60 [Hz] and a phase of almost 0 degrees. This means that the current control deviation Δis is substantially in phase with respect to the current command value is *. At this time, the phase difference between the current command value is * and the alternating current is is also eliminated, so that the power factor can be controlled with high accuracy.

By the way, in the current control unit, a method of using a bandpass filter as an amplifier having a resonance characteristic is known. Hereinafter, the current controller by this method will be referred to as "current controller according to Comparative Example 1" or simply "Comparative Example 1". FIG. 11 is a block diagram showing a configuration example of the current control unit 15A according to Comparative Example 1. In the current control unit 15A according to Comparative Example 1, the first amplifier 151 shown in FIG. 8 is replaced with the bandpass filter 154. B (s) is a transfer function of the bandpass filter 154.

Further, in the current control unit 15A of FIG. 11, the subtractor 152 shown in FIG. 8 is changed to the adder 155. The subtractor 152 shown in FIG. 8 generates a difference between the current control deviation Δis and the output of the first amplifier 151, whereas the adder 155 shown in FIG. 11 has a bandpass filter for the current control deviation Δis. It should be noted that the output of 154 is added. The other configurations are the same as those in FIG. 8, and the same components are indicated by the same reference numerals and symbols.

The transfer function B (S) of the bandpass filter 154 can be expressed by the following equation, where the proportional gain is α, the center frequency is ωr, and the attenuation coefficient is ζ.

In the above equation (9), "1 / 2ζ" is also called sharpness, and "2ζωr" is also called bandwidth.

Next, the frequency characteristics of the current control system to which the current control unit 15A according to Comparative Example 1 is applied will be described. The center frequency ωr is made to match the frequency of the power supply voltage vs. in order to make the conditions at the time of comparison with the current control unit 15 according to the first embodiment the same.

FIG. 12 is a Bode diagram showing the frequency characteristics of the bandpass filter 154 according to Comparative Example 1. FIG. 12 shows the frequency characteristics of the bandpass filter 154 when the proportional gain is 1 and the center frequency is 120π [rad / s] (60 [Hz]). According to the example of FIG. 12, the gain is maximized at the center frequency, and the phase response is “0 degrees” at the center frequency, that is, the response characteristic that is not negative is shown.

FIG. 13 is a Bode diagram showing the frequency characteristics of the open-loop transfer function of the current control system according to Comparative Example 1. The frequency characteristics of the open-loop transfer function according to Comparative Example 1 are shown by solid lines in the upper and lower parts of FIG. Further, in FIG. 13, the broken line in both the upper part and the lower part is the reference characteristic shown by the broken line in FIG.

In the open-loop transfer function according to Comparative Example 1, the gain is maximized at the center frequency of the bandpass filter 154, and the phase is approximately "-90 degrees". According to Comparative Example 1, the gain can be selectively increased at the center frequency. Therefore, as in the first embodiment, the control deviation of the center frequency with respect to the sinusoidal command input can be reduced, and the power factor can be controlled with high accuracy.

However, as shown in the lower part of FIG. 13, the characteristic of Comparative Example 1 has a phase advance on the low frequency side of the center frequency as compared with the reference characteristic, but has a phase lag on the high frequency side of the center frequency. Is getting bigger. Therefore, if the second amplifier 153 is designed so that the response frequency (gain crossing frequency) is higher than the frequency of the power supply voltage vs. in order to increase the response, there arises a problem that the phase margin is reduced. Here, the "phase margin" is an amount indicating how much the phase at the frequency at which the gain becomes 0 dB has a margin with respect to −180 degrees in the Bode diagram of the open-loop transfer function, that is, the phase difference. is there. The frequency at which the gain becomes 0 dB is generally called a "response frequency" or a "gain crossing frequency".

On the other hand, in the case of the current control unit 15 according to the first embodiment described above, the phase characteristic of the open-loop transfer function is only in the direction of advancing the phase with respect to the reference characteristic, as shown in the lower part of FIG. It does not change. Therefore, according to the first embodiment, even when the current control system is designed to have a high response, a sufficient phase margin can be secured.

Further, as another method of applying an amplifier having resonance characteristics to the current control unit, a method of using a high-pass filter is also known. Hereinafter, the current controller by this method will be referred to as "current controller according to Comparative Example 2" or simply "Comparative Example 2". FIG. 14 is a block diagram showing a configuration example of the current control unit 15B according to Comparative Example 2. In the current control unit 15B according to Comparative Example 2, the first amplifier 151 shown in FIG. 8 is replaced with the high-pass filter 156. C (s) is the transfer function of the high-pass filter 156. In the current control unit 15B of FIG. 14, the output of the high-pass filter 156 is added to the current control deviation Δis in the adder 155, similarly to the current control unit 15A of FIG. Other configurations are the same as those in FIG. 8, and the same components are indicated by the same reference numerals and symbols.

The transfer function C (s) of the high-pass filter 156 can be expressed by the following equation, where the proportional gain is α, the cutoff frequency is ωr, and the attenuation coefficient is ζ.

Next, the frequency characteristics of the current control system to which the current control unit 15B according to Comparative Example 2 is applied will be described. The cutoff frequency ωr is made to match the frequency of the power supply voltage vs. in order to make the conditions at the time of comparison with the current control unit 15 according to the first embodiment equivalent.

FIG. 15 is a Bode diagram showing the frequency characteristics of the high-pass filter 156 according to Comparative Example 2. FIG. 15 shows the frequency characteristics of the high-pass filter 156 when the proportional gain is 1 and the cutoff frequency is 120π [rad / s] (60 [Hz]). According to the example of FIG. 15, the gain is maximized at the cutoff frequency, and the phase response is “+90 degrees” at the cutoff frequency.

FIG. 16 is a Bode diagram showing the frequency characteristics of the open-loop transfer function of the current control system according to Comparative Example 2. The frequency characteristics of the open-loop transfer function according to Comparative Example 2 are shown by solid lines in the upper and lower parts of FIG. Further, in FIG. 16, the broken lines in both the upper part and the lower part are the reference characteristics shown by the broken lines in FIGS. 10 and 13.

In the open-loop transfer function according to Comparative Example 2, the gain is maximized at the cutoff frequency of the high-pass filter 156, and the phase is substantially "0 degree". This means that the current control deviation Δis is substantially in phase with respect to the current command value is *. At this time, the phase difference between the current command value is * and the alternating current is is also eliminated, so that the power factor can be controlled with high accuracy.

However, as shown in the upper part of FIG. 16, in the characteristic of Comparative Example 2, the gain is offset in the increasing direction on the high frequency side of the cutoff frequency as compared with the reference characteristic. At this time, the response frequency (gain crossing frequency) is shifted to the high frequency side as compared with the reference characteristic. Therefore, if the second amplifier 153 is designed so that the response frequency (gain crossing frequency) is higher than the frequency of the power supply voltage vs. in order to increase the response, there arises a problem that the gain margin and the phase margin are reduced.

Further, when the current control unit is configured only by the second amplifier 153, it is easily desired by simply determining the proportional gain of the second amplifier 153 corresponding to the magnitude of the inductance component L of the controlled object 52. A response frequency can be achieved. On the other hand, when the current control unit is configured by using the high-pass filter 156 in combination, the parameters of the high-pass filter 156 and the parameters of the second amplifier 153 influence each other to change the response frequency. There is the problem of complexity.

On the other hand, in the case of the current control unit 15 according to the first embodiment described above, the gain characteristic of the open-loop transfer function is on the higher frequency side than the natural frequency with respect to the reference characteristic, as shown in the upper part of FIG. The amount of offset at is small. Therefore, the control response on the higher frequency side than the natural frequency is predominantly determined only by the parameter of the second amplifier 153. As a result, the controller can be easily designed, and a sufficient gain margin and phase margin can be secured.

As described above, the current control unit in the first embodiment amplifies the deviation between the current command value, which is the target value of the current to be passed to the AC side of the power converter, and the actual current on the AC side. The entire transfer function including the first amplifier and the second amplifier by the first amplifier whose transfer function is represented by K1 (s) and the second amplifier whose transfer function is represented by K2 (s). Is represented by K2 (s) (1-K1 (s)). Then, the first amplifier resonates at least at the frequency of the AC power supply, and the second amplifier performs proportional calculation or proportional integration calculation. As a result, it is possible to reduce the steady-state deviation with respect to the sine wave command value while maintaining the suppression performance against step-like disturbance.

Further, in the control device according to the first embodiment, if the first amplifier has a negative phase response characteristic at the frequency of the power supply voltage and the gain monotonically decreases in the high frequency range than the frequency of the power supply voltage. It is possible to secure a sufficient gain margin and phase margin while ensuring the ease of design in the current controller.

Next, the hardware configuration for realizing the calculation function in the control calculation unit 10a of the first embodiment will be described with reference to the drawings of FIGS. 17 and 18. FIG. 17 is a block diagram showing an example of a hardware configuration that realizes a calculation function in the control calculation unit 10a of the first embodiment. FIG. 18 is a block diagram showing another example of a hardware configuration that realizes a calculation function in the control calculation unit 10a of the first embodiment.

When all or part of the calculation function in the control calculation unit 10a of the first embodiment is realized by software, as shown in FIG. 17, the processor 300 that performs the calculation and the program read by the processor 300 are stored. The memory 302 and the interface 304 for inputting / outputting signals can be included.

The processor 300 may be a computing means such as an arithmetic unit, a microprocessor, a microcomputer, a CPU (Central Processing Unit), or a DSP (Digital Signal Processor). Further, the memory 302 includes a non-volatile or volatile semiconductor memory such as a RAM (Random Access Memory), a ROM (Read Only Memory), a flash memory, an EPROM (Erasable Program ROM), and an EPROM (registered trademark) (Electrically EPROM). Examples thereof include magnetic discs, flexible discs, optical discs, compact discs, mini discs, and DVDs (Digital Versailles Disc).

The memory 302 stores a program that executes all or a part of the calculation functions of the control calculation unit 10a. The processor 300 exchanges necessary information via the interface 304, and the processor 300 executes a program stored in the memory 302 to perform PWM control on the converter 2.

Further, the processor 300 and the memory 302 shown in FIG. 17 may be replaced with the processing circuit 303 as shown in FIG. The processing circuit 303 corresponds to a single circuit, a composite circuit, an ASIC (Application Specific Integrated Circuit), an FPGA (Field-Programmable Gate Array), or a combination thereof.

Embodiment 2.
In the second embodiment, the stability conditions in the current control system of the control device 10 according to the first embodiment will be considered. First, the open-loop transfer function of the current control system to which the current control unit 15 according to the first embodiment is applied is expressed by the following equation.

In the above equation (11), if the denominator polynomial is D (s) and the numerator polynomial is N (s), the closed loop transfer function of the current control system is expressed by the following equation.

Further, the proportional gain is Kp, the integration time constant is Ti, and the transfer function K2 (s) of the second amplifier 153 is set as follows.

Here, in order to evaluate the stability of the current control system, the characteristic polynomial φ (s) of the current control system is obtained. The characteristic polynomial φ (s) is expressed by the following equation.

In the above equation (14), a0 to a4 are coefficients of the characteristic polynomial φ (s), and α is a proportional gain of the first amplifier 151. Of the coefficients a0 to a4, the expressions representing the coefficients a2 to a4 do not include a negative sign, and all the constants are positive. Therefore, the values of the coefficients a2 to a4 are positive. Therefore, in order for the current control system to be stable, at least the coefficients a0 and a1 of the polynomial must be positive. First, the following equation is obtained from the condition of a1> 0.

Further, in order for a0> 0, α <1 may be sufficient. Here, the second and third terms on the right side of the above equation (15) are clearly positive. Therefore, the necessary condition for the current control system to be stable is α <1. The condition for making the phase response of the first amplifier 151 negative is α> 0. Based on the above conditions, it is desirable that the proportional gain α of the first amplifier 151 is set in the range of 0 <α <1. Here, by substituting α = 1, which is a boundary condition, into the above equation (14) and evaluating the polynomial coefficients a0 to a4 based on Routh's stability determination method, it can be confirmed that the current control system is stable. Further, when α = 0, the first amplifier becomes invalid and the configuration is the same as that of the conventional current control system. Therefore, if the proportional gain α of the first amplifier 151 is set in the range of 0 <α <1, the stability of the current control system can be maintained without adding another control system.

The frequency characteristic of the transfer function may be calculated by replacing the Laplace operator of the transfer function with s = jω. Further, the equation (8) representing the transfer function of the first amplifier 151 agrees with the proportional gain α when s = 0. Therefore, "setting the proportional gain α of the first amplifier 151 in the range of 0 <α <1" can be rephrased as "setting the gain for the DC component in the range of 0 <α <1". Good.

As described above, in the control device according to the second embodiment, since the proportional gain α of the first amplifier is set in the range of 0 <α <1, current control is performed without adding another control system. The stability of the system can be maintained.

Embodiment 3.
In the third embodiment, a modification of the current control unit 15 according to the first embodiment shown in FIG. 8 will be described. The current control unit 15 according to the first embodiment may be configured as shown in the following modification.

FIG. 19 is a diagram showing a first modification of the current control unit 15 according to the first embodiment. In FIG. 19, the same or equivalent components as those in FIG. 8 are indicated by the same reference numerals and symbols.

In the current control unit 15C shown in FIG. 19, the second amplifier 153 amplifies the current control deviation Δis. Further, the control system including the first amplifier 151 amplifies the output of the second amplifier 153. Then, the subtractor 152 outputs the difference between the output of the second amplifier 153 and the output of the first amplifier 151 as the voltage compensation amount Δvc.

FIG. 20 is a diagram showing a second modification of the current control unit 15 according to the first embodiment. In the current control unit 15D in FIG. 20, the same or equivalent components as those in FIG. 8 are indicated by the same reference numerals and symbols.

In the current control unit 15D shown in FIG. 20, the first amplifier 151 amplifies the current control deviation Δis. Further, there are two second amplifiers 153, one second amplifier 153a amplifies the current control deviation Δis, and the other second amplifier 153b amplifies the output of the first amplifier 151. Then, the subtractor 157 outputs the difference between the output of one second amplifier 153a and the output of the other second amplifier 153b as the voltage compensation amount Δvc.

FIG. 21 is a diagram showing a third modification of the current control unit 15 according to the first embodiment. In the current control unit 15E in FIG. 21, the same or equivalent components as those in FIG. 8 are indicated by the same reference numerals and symbols.

Also in the current control unit 15E shown in FIG. 21, there are two second amplifiers 153, and both the second amplifier 153a and the second amplifier 153b amplify the current control deviation Δis. Further, the first amplifier 151 amplifies the output of the other second amplifier 153b. Then, the subtractor 157 outputs the difference between the output of the second amplifier 153a and the output of the first amplifier 151 as the voltage compensation amount Δvc.

In any of the first to third modifications shown in FIGS. 19 to 21, the second amplifier 153 or the second amplifiers 153a and 153b are P controllers or PI controllers. Further, the first amplifier 151 satisfies the same conditions as in the first embodiment.

The current control units 15, 15C, 15D, and 15E shown in FIGS. 8 and 19 to 21 can all be represented by transfer functions of the following equations that are equivalent in mathematical expressions.

As described above, the current control unit according to the present invention is a linear system composed of three types of controllers: a first amplifier, a second amplifier, and a subtractor. Therefore, any configuration other than those shown in FIGS. 8 and 19 to 21 may be used as long as it can be equivalently converted by the commutative law, the distributive law, or the associative law. That is, the current control unit according to the present invention may have any configuration as long as the transfer function of the current control unit can be expressed in the form of the above equation (16).

Embodiment 4.
FIG. 22 is a block diagram showing the configuration of the feedback control device 20 according to the fourth embodiment. FIG. 22 shows the current control system of the first embodiment shown in FIG. 6 in a block diagram of a more generalized feedback control system. In FIG. 22, the same or equivalent components as those in FIGS. 6 and 8 are indicated by the same reference numerals and symbols.

In FIG. 22, the feedback control device 20 according to the fourth embodiment has a subtractor 14 and a control unit 24. The control unit 24 corresponds to the current control unit 15 shown in FIG. There is a control target 26 on the output side of the control unit 24, and the model output y, which is the output of the control target 26, is fed back to the subtractor 14. The subtractor 14 generates a control deviation e by subtracting the model output y from the command signal r. The control unit 24 amplifies the control deviation e and calculates the operation amount u to be input to the control target 26. K (s) is the transfer function of the control unit 24, and P (s) is the transfer function of the controlled object 26.

Further, in FIG. 22, the control unit 24 has a first amplifier 241 and a subtractor 242, and a second amplifier 243. The first amplifier 241 corresponds to the first amplifier 151 shown in FIG. 8, the subtractor 242 corresponds to the subtractor 152 shown in FIG. 8, and the second amplifier 243 corresponds to the subtractor 152 shown in FIG. Corresponds to the second amplifier 153. K1 (s) is the transfer function of the first amplifier 241 and K2 (s) is the transfer function of the second amplifier 243.

Here, it is assumed that the first amplifier 241 is a controller that satisfies the following conditions when the command signal r is a sinusoidal signal that oscillates at the frequency ωr.

・ Resonance characteristics occur at the frequency ωr.
-Has a negative phase response characteristic at the frequency ωr.
-The gain is monotonically decreasing in the high frequency range from the frequency ωr.

Further, as in the first embodiment, the transfer function P (s) of the control target 26 is represented by P (s) = 1 / (R + sL) as the first-order lag format, and the model constant of the control target 26 is It is assumed that R << ωrL holds in the vicinity of the frequency ωr. At this time, the first amplifier 241 may satisfy the following conditions.

・ Resonance characteristics occur at the frequency ωr.
-The phase response at the frequency ωr is -90 degrees.
-The gain is monotonically decreasing in the high frequency range from the frequency ωr.

Further, as a simple amplifier that satisfies the above conditions, the first amplifier 241 may be a second-order lag system as represented by the above equation (8). In this case, α, which is the proportional gain of the first amplifier 241, is in the range of 0 <α <1. If the proportional gain α is set to a value within this range, the control system of the feedback control device 20 can be stabilized by the control unit 24. As the second amplifier 243, a P controller or a PI controller is used as in the first embodiment.

At this time, the open-loop transfer function of the feedback control system shown in FIG. 22 has a frequency characteristic as shown by the solid line in FIG. 10, for example. In FIG. 10, the frequency ωr corresponds to the natural frequency, and the open-loop transfer function of the feedback control system has a maximum gain at the frequency ωr and a phase of almost 0 degrees. This means that the control deviation e is substantially in phase with respect to the command signal r oscillating at the frequency ωr, which further means that the model output y is substantially in phase with respect to the command signal r. It means that. Therefore, the followability of the model output y with respect to the command signal r is improved.

Further, the open-loop transfer function of the feedback control system shown in FIG. 22 has a frequency ωr as compared with the characteristic shown by the broken line in FIG. 10, that is, the characteristic when the control unit 24 is configured only by the second amplifier 243. There is almost no gain increase or phase lag on the higher frequency side. Therefore, the control response on the high frequency side of the frequency ωr is predominantly determined only by the parameter of the second amplifier 243. Therefore, the feedback control device 20 according to the fourth embodiment can secure a sufficient gain margin and a phase margin while ensuring the ease of design in the control unit 24.

The configuration shown in the above-described embodiment shows an example of the content of the present invention, can be combined with another known technique, and is configured without departing from the gist of the present invention. It is also possible to omit or change a part of.

1 AC power supply, 2 converter, 3 load, 4 impedance element, 10 controller, 10a control calculation unit, 11 phase calculation unit, 12, 14, 16, 152, 157, 242 subtractor, 13 voltage control unit, 13a unit sine Wave generator, 13b PI controller, 13c multiplier, 15, 15A, 15B, 15C, 15D, 15E current control unit, 17 switching command generator, 20 feedback control device, 24 control unit, 26, 52 control target, 100 Drive system, 151,241 first amplifier, 153, 153a, 153b, 243 second amplifier, 154 bandpass filter, 155 adder, 156 highpass filter, 300 processor, 302 memory, 303 processing circuit, 304 interface.

Claims (4)

A feedback control device including a control unit that amplifies the deviation between the command signal and the output of the control target and calculates the amount of operation to be input to the control target.
The control unit
A first amplifier that resonates at the frequency of the command signal,
A second amplifier that performs proportional or proportional integral operations,
With
The transfer function of the first amplifier is represented by K1 (s).
When the transfer function of the second amplifier is represented by K2 (s),
The transfer function of the control unit in the previous term is represented by K2 (s) (1-K1 (s)).
The first amplifier
A feedback control device characterized in that the phase response to the frequency of the command signal is negative and the gain is monotonically decreased in a frequency region higher than the frequency of the command signal. The feedback control device according to claim 1, wherein the first amplifier is a secondary lag system in which the natural frequency matches the frequency of the command signal. The feedback control device according to claim 1 or 2, wherein the first amplifier has a gain for a DC component of more than 0 and less than 1. A control device for a power converter that converts AC power output from an AC power supply into DC power.
The control device is
A current control unit that generates a voltage compensation amount based on the deviation between the current command value, which is the target value of the current to be passed to the AC side of the power converter, and the actual current on the AC side, is provided.
The current control unit includes the control unit provided in the feedback control device according to any one of claims 1 to 3.
The frequency of the command signal is the frequency of the AC power supply.
A control device for a power converter, wherein the operation amount is the voltage compensation amount.

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