JPWO2018167952A1 - Adaptive array antenna device - Google Patents

Abstract

A plurality of signals respectively received by the M subarray antennas (1-m) during the listening period, and a plurality of signals respectively received by the M subarray antennas (1-m) during the beam transmission period From the above, a null shift estimation unit (11) is provided to estimate a null shift which is a change between the arrival direction of the interference wave in the listening period and the arrival direction of the interference wave in the beam transmission period, and the compensation weight calculation unit (12) Based on the null deviation estimated by the deviation estimation unit (11), null deviation is compensated during the beam transmission period as a weighting factor for a plurality of signals respectively received by the M sub-array antennas (1-m). Calculate compensation weights.

Description

  The present invention relates to an adaptive array antenna apparatus that multiplies a plurality of signals respectively received by a plurality of subarray antennas by a weighting factor, and combines the plurality of signals after weighting factor multiplication.

The radar apparatus is mounted on a platform such as an aircraft, for example, and may be operated in an environment where disturbance waves arrive.
The antenna device included in the radar device forms an adaptive beam by an array antenna in which a plurality of subarray antennas are arranged.
At this time, the antenna apparatus suppresses interference waves included in the reception signal of the array antenna to improve signal to jamming and noise ratio (SJNR) of the target signal related to the target to be observed. Adaptive weights are determined such that nulls are formed in the direction of arrival of. The adaptive weight is a weighting factor for a plurality of signals respectively received by a plurality of subarray antennas.

The determination of the adaptive weight by the antenna device is performed, for example, by the following method.
First, the antenna apparatus temporarily stops the beams transmitted from the array antenna, and sets a plurality of signals respectively received by the plurality of sub-array antennas as disturbance signals during a listening period in which no beam is transmitted. get.
Next, the antenna apparatus determines adaptive weights so that nulls are formed in the arrival direction of the interference wave from the acquired interference wave signal.

When a radar device provided with an antenna device is installed at a land base or the like and the radar device does not move, the direction of arrival of the disturbance wave does not change unless the source of the disturbance wave is moved.
However, when the radar device provided with the antenna device or the source of the disturbance wave is moving, the arrival direction of the disturbance wave changes.
For this reason, even if the antenna device determines an adaptive weight in which nulls are formed in the direction of arrival of the disturbance during the listening period, the disturbance is generated during a short time until the transmission of the beam from the array antenna is resumed. The direction of arrival of the waves changes.
That is, the arrival direction of the interference wave in the listening period and the arrival direction of the interference wave in the beam transmission period, which is a period in which the beam is transmitted after the listening period ends, are different.

According to the change in the arrival direction of the disturbance wave, the deviation between the direction of the null formed by the adaptive weight determined during the listening period and the actual arrival direction of the disturbance wave in the beam transmission period (hereinafter referred to as “null deviation” (Referred to as
The occurrence of the null shift degrades the suppression performance of the interference wave by the antenna device, and thus degrades the detection performance of the target by the radar device.
Non-Patent Document 1 below discloses an antenna device that performs processing to widen the width of a null formed by an adaptive weight determined during a listening period in order to avoid the influence of null shift.

J. R. Guerci, “Theory and application of covariance matrix tapers for robust adaptive beamforming,” IEEE Transactions on Signal Processing, vol. 47, no. 4, pp. 977-985, Apr 1999.

  The conventional antenna device carries out processing to widen the width of the null formed by the adaptive weight, but does not properly set the extension width of the null width based on the change in the arrival direction of the interference wave. For this reason, the arrival direction of the disturbance may cause a large change such as to exceed the expanded null width. As described above, when the change in the arrival direction of the interference wave is large, there is a problem that the interference wave contained in the reception signal of the array antenna can not be suppressed.

  The present invention has been made to solve the above problems, and an antenna apparatus capable of suppressing an interference wave contained in a received signal of an array antenna even if the arrival direction of the interference wave changes significantly. The aim is to get

  An antenna apparatus according to the present invention includes an array antenna in which a plurality of sub array antennas including one or more element antennas are arranged, and a plurality of sub array antennas during a listening period in which a beam is not transmitted from the array antenna. An interference wave in the listening period is made up of a plurality of signals respectively received and a plurality of signals respectively received by a plurality of sub array antennas during a beam transmission period in which a beam is transmitted after the end of the listening period. And a plurality of null deviation estimation units for estimating null deviation, which is a change between the arrival direction of the interference wave and the arrival direction of the interference wave in the beam transmission period, and a plurality of beam transmission periods based on the null deviation estimated by the null deviation estimation unit. Weighting coefficients for a plurality of signals respectively received by the subarray antenna of And a compensation weight calculation unit for calculating a compensation weight for compensating for a null shift, and the beam forming unit calculates a plurality of signals respectively received by a plurality of sub-array antennas by the compensation weight calculation unit during a beam transmission period. The above compensation weights are multiplied to synthesize a plurality of signals after the compensation weight multiplication.

  According to the present invention, interference in the listening period is caused by the plurality of signals respectively received by the plurality of subarray antennas during the listening period and the plurality of signals respectively received by the plurality of subarray antennas during the beam transmission period. A null shift estimation unit is provided for estimating a null shift which is a change between the arrival direction of the wave and the arrival direction of the interference wave in the beam transmission period, and the compensation weight calculation unit calculates the null shift based on the null shift estimated by the null shift estimation unit. Since the compensation weight for compensating for the null shift is calculated as a weighting factor for the plurality of signals respectively received by the plurality of subarray antennas during the beam transmission period, the arrival direction of the disturbance wave changes significantly. Also, there is an effect that the interference wave contained in the reception signal of the array antenna can be suppressed.

It is a block diagram which shows the adaptive array antenna apparatus by Embodiment 1 of this invention. It is a hardware block diagram which shows the signal processing apparatus 2 of the adaptive array antenna apparatus by Embodiment 1 of this invention. FIG. 17 is a hardware configuration diagram of a computer when the signal processing device 2 is realized by software or firmware. It is a block diagram which shows the null shift | offset | difference estimation part 11 of the adaptive array antenna apparatus by Embodiment 1 of this invention. It is a block diagram which shows the compensation weight calculation part 12 of the adaptive array antenna apparatus by Embodiment 1 of this invention. It is a flowchart which shows the process sequence in case the signal processing apparatus 2 is implement | achieved by software or a firmware. It is a block diagram which shows the compensation weight calculation part 12 of the adaptive array antenna apparatus by Embodiment 2 of this invention.

  Hereinafter, in order to explain the present invention in more detail, a mode for carrying out the present invention will be described according to the attached drawings.

Embodiment 1
FIG. 1 is a block diagram showing an adaptive array antenna apparatus according to Embodiment 1 of the present invention.
FIG. 2 is a hardware configuration diagram showing a signal processing device 2 of the adaptive array antenna device according to the first embodiment of the present invention.
1 and 2, the array antenna 1 is an antenna in which M sub array antennas 1-m (m = 1, 2,..., M) are arranged.
The sub array antenna 1-m is an antenna including at least one or more element antennas.

The signal processing device 2 includes a null deviation estimation unit 11, a compensation weight calculation unit 12, and a beam forming unit 13, and multiplies a plurality of signals respectively received by M sub-array antennas 1-m by weighting factors. , And an apparatus for combining a plurality of signals after multiplication by weight coefficients.
In FIG. 1, for simplification of the drawing, a receiver for detecting a signal received by the sub array antenna 1-m and a converter for converting a received signal of the receiver from an analog signal to a digital signal are omitted. In practice, it comprises a receiver and a converter.
For this reason, the null shift estimation unit 11 and the beam forming unit 13 are supplied with digital received signals corresponding to the signals received by the M sub-array antennas 1-m.
In FIG. 1, received signal vectors including M digital received signals are shown.

The null deviation estimation unit 11 is realized by, for example, the null deviation estimation circuit 21 shown in FIG.
The null shift estimation unit 11 receives M digital reception signals corresponding to M signals respectively received by M subarray antennas 1-m during a listening period in which no beam is transmitted from the array antenna 1. get.
Also, the null shift estimation unit 11 corresponds to M signals received by the M sub-array antennas 1-m, respectively, during a beam transmission period in which a beam is transmitted after the end of the listening period. Acquire M digital reception signals.
From the M digital reception signals in the listening period and the M digital reception signals in the beam transmission period, the null deviation estimation unit 11 receives the arrival direction of the interference wave in the listening period and the arrival of the interference wave in the beam transmission period. A process is performed to estimate the null deviation which is a change from the direction.
The beam transmission period includes a period in which the array antenna 1 receives a beam, as well as a period in which the array antenna 1 transmits a beam.

The compensation weight calculation unit 12 is realized by, for example, the compensation weight calculation circuit 22 shown in FIG.
The compensation weight calculation unit 12 calculates M signals corresponding to M signals respectively received by M sub-array antennas 1-m during the beam transmission period based on the null shift estimated by the null shift estimation unit 11. A process of calculating a compensation weight that compensates for null deviation is performed as a weighting factor for the digital received signal of.

The beam forming unit 13 is realized by, for example, a beam forming circuit 23 shown in FIG.
The beam forming unit 13 includes M multipliers 14-m and an adder 15, and M beams corresponding to M signals received by M sub-array antennas 1-m during a beam transmission period. A process of multiplying M digital received signals after the compensation weight multiplication is performed by multiplying the digital received signals by the compensation weight calculated by the compensation weight calculating unit 12.
The multiplier 14-m multiplies the compensation weight calculated by the compensation weight calculator 12 by the digital reception signal corresponding to the signal received by the sub array antenna 1-m during the beam transmission period, and performs multiplication by the compensation weight. The digital reception signal is output to the adder 15.
The adder 15 combines the digital reception signals after M compensation weights multiplied respectively output from the M multipliers 14-m and outputs the combined digital reception signal as a reception beam.

In FIG. 1, it is assumed that each of the null deviation estimation unit 11, the compensation weight calculation unit 12 and the beam forming unit 13 which are components of the signal processing apparatus 2 is realized by dedicated hardware as shown in FIG. doing. That is, what is realized by the null deviation estimation circuit 21, the compensation weight calculation circuit 22, and the beam forming circuit 23 is assumed.
The null deviation estimation circuit 21, the compensation weight calculation circuit 22 and the beam forming circuit 23 may be, for example, a single circuit, a composite circuit, a programmed processor, a parallel programmed processor, an application specific integrated circuit (ASIC), an FPGA (field- Programmable Gate Array) or a combination thereof is applicable.

However, the components of the signal processing device 2 are not limited to those realized by dedicated hardware, and the signal processing device 2 is realized by software, firmware, or a combination of software and firmware. May be
The software or firmware is stored as a program in the memory of the computer. A computer means hardware that executes a program, and corresponds to, for example, a central processing unit (CPU), a central processing unit, a processing unit, an arithmetic unit, a microprocessor, a microcomputer, a processor, a digital signal processor (DSP), etc. .

FIG. 3 is a hardware configuration diagram of a computer when the signal processing device 2 is realized by software or firmware.
When the signal processing apparatus 2 is realized by software or firmware, a program for causing a computer to execute the processing procedure of the null deviation estimation unit 11, the compensation weight calculation unit 12 and the beam forming unit 13 is stored in the memory 31 of the computer. The processor 32 of the computer may execute the program stored in the memory 31.
FIG. 6 is a flowchart showing a processing procedure when the signal processing device 2 is realized by software or firmware.
The memory 31 of the computer is, for example, non-volatile such as random access memory (RAM), read only memory (ROM), flash memory, erasable programmable read only memory (EPROM) or electrically erasable programmable read only memory (EEPROM). Volatile semiconductor memory, magnetic disk, flexible disk, optical disk, compact disk, mini disk, DVD (Digital Versatile Disc), etc. correspond.

FIG. 4 is a block diagram showing the null deviation estimation unit 11 of the adaptive array antenna device according to the first embodiment of the present invention.
In FIG. 4, the first correlation matrix calculating unit 41 is configured to receive the interference wave from the M digital reception signals corresponding to the M signals respectively received by the M subarray antennas 1-m during the listening period. Implement the process of calculating the correlation matrix.

The vector calculation unit 42 scans the scan direction of the interference wave using the weight constraint vector for scanning the main beam direction of the beam and the correlation matrix of the interference wave calculated by the first correlation matrix calculation unit 41. The process of calculating the weight vector to carry out is performed.
That is, the vector calculation unit 42 calculates the diagonal matrix having in the diagonal terms the path difference phase component due to the difference between the main beam direction and the scan direction, the weight constraint vector, and the disturbance calculated by the first correlation matrix calculation unit 41. A process of calculating weight vectors is performed by multiplying the correlation matrix of waves.
The second correlation matrix calculation unit 43 receives the reception signal of the array antenna 1 from M digital reception signals corresponding to the M signals respectively received by the M subarray antennas 1-m during the beam transmission period. The process of calculating the correlation matrix of

The evaluation function calculation unit 44 uses the weight vector calculated by the vector calculation unit 42 and the correlation matrix of the received signal calculated by the second correlation matrix calculation unit 43 to use an evaluation function used for estimation of null deviation. Implement the process to calculate.
The null deviation estimation processing unit 45 uses the evaluation function calculated by the evaluation function calculation unit 44 to carry out processing for estimating null deviation.

FIG. 5 is a block diagram showing the compensation weight calculation unit 12 of the adaptive array antenna device according to the first embodiment of the present invention.
In FIG. 5, the CMT matrix calculation unit 51 calculates a CMT (Covariance Matrix Taper) matrix, which is a matrix for setting a null width, from the null deviation estimated by the null deviation estimation processing unit 45 of the null deviation estimation unit 11. Conduct.
The weight calculation processing unit 52 includes a compensation type correlation matrix calculation unit 53 and a compensation type weight calculation unit 54.
The weight calculation processing unit 52 compensates for null deviation from the CMT matrix calculated by the CMT matrix calculation unit 51 and the correlation matrix of the disturbance wave calculated by the first correlation matrix calculation unit 41 of the null deviation estimation unit 11. The process of calculating the compensation weight is performed.

The compensation type correlation matrix calculation unit 53 performs null shift from the CMT matrix calculated by the CMT matrix calculation unit 51 and the correlation matrix of the disturbance wave calculated by the first correlation matrix calculation unit 41 of the null shift estimation unit 11. A process of calculating a compensation type correlation matrix is performed.
The compensation type weight calculation unit 54 performs processing of calculating a compensation weight for compensating for null deviation from the weight constraint vector and the null deviation compensation type correlation matrix calculated by the compensation type correlation matrix calculation unit 53.

Next, the operation will be described.
First, the signal processing device 2 temporarily stops the beam transmitted from the array antenna 1, and during a listening period in which no beam is transmitted, M sub-array antennas 1-m (m = 1, 2,. , M) acquire M signals respectively received as interference signal.
The signal processing device 2 detects M signals respectively received by the M subarray antennas 1-m, and converts each of the detected M signals from an analog signal to a digital signal.
The null shift estimation unit 11 and the beam forming unit 13 of the signal processing device 2 are provided with a received signal vector including digital received signals which are M converted digital signals.

式（１）において、ｔは時刻、ａ_Ｊ（ｕ_０ ^（ｋ））は、第ｋ番目の妨害波の到来方向ｕ_０ ^（ｋ）＝［ｕ_０ ^（ｋ），ｖ_０ ^（ｋ）］^Ｔに対するステアリングベクトルである。Ｔは転置を示す記号である。
ｊ_ｋ（ｔ）は、第ｋ番目の妨害波の複素振幅、ｎ_０（ｔ）は、受信機雑音ベクトルである。During the listening period, the received signal vector x ₀ (t) shown in the following equation (1) is given to the null deviation estimation unit 11 and the beam forming unit 13.

In equation (1), t is time, a _J (u ₀ ^(k) ) is the direction of arrival of the k th disturbance wave u ₀ ^(k) = [u ₀ ^(k) , v ₀ ^(k) ] ^T Is the steering vector for T is a symbol indicating transposition.
j _k (t) is the complex amplitude of the k th disturbance, and n ₀ (t) is the receiver noise vector.

式（２）において、Ｅ［・］は、・に関するアンサンブル平均を示す記号である。Ｈは、エルミート転置を示す記号である。
式（２）では、リスニング期間における異なる時刻ｔの有限個数の受信信号ベクトルｘ_０（ｔ）を用いている。
第１の相関行列算出部４１は、算出した妨害波の相関行列Ｒ_０をベクトル算出部４２及び補償ウェイト算出部１２の補償型相関行列算出部５３に出力する。When the reception signal vector x ₀ (t) in the listening period is given, the first correlation matrix calculation unit 41 of the null shift estimation unit 11 receives the reception signal in the listening period as shown in the following equation (2) From the vector x ₀ (t), the correlation matrix R ₀ of the interference wave is calculated (step ST1 in FIG. 6).

In Equation (2), E [·] is a symbol indicating an ensemble average for ·. H is a symbol indicating Hermite displacement.
Equation (2) uses a finite number of received signal vectors x ₀ (t) at different times t in the listening period.
The first correlation matrix calculating unit 41 outputs the calculated correlation matrix R ₀ of the interference wave to the vector calculating unit 42 and the compensation type correlation matrix calculating unit 53 of the compensation weight calculating unit 12.

The vector calculation unit 42 obtains the scan direction u of the disturbance wave, and obtains a diagonal matrix D (u-u _s ) having a path difference phase component in a diagonal term by the difference between the scan direction u and the main beam direction u _s .
The main beam direction u _s is in the beam transmission period after the listening period ends, the direction of the main beam in the beam transmitted from the array antenna 1.
Furthermore, the vector calculation unit 42, obtained during beam transmission period after the listening period has ended, the wait constraint vector a to scan the main beam direction u _s of the beam transmitted from the array antenna 1 (u _s) Do.

式（３）において、αは、事前に設定された規格化係数である。
ベクトル算出部４２は、算出したウェイトベクトルｗ（ｕ）を評価関数算出部４４に出力する。
なお、ベクトル算出部４２により算出されるウェイトベクトルｗ（ｕ）は、妨害波の到来方向ｕ_０ ^（ｋ）にヌルを形成するためのベクトルであり、メインビーム方向ｕ_ｓにおけるウェイトベクトルｗ（ｕ_ｓ）＝αＲ_０ ^−１ａ（ｕ_ｓ）を妨害波のスキャン方向ｕに走査するウェイトベクトルに相当する。The vector calculation unit 42 outputs the diagonal matrix D (u−u _s ), the weight constraint vector a (u _s ), and the first correlation matrix calculation unit 41 as shown in the following equation (3). The weight vector w (u) for scanning the scan direction u of the interference wave is calculated by multiplying the interference matrix R ₀ of the interference wave (step ST2 in FIG. 6).

In equation (3), α is a preset normalization factor.
The vector calculation unit 42 outputs the calculated weight vector w (u) to the evaluation function calculation unit 44.
Incidentally, the weight vector w to be calculated by the vector calculating unit 42 (u) is a vector to form a null in the arrival direction u ₀ of the disturbance ^_(k), the weight vector w (u in the main beam direction u _{s s)} = _{αR 0} ^-1 a a _{(u s)} corresponding to the weight vector for scanning the scanning direction u of the disturbance.

式（４）において、ｓ（ｔ）は目標信号、ａ_ｓは、目標信号ｓ（ｔ）の到来方向に対するステアリングベクトル、ｎ（ｔ）は、受信機雑音ベクトルであり、受信機雑音ベクトルｎ（ｔ）の特性は、リスニング期間における受信機雑音ベクトルｎ_０（ｔ）の特性と同じである。
ｕ_０ ^（ｋ）＋δｕ_ｋは、ビーム送信期間における第ｋ番目の妨害波の到来方向であり、リスニング期間における第ｋ番目の妨害波の到来方向とずれている。
δｕ_ｋ＝［δｕ_ｋ，δｖ_ｋ］^Ｔは、ビーム送信期間における第ｋ番目の妨害波の到来方向と、リスニング期間における第ｋ番目の妨害波の到来方向とのずれである。During the beam transmission period after the end of the listening period, the received signal vector x (t) shown in the following equation (4) is given to the null deviation estimation unit 11 and the beam forming unit 13.

In equation (4), s (t) is a target signal, a _s is a steering vector for the direction of arrival of the target signal s (t), n (t) is a receiver noise vector, and a receiver noise vector n (t) The characteristics of t) are the same as the characteristics of the receiver noise vector n ₀ (t) in the listening period.
u ₀ ^(k) + δ u _k is the arrival direction of the kth disturbance in the beam transmission period, which is offset from the arrival direction of the kth interference in the listening period.
δ u _k = [δ u _k , δ v _k ] ^T is the deviation between the arrival direction of the k th disturbance in the beam transmission period and the arrival direction of the k th interference in the listening period.

第２の相関行列算出部４３は、算出した受信信号の相関行列Ｒ_ｘを評価関数算出部４４に出力する。The second correlation matrix calculation unit 43 of the null shift estimation unit 11 receives the reception signal vector x (t) in the beam transmission period after the end of the listening period, as shown in the following equation (5) , from the received signal vector x in the beam transmission period (t), calculates a correlation matrix R _x of the received signal of the array antenna 1 (step ST3 in FIG. 6).

Second correlation matrix calculator 43 outputs the correlation matrix R _x of the calculated reception signal to the evaluation function calculating unit 44.

式（６）の分母であるｗ^Ｈ（ｕ）Ｒ_ｘｗ（ｕ）は、ウェイトベクトルｗ（ｕ）によるビームの走査出力電力である。
ウェイトベクトルｗ（ｕ）は、上述したように、メインビーム方向ｕ_ｓにおけるウェイトベクトルｗ（ｕ_ｓ）を妨害波のスキャン方向ｕに走査するためのウェイトベクトルであるため、ウェイトベクトルｗ（ｕ）により形成される妨害波の到来方向ｕに対するヌルもスキャンされる。The evaluation function calculation unit 44 of the null shift estimation unit 11 calculates the weight vector w (u) output from the vector calculation unit 42 and the correlation matrix R _x of the received signal output from the second correlation matrix calculation unit 43. As shown in the following equation (6), the evaluation function P (u) used for estimation of the null deviation is calculated (step ST4 in FIG. 6).

The denominator w ^H (u) R _x w (u) of equation (6) is the scan output power of the beam according to the weight vector w (u).
Weight vector w (u), as described above, since a weight vector for scanning the weight vector in the main beam direction u _s w a (u _s) in the scanning direction u of the disturbance, weight vector w (u) Nulls are also scanned for the arrival direction u of the disturbance formed by.

Therefore, the weight vector w (u), the scanning direction u is if _{u s} + .delta.u _k, to form a null in the direction of _{^{u 0 (k) + δu k}} .
This null the formation direction _u ^{0 _(k)} + _δu ^k coincides with the arrival direction _u ^{0 _(k)} + _δu ^k of the k-th interference wave contained in the correlation matrix _{R x} of the received signal, ^{w H} The power of the k-th disturbance contained in (u _s + δu _k ) R _x w (u _s + δu _k ) is minimized.
Therefore, the evaluation function calculating unit 44 the evaluation function P calculated by (u), in the scanning direction u = u s ₊ .delta.u _k, a peak is a function value, .delta.u _k is null and u corresponding to the function value of the peak It becomes a candidate for deviation.
Since a total of K null shift candidates exist, among the K candidates, the candidate with the largest function value is the null shift estimated value δu hat. In the text of the specification, for convenience of the electronic application, since it is not possible to put the symbol of "^" on the letter, it is written as "δu hat".
The evaluation function calculation unit 44 outputs the calculated evaluation function P (u) to the null deviation estimation processing unit 45.

  The null deviation estimation processing unit 45 of the null deviation estimation unit 11 estimates the null deviation using the evaluation function P (u) output from the evaluation function calculation unit 44, and calculates the compensation weight of the estimated value δu of the null deviation. It outputs to the part 12 (step ST5 of FIG. 6).

式（７）において、Ｄ_Δは、差ビーム形成用のテーパを対角項に有する対角行列である。
ｗ（ｕ）は、妨害波の到来方向ｕ及びメインビーム方向ｕ_ｓにヌルを形成するウェイトベクトルである。
メインビーム方向ｕ_ｓにヌルを形成するため、メインビーム方向ｕ_ｓの近傍から到来する目標信号は抑圧され、ｗ^Ｈ（ｕ_ｓ＋δｕ_ｋ）Ｒ_ｘｗ（ｕ_ｓ＋δｕ_ｋ）に含まれている目標信号の電力が低減される。Here, the power of the k-th disturbance is minimized, but the power of the target signal and the power of the disturbances other than the k-th included in the correlation matrix R _x of the received signal still remain .
When calculating the estimated value .delta.u hat null ^shift, power _{w H (u s + δu k} ) R x w (u s + δu k) to Including target signal, because it is unnecessary power, advance the target signal If the power of V can be reduced, the estimation accuracy of the null deviation using the evaluation function P (u) is improved.
Therefore, the vector calculation unit 42 may calculate the weight vector w (u) for scanning the scan direction u of the interference wave by the following equation (7) instead of the equation (3).

In equation (7), D _Δ is a diagonal matrix having tapers for difference beam formation in diagonal terms.
w (u) is a weight vector that forms nulls in the arrival direction u of the disturbance wave and the main beam direction u _s .
To form a null in the main beam direction _{u s,} target signal coming from the vicinity of the main beam direction _{u s} is ^suppressed, it is included in _{w H (u s + δu k} ) R x w (u s + δu k) The power of the target signal is reduced.

In the description up to this point, and the direction of arrival of the k-th disturbance in the listening period, the deviation between the arrival direction of the k-th interference wave in the beam transmission period _{δu k = [δu k, δv} k] and ^T However, if the difference in the deviation due to the K interference waves is so small as to be negligible, the deviation may be treated as δu = [δu, δv] ^T.
In this case, the evaluation function P (u), in the scanning direction u = u s ₊ .delta.u, exhibits a peak function value, .delta.u a u corresponding to the function value of the peak is a candidate for null shift, the peak of the function values , Which corresponds to u, becomes the estimated value of null shift δu hat.

式（８）において、ｉ，ｊは行列要素番号であり、ｉ＝１，２，・・・，Ｍ、ｊ＝１，２，・・・，Ｍである。
式（８）では、第ｍ番目のサブアレーアンテナ１−ｍの座標をＰ_ｍ＝［ｘ_ｋ，ｙ_ｋ］^Ｔとし、Δｘ_ｉｊ及びΔｙ_ｉｊは、それぞれ相対座標である。
Δｘ_ｉｊ＝ｘ_ｉ−ｘ_ｊであり、Δｙ_ｉｊ＝ｙ_ｉ−ｙ_ｊである。λは送信波長である。
式（８）では、ＣＭＴ行列Ｔ_ＣＭＴによる設定ヌル幅をΔｕ＝［Δｕ，Δｖ］^Ｔとしている。
ＣＭＴ行列Ｔ_ＣＭＴによる設定ヌル幅Δｕは、ヌルずれの推定値δｕハットに基づいて、以下の式（９）のように決定される。

式（９）において、ｋ_ｕ，ｋ_ｖは、任意の係数であり、ヌルずれの推定値δｕハットと関係なく、ビーム形成で用いるテーパに応じて事前に設定される値である。
ＣＭＴ行列算出部５１は、算出したＣＭＴ行列Ｔ_ＣＭＴを補償型相関行列算出部５３に出力する。When the CMT matrix calculation unit 51 of the compensation weight calculation unit 12 receives the estimated value δu of null deviation output from the null deviation estimation processing unit 45 of the null deviation estimation unit 11, as shown in the following equation (8) The CMT matrix T _CMT for setting the null width is calculated using the estimated value δu of the null shift (step ST6 in FIG. 6).

In equation (8), i and j are matrix element numbers, and i = 1, 2,..., M and j = 1, 2,.
In equation (8), the coordinates of the m-th subarray antenna 1-m are P _m = [x _k , y _k ] ^T, and Δx _ij and Δy _ij are relative coordinates.
Δx _ij = x _i −x _j and Δy _ij = y _i −y _j . λ is a transmission wavelength.
In the equation (8), the set null width by the CMT matrix T _CMT is Δu = [Δu, Δv] ^T.
The set null width Δu by the CMT matrix T _CMT is determined as in the following equation (9) based on the estimated value δu of the null deviation.

In equation (9), k _u and k _v are arbitrary coefficients and are values that are set in advance according to the taper used in beam formation, regardless of the estimated value of the null shift δu hat.
The CMT matrix calculation unit 51 outputs the calculated CMT matrix T _CMT to the compensation type correlation matrix calculation unit 53.

式（１０）において、二重丸の記号は、ＣＭＴ行列Ｔ_ＣＭＴと妨害波の相関行列Ｒ_０との行列要素同士の乗算を行うアダマール積を示す記号である。
補償型相関行列算出部５３は、算出したヌルずれ補償型相関行列Ｒを補償型ウェイト算出部５４に出力する。The compensation type correlation matrix calculation unit 53 of the compensation weight calculation unit 12 calculates the CMT matrix T _CMT output from the CMT matrix calculation unit 51 and the first of the null shift estimation unit 11 as shown in the following equation (10). From the correlation matrix R ₀ of the interference wave output from the correlation matrix calculation unit 41, the null shift compensation type correlation matrix R is calculated (step ST7 in FIG. 6).

In Equation (10), the double circle symbol is a symbol indicating a Hadamard product in which the matrix elements of the CMT matrix T _CMT and the correlation matrix R ₀ of the disturbance wave are multiplied.
The compensation type correlation matrix calculation unit 53 outputs the calculated null deviation compensation type correlation matrix R to the compensation type weight calculation unit 54.

式（１１）において、βは規格化係数である。
補償型ウェイト算出部５４は、算出した補償ウェイトベクトルｗ_Ａをビーム形成部１３に出力する。The compensation weight calculator 54 of the compensation weight calculator 12 is a weight constraint vector a for scanning the main beam direction u _s of the beam transmitted from the array antenna 1 during the beam transmission period after the end of the listening period. to get the _{(u s).}
The compensation type weight calculation unit 54 uses the weight constraint vector a (u _s ) and the null shift compensation type correlation matrix R output from the compensation type correlation matrix calculation unit 53 as shown in the following equation (11) _A compensation weight vector w _A indicating a compensation weight for compensating for the null shift is calculated (step ST8 in FIG. 6).

In equation (11), β is a normalization factor.
The compensation weight calculating unit 54 outputs the calculated compensation weight vector w _A to the beam forming unit 13.

When the received signal vector x (t) in the beam transmission period after the end of the listening period is given, the beam forming unit 13 compensates the received signal vector x (t) as shown in the following equation (12) The reception beam y (t) is calculated by multiplying the weight vector w _A and combining the reception signal vector x (t) after the compensation weight multiplication (step ST9 in FIG. 6).

That is, the M multipliers 14-m in the beam forming unit 13 are included in the compensation weight vector w _A in the digital received signal related to the sub-array antenna 1-m included in the received signal vector x (t). A compensation weight related to the existing sub array antenna 1-m is multiplied, and the digital reception signal after the compensation weight multiplication is output to the adder 15.
The adder 15 of the beam forming unit 13 combines the M digital reception signals respectively output from the M multipliers 14-m, and outputs the combined digital reception signal as a reception beam y (t).
Since the received signal vector x (t) is multiplied by the compensation weight vector w _A , a null having a setting null width Δu is formed in the received beam y (t) with the arrival direction u ₀ of the interference wave as the center. ing.

  As apparent from the above, according to the first embodiment, the plurality of signals respectively received by the M subarray antennas 1-m during the listening period, and the M subarray antennas during the beam transmission period A null deviation estimation unit 11 is provided for estimating a null deviation which is a change between the arrival direction of the interference wave in the listening period and the arrival direction of the interference wave in the beam transmission period from a plurality of signals respectively received by 1-m. Based on the null shift estimated by the null shift estimation unit 11, the compensation weight calculation unit 12 determines nulls as weighting coefficients for a plurality of signals respectively received by the M sub-array antennas 1-m during the beam transmission period. Since the compensation weight for compensating for the deviation is calculated, even if the arrival direction of the interference wave changes significantly, An effect that can suppress interference waves contained in the signal signal.

Second Embodiment
In the first embodiment, the compensation weight calculation unit 12 includes the CMT matrix calculation unit 51, and uses the CMT matrix T _CMT calculated by the CMT matrix calculation unit 51 to indicate the compensation weight indicating the compensation weight that compensates for the null shift. It shows an example of calculating a vector w _a.
In the second embodiment, an example will be described in which the compensation weight calculation unit 12 calculates a compensation weight vector w _A indicating a compensation weight that compensates for null shift without using the CMT matrix T _CMT .

FIG. 7 is a block diagram showing a compensation weight calculator 12 of an adaptive array antenna apparatus according to a second embodiment of the present invention. In FIG. 7, the same reference numerals as those in FIG. 5 denote the same or corresponding parts, and therefore the description thereof will be omitted.
The weight calculation processing unit 55 includes a compensation type correlation matrix calculation unit 56 and a compensation type weight calculation unit 54.
The compensation type correlation matrix calculation unit 56 calculates the null deviation estimated by the null deviation estimation processing unit 45 of the null deviation estimation unit 11 and the disturbance wave calculated by the first correlation matrix calculation unit 41 of the null deviation estimation unit 11. A process of calculating a null shift compensation type correlation matrix from the correlation matrix is performed.

Next, the operation will be described.
In the first embodiment, the arrival direction of the k-th disturbance in the listening period, the deviation between the arrival direction of the k-th interference wave in the beam transmission period _{δu k = [δu k, δv} k] as ^T There is.
In the second embodiment, it is assumed that the difference in the deviation of the K interference waves is small enough to be ignored, and an example in which the deviation is treated as δu = [δu, δv] ^T will be described.

式（１３）において、ｕ_０ ^（ｋ）＋δｕは、ビーム送信期間における第ｋ番目の妨害波の到来方向であり、リスニング期間における第ｋ番目の妨害波の到来方向とずれている。Therefore, in the second embodiment, during the beam transmission period after the end of the listening period, the received signal vector x (t) shown in the following equation (13) is null deviation estimation unit 11 and beam formation unit 13 Given to

In equation (13), u ₀ ^(k) + δu is the arrival direction of the kth disturbance in the beam transmission period, and is offset from the arrival direction of the kth interference in the listening period.

The second correlation matrix calculator 43 of the null deviation estimator 11 receives the received signal vector x (t) in the beam transmission period shown in equation (13), and transmits the beam as in the first embodiment. from the received signal vector x in period (t), calculates a correlation matrix R _x of the received signal of the array antenna 1.
Second correlation matrix calculator 43 outputs the correlation matrix R _x of the calculated reception signal to the evaluation function calculating unit 44.

The evaluation function calculation unit 44 of the null shift estimation unit 11 calculates the weight vector w (u) output from the vector calculation unit 42 and the correlation matrix R _x of the received signal output from the second correlation matrix calculation unit 43. As in the first embodiment, the evaluation function P (u) used to estimate the null deviation is calculated.
The denominator w ^H (u) R _x w (u) of equation (6) is the scan output power of the beam according to the weight vector w (u).
Weight vector w (u), as described above, since a weight vector for scanning the weight vector in the main beam direction u _s w a (u _s) in the scanning direction u of the disturbance, weight vector w (u) Nulls are also scanned for the arrival direction u of the disturbance formed by.

Therefore, the weight vector w (u) forms a null in the direction of u ₀ ^(k) + δu if the scan direction u is u _s + δu.
This null the formation direction _u ^{0 (k)} + _δu is consistent with the direction of arrival _u ^{0 (k)} + _δu k-th interference wave contained in the correlation matrix _{R x} of the received ^{signal, w} H (u _s + _.delta.u) power R x w _{(u s} + _δu k-th interference wave contained in) is minimized.
Therefore, the evaluation function P calculated by the evaluation function calculating unit 44 (u), in the scanning direction u = u s ₊ .delta.u, a peak is a function value, .delta.u is null shift and u corresponding to the function value of the peak It becomes an estimated value δu hat.
The evaluation function calculation unit 44 outputs the calculated evaluation function P (u) to the null deviation estimation processing unit 45.

  The null deviation estimation processing unit 45 of the null deviation estimation unit 11 estimates the null deviation using the evaluation function P (u) output from the evaluation function calculation unit 44, as in the first embodiment, and performs the null deviation. The estimated value .delta.u hat is output to the compensation weight calculation unit 12.

The compensation type correlation matrix calculation unit 56 of the compensation weight calculation unit 12 outputs the estimated null deviation value δu hat output from the null deviation estimation processing unit 45 of the null deviation estimation unit 11 and the first correlation matrix calculation unit 41. A null deviation compensation type correlation matrix R ₀ ′ is calculated from the detected interference wave correlation matrix R ₀ .
The compensation type correlation matrix calculation unit 56 outputs the calculated null deviation compensation type correlation matrix R ₀ ′ to the compensation type weight calculation unit 54.

式（１４）において、Ａ_０は、Ｋ個のステアリングベクトルａ_Ｊ（ｕ_０ ^（ｋ））を並べた行列、Ｊはｊ_ｋ（ｔ）の相関行列、σ^２は受信機雑音電力である。
一方、ビーム送信期間中の妨害波の相関行列Ｒ_０’は、以下の式（１５）のように表すことができる。ビーム送信期間中の妨害波の相関行列Ｒ_０’は、ヌルずれ補償型相関行列に相当する。

式（１５）において、Ａ_０’は、Ｋ個のステアリングベクトルａ_Ｊ（ｕ_０ ^（ｋ）＋δｕ）を並べた行列である。Here, the correlation matrix R ₀ of the interference wave calculated by the first correlation matrix calculation unit 41 can be expressed as the following equation (14).

In equation (14), A ₀ is a matrix in which K steering vectors a _J (u ₀ ^(k) ) are arranged, J is a correlation matrix of j _k (t), and σ ² is receiver noise power.
On the other hand, the correlation matrix R ₀ ′ of the interference wave during the beam transmission period can be expressed as the following equation (15). The correlation matrix R ₀ ′ of interference waves during the beam transmission period corresponds to the null shift compensation type correlation matrix.

In equation (15), A ₀ ′ is a matrix in which K steering vectors a _J (u ₀ ^(k) + δu) are arranged.

よって、式（１５）は、以下の式（１７）のように表すことができる。

式（１７）の関係より、ヌルずれの推定値δｕハットと、妨害波の相関行列Ｒ_０とから、ヌルずれ補償型相関行列として、以下の式（１８）のようにＲ_０’を算出することができる。

At this time, if a diagonal matrix having a path difference phase component due to the shift δu as a diagonal term is D (δu), the following equation (16) is established.

Therefore, equation (15) can be expressed as equation (17) below.

From the relation of the null shift and the correlation matrix R ₀ of the interference wave from the relationship of the equation (17), R ₀ ′ is calculated as the null shift compensation type correlation matrix as the following equation (18) be able to.

補償型ウェイト算出部５４は、算出した補償ウェイトベクトルｗ_Ａをビーム形成部１３に出力する。The compensation weight calculator 54 of the compensation weight calculator 12 is a weight constraint vector a for scanning the main beam direction u _s of the beam transmitted from the array antenna 1 during the beam transmission period after the end of the listening period. to get the _{(u s).}
The compensation weight calculation unit 54 calculates the weight constraint vector a (u _s ) and the null shift compensation correlation matrix R ₀ ′ output from the compensation correlation matrix calculation unit 53 as shown in the following equation (19). Using this, a compensation weight vector w _A indicating a compensation weight that compensates for the null shift is calculated.

The compensation weight calculating unit 54 outputs the calculated compensation weight vector w _A to the beam forming unit 13.

When receiving signal vector x (t) in the beam transmission period after the end of the listening period, beam forming unit 13 adds the compensation weight vector to reception signal vector x (t) as in the first embodiment. The reception beam y (t) is calculated by multiplying w _A and combining the reception signal vector x (t) after the compensation weight multiplication.
Since the received signal vector x (t) is multiplied by the compensation weight vector w _A , a null having a setting null width Δu is formed in the received beam y (t) with the arrival direction u ₀ of the interference wave as the center. ing.

  As apparent from the above, according to the second embodiment, as in the first embodiment, even if the arrival direction of the interference wave changes significantly, the interference wave included in the received signal of the array antenna 1 Produces an effect of suppressing

  In the scope of the invention, the present invention allows free combination of each embodiment, or modification of any component of each embodiment, or omission of any component in each embodiment. .

  The present invention is suitable for an adaptive array antenna apparatus that multiplies a plurality of signals received by a plurality of subarray antennas by a weighting factor and combines a plurality of signals after weighting factor multiplication.

  Reference Signs List 1 array antenna, 1-1 to 1-M sub array antenna, 2 signal processing device, 11 null deviation estimation unit, 12 compensation weight calculation unit, 13 beam forming unit, 14-1 to 14-M multiplier, 15 adder, 21 null shift estimation circuit, 22 compensation weight calculation circuit, 23 beam forming circuit, 31 memory, 32 processors, 41 first correlation matrix calculation unit, 42 vector calculation unit, 43 second correlation matrix calculation unit, 44 evaluation function calculation 51 CMT matrix calculation unit 52 weight calculation processing unit 53 compensation type correlation matrix calculation unit 54 compensation type weight calculation unit 55 weight calculation processing unit 56 compensation type correlation matrix calculation unit.

Claims (5)

An array antenna in which a plurality of subarray antennas including one or more element antennas are arranged;
Beam transmission which is a period during which a plurality of signals respectively received by the plurality of sub-array antennas and the beam after the end of the listening period are transmitted during a listening period during which a beam is not transmitted from the array antenna During a period, from a plurality of signals respectively received by the plurality of subarray antennas, a null deviation which is a change between an arrival direction of an interference wave in the listening period and an arrival direction of the interference wave in the beam transmission period is estimated. A null deviation estimation unit,
Based on the null deviation estimated by the null deviation estimation unit, a compensation weight for compensating the null deviation is calculated as a weighting factor for a plurality of signals respectively received by the plurality of subarray antennas during the beam transmission period. Compensation weight calculation unit,
A beam forming unit configured to combine a plurality of signals after the compensation weight multiplication by multiplying the plurality of signals respectively received by the plurality of subarray antennas during the beam transmission period by the compensation weight calculated by the compensation weight calculation unit; Adaptive array antenna device with and. The null deviation estimation unit
A first correlation matrix calculation unit that calculates a correlation matrix of the interference wave from a plurality of signals respectively received by the plurality of subarray antennas during the listening period;
Weight for scanning the scan direction of the interference wave using the weight constraint vector for scanning the main beam direction of the beam and the correlation matrix of the interference wave calculated by the first correlation matrix calculation unit A vector calculation unit that calculates a vector;
A second correlation matrix calculation unit that calculates a correlation matrix of reception signals of the array antenna from the plurality of signals respectively received by the plurality of subarray antennas during the beam transmission period;
An evaluation function calculation unit that calculates an evaluation function used to estimate the null using the weight vector calculated by the vector calculation unit and the correlation matrix of the received signal calculated by the second correlation matrix calculation unit; ,
The adaptive array antenna apparatus according to claim 1, further comprising: a null shift estimation processing unit configured to estimate the null shift using the evaluation function calculated by the evaluation function calculation unit.   The vector calculation unit is calculated by the first correlation matrix calculation unit, a diagonal matrix having in a diagonal term a path difference phase component due to a difference between the main beam direction and the scan direction, the weight constraint vector, and the first correlation matrix calculation unit. The adaptive array antenna apparatus according to claim 2, wherein the weight vector is calculated by multiplying the interference vector by a correlation matrix of the interference wave. The compensation weight calculation unit
A CMT matrix calculation unit that calculates a CMT matrix that is a matrix for setting a null width from the null deviation estimated by the null deviation estimation processing unit;
And a weight calculation processing unit that calculates a compensation weight for compensating for the null deviation from the CMT matrix calculated by the CMT matrix calculation unit and the correlation matrix of the interference wave calculated by the first correlation matrix calculation unit. The adaptive array antenna apparatus according to claim 2, characterized in that: The compensation weight calculation unit
It has a weight calculation processing unit that calculates a compensation weight for compensating for the null deviation from the null deviation estimated by the null deviation estimation processing unit and the correlation matrix of the interference wave calculated by the first correlation matrix calculation unit. The adaptive array antenna apparatus according to claim 2, characterized in that:

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