JP6164936B2 - Radar equipment - Google Patents

Description

  The present invention relates to a radar apparatus that enhances a target detection distance and distance resolution by pulse-compressing a received signal.

In general, a radar apparatus emits a pulse signal (transmission pulse) subjected to frequency modulation or phase modulation to a space, and receives a scattered wave of the pulse signal that has been reflected back by a target or clutter.
When the radar apparatus receives the scattered wave of the pulse signal reflected and returned by the target or the clutter, the radar apparatus performs pulse compression on the received signal using a reference signal having a high correlation with the transmission pulse.
By performing pulse compression, it is possible to increase the transmission energy by increasing the transmission pulse width and to narrow the pulse width, so that the target detection distance and the distance resolution of the radar apparatus can be increased at the same time.

By performing pulse compression, a main lobe is formed at a target delay time, and a range side lobe is formed in a range corresponding to the transmission pulse width before and after the main lobe.
This range side lobe may degrade other target signal-to-interference ratios (SIRs) present in the same region. Here, Signal is another target of the range side lobe region, and Interference is a target in the main lobe generating the range side lobe.
As such a scenario, for example, there is a case where it is necessary to detect a new target with a relatively small received power separated from a target that is already being tracked at an early stage. In this case, the new target has a problem that the SIR is lowered due to interference by the target range side lobe and detection becomes difficult.
In addition, in the target detection scenario under the clutter environment, there is a problem that the target detection becomes difficult due to the low SIR due to the interference by the range side lobe of the ground clutter existing around the target.

In order to solve the above problem, there is a radar apparatus that reduces a range side lobe by using a window function in pulse compression using an LFM (Linear Frequency Modulation) wave (see, for example, Non-Patent Document 1). .
However, in the case of using the window function, in principle, a post-compression pulse width that deteriorates the distance resolution and a mismatch loss that reduces the detection distance occur.
In other words, pulse compression using a window function can achieve low-range sidelobe characteristics, but it is impossible to avoid an increase in pulse width and mismatch loss after compression, resulting in degradation of distance resolution and degeneration of detection distance. Occur.

The following Patent Document 1 discloses a radar apparatus that suppresses the occurrence of degradation in distance resolution and degeneration of detection distance.
In this radar apparatus, the target and clutter are detected from the amplitude detection waveform of the received signal, and the distance to the target and clutter is extracted.
Next, this radar apparatus gives a new pulse compression weight based on the MSN (Maximum Signal to Noise Ratio) method to the range bin where the target exists, and performs pulse compression related to the range bin. At this time, by giving a pulse compression weight based on the MSN method or the like that forms a null based on the distance to the previously extracted clutter, the range sidelobe interference caused by the clutter is suppressed, and the SIR is improved. Yes.

As a result, only the range side lobe in which clutter is present is selectively reduced, so that it is expected that expansion of the compressed pulse width and mismatch loss can be suppressed as compared with the case of using the window function.
However, in this radar device, it is necessary to obtain the distance to the clutter as an interference source from the amplitude detection waveform of the received signal in advance. For this reason, the pulse compression weight cannot be obtained before obtaining the amplitude detection waveform of the received signal. Further, the interference source is limited to the clutter, and there is a problem that it cannot be directly applied to the interference of the range side lobe due to the existing target.

The radar apparatus disclosed in Patent Document 2 below performs conventional pulse compression, selects an interference wave based on an amplitude value, and determines the pulse of the interference wave from the distance and amplitude value of the interference wave. Estimate the waveform before compression.
Next, the radar apparatus suppresses the interference wave by subtracting the pre-pulse compression waveform of the interference wave from the reception signal before the pulse compression, and again performs the conventional pulse compression to suppress the interference wave. A pulse compression output is obtained.
In this radar apparatus, since the interference wave is selected based on the amplitude value, only the range side lobe where the clutter, the existing target, etc. are present is selectively reduced. For this reason, compared with the case where a window function is used, it is expected that expansion of the pulse width after compression and mismatch loss can be suppressed.
However, in this radar apparatus, when the amplitude value is obtained after integrating the pulse compression output for each of the plurality of PRIs, the amplitude value before integration is estimated from the integrated amplitude value due to the stochastic fluctuation of the amplitude value between pulses. There is a problem that becomes difficult.

In order to avoid this problem, there is a method of directly obtaining the amplitude value before integration, but the calculation scale increases, and the interference wave before integration has a small amplitude value, and the distance of the interference wave and the estimation accuracy of the amplitude value are reduced. There is a problem of deterioration.
Similarly to the radar apparatus disclosed in Patent Document 1, it is necessary to obtain the distance to the interference source from the amplitude detection waveform of the received signal in advance, and before acquiring the amplitude detection waveform of the received signal, the pulse The compression weight cannot be obtained.

Japanese Patent Laying-Open No. 2005-201727 (paragraph number [0004]) JP 10-197624 A (paragraph number [0008])

M.A.Richards, J.A.Scheer, W.A.Holms, Principles of Modern Radar Basic Principles, Scitech Publishing Inc., Raleigh, NC, 2010.

Since the conventional radar apparatus is configured as described above, when applying a pulse compression weight based on the MSN method or the like that forms a null based on the distance to the previously extracted clutter (Patent Document 1), a window Compared to the case where the function is used, the pulse width after compression and mismatch loss can be reduced, but the interference source is limited to clutter. There were issues that could not be applied.
In addition, the interference wave is selected based on the amplitude value, the pre-pulse compression waveform of the interference wave is estimated from the distance and amplitude value of the interference wave, and the pre-pulse compression waveform of the interference wave from the received signal before the pulse compression. When the interference wave is suppressed by subtracting (Patent Document 2), it is possible to suppress the expansion of the compressed pulse width and the mismatch loss as compared with the case of using the window function. However, when the amplitude value is obtained after integrating the pulse compression output for each of the plurality of PRIs, it is difficult to estimate the amplitude value before integration from the integrated amplitude value due to the stochastic fluctuation of the amplitude value between pulses. Sometimes. On the other hand, when the amplitude value before integration is directly obtained, the calculation scale increases, and the amplitude value of the interference wave before integration is small, so that the estimation accuracy of the distance and amplitude value of the interference wave may deteriorate. For this reason, there has been a problem that interference waves cannot be sufficiently suppressed.

  The present invention has been made to solve the above-described problems, and an object of the present invention is to obtain a radar apparatus that can sufficiently suppress the interference of range side lobes due to clutter and existing targets.

The radar apparatus according to the present invention uses a predetermined frequency signal to generate pulse signal generation means for generating a pulse signal, and radiates the pulse signal generated by the pulse signal generation means toward space in a predetermined beam pointing direction. On the other hand, a transmission / reception unit that receives the scattered wave of the pulse signal that has been reflected back by the target or the clutter is provided, and the pulse compression unit performs pulse compression on the reception signal of the transmission / reception unit, and provides a known interference source. The range side of the distance where the main lobe of the scattered wave reflected by the interference source appears among the range side lobes of the received signal formed by performing pulse compression using the scattered wave information on the reflected scattered wave. The lobe is suppressed.
The pulse compression means has first to Kth pulse compression units, and the kth (k = 1, 2,..., K) pulse compression units are included in the scattered wave information. Among the information on the first to K-th scattered waves reflected by the first to K-th interference sources, the information on the scattered waves other than the k-th scattered wave is used as the interference wave information, and the interference wave is obtained from the interference wave information. An interference wave suppression matrix for suppression is calculated, and using the interference wave suppression matrix, scattered waves other than the k-th scattered wave included in the received signal are suppressed as interference waves .

  According to the present invention, the pulse compression unit performs pulse compression on the reception signal of the transmission / reception unit and performs pulse compression using the scattered wave information regarding the scattered wave reflected by the known interference source. Since the range side lobe of the distance where the main lobe of the scattered wave reflected by the interference source appears is suppressed among the range side lobes of the received signal, the interference of the range side lobe due to clutter and the existing target is suppressed. There is an effect that can be sufficiently suppressed.

It is a block diagram which shows the radar apparatus by Embodiment 1 of this invention. It is a block diagram which shows KA type pulse compression part 10-k (k = 1, 2, ..., K) of the radar apparatus by Embodiment 1 of this invention.

Embodiment 1 FIG.
1 is a block diagram showing a radar apparatus according to Embodiment 1 of the present invention.
In FIG. 1, a transmitter 1 generates a predetermined frequency signal, and generates a pulse signal having a predetermined pulse width by performing modulation using the frequency signal in a predetermined method, and the pulse signal is converted to a duplexer 2. Execute the process to output to. The transmitter 1 constitutes pulse signal generation means.
The duplexer 2 outputs the pulse signal output from the transmitter 1 to the antenna 3, while executing the process of outputting the scattered wave incident from the antenna 3 to the receiver 3.
The antenna 3 radiates the pulse signal output from the duplexer 2 toward the space in a predetermined beam directing direction, and receives the scattered wave of the pulse signal that has been reflected back by the target or clutter.

The receiver 4 performs predetermined reception processing such as detection processing and demodulation processing on the signal received by the antenna 3, and converts the signal after reception processing to a baseband frequency to generate an analog reception signal Perform the process.
The AD converter 5 performs A / D conversion on the analog reception signal generated by the receiver 4 and outputs the digital reception signal to the pulse compressor 6.
The duplexer 2, the antenna 3, the receiver 4, and the AD converter 5 constitute transmission / reception means.

  The pulse compressor 6 is composed of, for example, a semiconductor integrated circuit on which a CPU is mounted, a one-chip microcomputer, or the like, and performs pulse compression on the digital reception signal output from the AD converter 5 and also known interference. The main lobe of the scattered wave reflected by the interference source appears among the range side lobes of the digital reception signal formed by performing the pulse compression using the scattered wave information regarding the scattered wave reflected by the source. Performs processing to suppress the range side lobe of the distance. The pulse compressor 6 constitutes pulse compression means.

The discrete Fourier transform unit 7 of the pulse compressor 6 performs a process of calculating a frequency spectrum of the digital reception signal by performing a discrete Fourier transform on the digital reception signal output from the AD converter 5. The discrete Fourier transform unit 7 constitutes a first frequency spectrum calculation unit.
The discrete Fourier transform unit 8 inputs a pulse signal generated by the transmitter 1 as a reference signal, and performs a process of calculating a frequency spectrum of the reference signal by performing a discrete Fourier transform on the reference signal. The discrete Fourier transform unit 8 constitutes second frequency spectrum calculation means.

The spectrum multiplier 9 multiplies the frequency spectrum calculated by the discrete Fourier transform unit 7 and the frequency spectrum calculated by the discrete Fourier transform unit 8 to perform a process of calculating a received signal vector after spectrum multiplication. The spectrum multiplier 9 constitutes a spectrum multiplier.
The KA type pulse compression units 10-1 to 10-K perform the pulse compression on the reception signal vector calculated by the spectrum multiplication unit 9, and the pulse compression is performed by using the scattered wave information. Among the range side lobes of the digital reception signal, processing is performed to suppress the range side lobes at the distance at which the main lobe of the scattered wave reflected by the interference source appears. The KA type pulse compression units 10-1 to 10-K constitute first to Kth KA type pulse compression means.

In the example of FIG. 1, it is assumed that the pulse compressor 6 of the radar apparatus is configured by dedicated hardware, but the pulse compressor 6 may be configured by a computer.
When the pulse compressor 6 is configured by a computer, the discrete Fourier transform unit 7, the discrete Fourier transform unit 8, the spectrum multiplication unit 9, and the KA type pulse compression units 10-1 to 10-which are components of the pulse compressor 6. A program describing the processing contents of K may be stored in a memory of a computer so that the CPU of the computer executes the program stored in the memory.

FIG. 2 is a block diagram showing the KA type pulse compression unit 10-k (k = 1, 2,..., K) of the radar apparatus according to Embodiment 1 of the present invention.
In FIG. 2, the sidelobe interference wave information extraction unit 11 relates to a scattered wave other than the scattered wave reflected by the kth interference source among the information about the first to Kth scattered waves included in the scattered wave information. A process of extracting information as k-th interference wave information is performed.
The interference wave steering vector calculation unit 12 performs a process of calculating the kth interference wave steering vector using the kth interference wave information extracted by the side lobe interference wave information extraction unit 11.

The interference wave correlation matrix calculation unit 13 calculates the k th interference wave information from the k th interference wave information extracted by the side lobe interference wave information extraction unit 11 and the k th interference wave steering vector calculated by the interference wave steering vector calculation unit 12. A process for calculating an interference wave correlation matrix is performed.
The interference wave suppression matrix calculation unit 14 performs a process of calculating the kth interference wave suppression matrix from the kth interference wave correlation matrix calculated by the interference wave correlation matrix calculation unit 13.

The interference wave suppression unit 15 multiplies the reception signal vector calculated by the spectrum multiplication unit 9 by the k-th interference wave suppression matrix calculated by the interference wave suppression matrix calculation unit 14 so as to be included in the reception signal vector. Processing for suppressing scattered waves other than the k-th scattered wave as interference waves is performed.
The post-pulse compression signal calculation unit 16 performs a process of calculating a reception signal vector after pulse compression from the reception signal vector after interference wave suppression by the interference wave suppression unit 15.

Next, the operation will be described.
The transmitter 1 generates a predetermined frequency signal, and performs modulation (for example, frequency modulation) using the frequency signal in a predetermined method, so that a pulse width τ at a predetermined PRI (Pulse Repetition Interval) interval. _{A TX} pulse signal r (t _m ) is generated, and the pulse signal r (t _m ) is output to the duplexer 2.
Here, the center frequency _{f c} of the pulse signal r _{(t m),} the modulation bandwidth of the pulse signal r _{(t m)} and B.

When receiving the pulse signal r (t _m ) from the transmitter 1, the duplexer 2 outputs the pulse signal r (t _m ) to the antenna 3.
Thereby, the pulse signal r (t _m ) is radiated from the antenna 3 to the space in a predetermined beam directing direction.
A part of the pulse signal r (t _m ) radiated into the space is reflected by the target or the clutter, and the scattered wave of the pulse signal is received by the antenna 3.

式（１）（２）において、ｔ_ｍは送信開始時刻を基準とする受信ゲート内の第ｍ番目の時間サンプルである。即ち、ｔ_１＝τ_ＴＸであり、送受切替時間は無視する。
時間サンプルのサンプリング時間間隔はＡＤサンプリング間隔でありΔｔとする。また、α_ｋは距離等による減衰係数、ｎ（ｔ_ｍ）は受信機雑音である。
この実施の形態１では、１ＰＲＩ中の受信ゲート内の受信信号を考えるものとする。 The receiver 4 performs predetermined reception processing such as detection processing and demodulation processing on the signal received by the antenna 3 and converts the signal after reception processing to a baseband frequency to generate an analog reception signal. To do.
When the receiver 4 generates an analog reception signal, the AD converter 5 performs A / D conversion on the analog reception signal and outputs a digital reception signal z (t _m ) to the pulse compressor 6.
Here, the digital received signal z (t _m ) is a K-wave scattered wave with a delay time t _k ^(s) and a Doppler frequency f _k ^(D) with respect to the pulse signal r (t _m ) radiated from the antenna 3. And is expressed as the following formula (2).

In equations (1) and (2), t _m is the mth time sample in the reception gate with reference to the transmission start time. That is, t ₁ = τ _TX and the transmission / reception switching time is ignored.
The sampling time interval of the time sample is an AD sampling interval and is assumed to be Δt. Further, α _k is an attenuation coefficient due to distance or the like, and n (t _m ) is receiver noise.
In the first embodiment, a received signal in a receiving gate in 1 PRI is considered.

When the discrete Fourier transform unit 7 receives the digital reception signal z (t _m ) from the AD converter 5, the discrete Fourier transform unit 7 performs discrete Fourier transform on the digital reception signal z (t _m ), as shown in the following formula (3). to, to calculate the frequency spectrum Z _{(f m)} of the digital received signal z _{(t m).}

ただし、第ｍ（ｍ＝１，・・・，Ｍ）番目の周波数ｆ_ｍは、下記の式（５）のように与えるものとする。

式（５）において、Ｂはリファレンス信号の帯域幅、Δｆはサンプル周波数間隔であり、Ｂ≦ＭΔｆを満たすものとする。 Hereinafter, the processing content of the discrete Fourier transform part 7 is demonstrated concretely.
The total sample point after zero padding of the digital reception signal z (t _m ) in the reception gate is set to M point, and the reception signal before spectrum multiplication composed of M points as shown in the following equation (4) A vector z (output of the discrete Fourier transform unit 7) is defined.

However, the m-th (m = 1,..., M) -th frequency f _m is given by the following equation (5).

In Equation (5), B is the bandwidth of the reference signal, Δf is the sampling frequency interval, and B ≦ MΔf is satisfied.

式（６）において、ｒ_ｋは下記の式（７）のように第ｋ番目の散乱波の周波数スペクトルが並べられたスペクトル乗算前散乱波ベクトル、Ｒ^{（ｓｉｇ）}は下記の式（８）のようにＫ個のスペクトル乗算前散乱波ベクトルｒ_ｋが並べられた行列である。
また、ｓは下記の式（９）のような複素振幅ベクトル、ｎ_０は下記の式（１０）のような受信機雑音の周波数スペクトルが並べられたベクトルである。 By substituting equation (3) into equation (4), the following equation (6) can be transformed.

In the formula (6), r _k is the k-th spectral multiplication before the scattered wave vector frequency spectrum is arranged in the scattered wave as follows equation (7), R ^(sig) is the following formula (8) is the K spectrum multiplied before the scattered wave vector r _k is arranged matrix as.
Further, s is a complex amplitude vector as shown in the following equation (9), and n ₀ is a vector in which the frequency spectrum of the receiver noise as shown in the following equation (10) is arranged.

That is, the discrete Fourier transform unit 7 performs a discrete Fourier transform on the digital received signal z (t _m ) output from the AD converter 5 and obtains a received signal vector z before spectral multiplication represented by Expression (6). I do.

The discrete Fourier transform unit 8 receives the pulse signal r (t _m ) generated by the transmitter 1 as a reference signal, and performs a discrete Fourier transform on the reference signal r (t _m ), whereby the reference signal r (t _m calculates the frequency spectrum R _{(f m)} of _m).
When the discrete Fourier transform unit 8 calculates the frequency spectrum R (f _m ) of the reference signal r (t _m ), the following formula (11) in which complex conjugates of the frequency spectrum R (f _m ) are arranged in a diagonal term: A matrix R _ref as shown in FIG.

式（１２）において、Ａは下記の式（１３）で表され、ｎは下記の式（１４）で表される。

When the discrete Fourier transform unit 7 obtains the received signal vector z before the spectrum multiplication and receives the matrix R _ref from the discrete Fourier transform unit 8, the spectrum multiplying unit 9 receives the received signal as shown in the following equation (12). By calculating the product of the vector z and the matrix R _ref , the received signal vector x after spectral multiplication is calculated.

In the formula (12), A is represented by the following formula (13), and n is represented by the following formula (14).

Here, regarding Expression (13), a _k expressed by Expression (15) below is set as a steering vector of the k-th scattered wave.

When the spectrum multiplication unit 9 calculates the received signal vector x after the spectrum multiplication, the KA type pulse compression units 10-1 to 10-K perform pulse compression on the received signal vector x and use the scattered wave information. In the range side lobe of the digital reception signal z (t _m ) formed by performing the pulse compression, the processing for suppressing the range side lobe of the distance where the main lobe of the scattered wave reflected by the interference source appears is performed. To do.
The processing contents of the KA type pulse compression unit 10-k (k = 1, 2,..., K) will be specifically described below.

First, the scattered wave information input to the KA type pulse compression unit 10-k includes the estimated value of the delay time t _k ^(s) of the K scattered waves and the Doppler frequency f _k ^(D) of the K scattered waves. The estimated value, the estimated accuracy σ _{tk of} the estimated value of the delay time t _k ^(s) , and the estimated accuracy σ _fk of the estimated value of the Doppler frequency f _k ^(D) .
However, the estimation accuracy σ _tk is the estimation error of the delay time t _k ^(s) , the estimation accuracy σ _fk is the estimation error of the Doppler frequency f _k ^(D) , and it goes without saying that the true value is known. Value shall be used. In this case, the estimation accuracy σ _tk and σ _fk are zero.

Various ways of giving scattered wave information to the KA type pulse compression unit 10-k are conceivable.
For example, as the scattered wave information, obtain the predicted value of the distance to the target obtained by tracking the target that is the interference source, the predicted value of the speed of the target, and the predicted accuracy of each predicted value, They can be given to the KA type pulse compression unit 10-k.
Further, as the scattered wave information, it is possible to acquire the distance from the clutter map to the clutter that is the interference source, and to give the distance to the clutter to the KA type pulse compression unit 10-k.
Further, as the scattered wave information, the distance from the synthetic aperture radar image to the interference source can be acquired, and the distance to the interference source can be given to the KA type pulse compression unit 10-k.
Similarly, as the scattered wave information, the distance from the optical image to the interference source can be acquired, and the distance to the interference source can be given to the KA type pulse compression unit 10-k.
Similarly, the distance from the map information to the interference source can be acquired as the scattered wave information, and the distance to the interference source can be given to the KA type pulse compression unit 10-k.
Further, the user can set the scattered wave information as appropriate and give the scattered wave information to the KA type pulse compression unit 10-k.

The sidelobe interference wave information extraction unit 11 of the KA type pulse compression unit 10-k reflects the first to Kth scattered wave information included in the given scattered wave information and reflects it to the kth interference source. Information regarding scattered waves other than the scattered wave to be extracted is extracted as k-th interference wave information.
The KA type pulse compression unit 10-k does not suppress the scattered wave reflected by the k-th interference source, but suppresses the scattered waves other than the scattered wave, so that the scattering reflected by the k-th interference source is suppressed. Information related to scattered waves other than waves is extracted as k-th interference wave information.

ただし、ｋ‘は第ｋの干渉波情報に含まれている第ｋ‘番目の散乱波に対応するインデックスである。
これらのステアリングベクトルは、下記の式（１７）に示すように、計Ｋ−１個になる。

式（１７）において、ａの文字の上に＾の記号を付しているが、明細書の文章中では、電子出願の関係上、文字の上に＾の記号を付することができないので、例えば、ａハットのように表記する。 When the sidelobe interference wave information extraction unit 11 extracts the k-th interference wave information, the interference wave steering vector calculation unit 12 uses the k-th interference wave information to generate the k-th interference wave based on Expression (15). A steering vector is calculated.

Here, k ′ is an index corresponding to the k′-th scattered wave included in the k-th interference wave information.
These steering vectors are K-1 in total as shown in the following equation (17).

In the formula (17), the symbol “^” is added to the letter “a”. However, in the text of the specification, the symbol “^” cannot be added to the letter because of the electronic application. For example, it is expressed as a hat.

When the interference wave steering vector calculation unit 12 calculates the kth interference wave steering vector, the interference wave correlation matrix calculation unit 13 calculates the kth interference wave correlation from the kth interference wave information and the kth interference wave steering vector. A matrix R _{k′k ′} ^(k) is calculated.
Hereinafter, the processing content of the interference wave correlation matrix calculation part 13 is demonstrated concretely.
First, consider the following interference wave correlation matrix R _{k′k ′} ^(k) for K−1 a hats _{k ′} . However, the average interference wave power is 1.

ただし、Δｔ_ｋ’は推定精度σ_ｔｋ’に基づく値であり、Δｆ_ｋ’ ^（Ｄ）は推定精度σ_ｆｋ’に基づく値である。
式（１９）において、ａ（ｔ，ｆ^（Ｄ））は、下記の式（２０）の通りである。

また、ｐ（ｔ，ｆ^（Ｄ））は、相対電力の分布関数である。例えば、推定値ｔハット_ｋ’，ｆハット_ｋ’ ^（Ｄ）を中心として、推定精度Δｔ_ｋ’，Δｆ_ｋ’ ^（Ｄ）を標準偏差とするガウス関数等を与える。 Also, using the estimation accuracy included in the k-th interference wave information, the k-th interference wave correlation matrix R _{k′k ′} ^(k) is expanded as in the following Expression (19).

However, Δt _{k ′} is a value based on the estimation accuracy σ _{tk ′} , and Δf _{k ′} ^(D) is a value based on the estimation accuracy σ _{fk ′} .
In the formula (19), a (t, f ^(D) ) is as the following formula (20).

Further, p (t, f ^(D) ) is a distribution function of relative power. For example, a Gaussian function or the like having the estimated accuracy Δt _{k ′} , Δf _{k ′} ^(D) as the standard deviation around the estimated values t hat _{k ′} , f hat _{k ′} ^(D) is given.

As described above, the k-th interference wave correlation matrix R _kk is given by the following equation (21).

One method of calculating the k-th interference wave correlation matrix R _kk is to directly calculate Equation (21) and use the projection matrix obtained from this as the k-th interference wave correlation matrix.
More specifically, the estimated values t hat _{k ′} and f hat _{k ′} ^(D) , the estimation accuracy Δt _{k ′} and Δf _{k ′} ^(D) , and the assumed distribution function p (t, f ^(D) ) of relative power. (21) is directly calculated by using the eigenvector corresponding to the large eigenvalue selected by the eigenvalue / eigenvector analysis.
Further, instead of using the projection matrix, an appropriate DL (Diagonal Loading) may be performed on Equation (21) to obtain the inverse matrix.

Next, a calculation method of the k-th interference wave suppression matrix when the reference is an LFM (Linear Frequency Modulation) waveform will be described.
In the case of the LFM waveform, a (t, f ^(D) ) is expressed by the following equation (22). However, the tau _TX pulse width of the pulse signal r _{(t m),} B is the bandwidth of the pulse signal r _{(t m).} Further, τ _TX B> 10.

即ち、ａ（ｔ，ｆ^（Ｄ））は、目標信号のドップラ周波数に関するステアリングベクトルａ（ｆ^（Ｄ））と、目標信号の到来時刻に関するステアリングベクトルａ（ｔ）に分解して考えることができる。 Expression (22) can be transformed into the following expression (24).

That is, a (t, f ^(D) ) can be considered by being decomposed into a steering vector a (f ^(D) ) related to the Doppler frequency of the target signal and a steering vector a (t) related to the arrival time of the target signal. .

ただし、簡単のために、相対電力の分布関数はｐ（ｔ，ｆ^（Ｄ））は下記の式（２６）で表し、式中には表記しない。

Next, equation (19) and the equation _{(20), R} k'k _{'the (m} i, _{m j)} of the ^(k) elements _{[R k'k'} ^(k)], the following equation (25) Is given as follows.

However, for the sake of simplicity, in the distribution function of relative power, p (t, f ^(D) ) is expressed by the following equation (26) and is not described in the equation.

式（２７）において、Δｆ_{ｍｉ，ｍｊ}は周波数差ｆ_ｍｉ−ｆ_ｍｊである。 Substituting equation (22) into equation (25) yields equation (27) below.

In Expression (27), Δf _{mi, mj} is a frequency difference f _mi− f _mj .

ここで、これらについて、以下のような変形を行う。
ただし、ｓｉｎｃ（ｘ）＝（ｓｉｎ（πｘ））／πｘである。

Since Expression (27) is an element of the correlation matrix, it is a cross-correlation function relating to Δf _{mi, mj} of delay time and Doppler frequency, and is expressed as Expressions (28) and (29) below, respectively.

Here, the following modifications are performed for these.
However, sinc (x) = (sin (πx)) / πx.

ただし、［Ｒ_ｋ’ｋ’ ^０（ｋ）］_{ｍｉ，ｍｊ}は、Δｔ_ｋ’＝Δｆ_ｋ’ ^（Ｄ）＝０の場合、即ち、推定精度を無視する場合の相関行列Ｒ_ｋ’ｋ’ ^０（ｋ）の第（ｍ_ｉ，ｍ_ｊ）要素であり、以下の通りである。

From the modified results of Expression (30) and Expression (31), [R _{k′k ′} ^(k) ] _{mi, mj} in Expression (27) is expressed as in Expression (32) below.

However, [R _{k′k ′} ^{0 (k)} ] _{mi, mj} is the correlation matrix R _{k′k ′} ⁰ when Δt _{k ′} = Δf _{k ′} ^(D) = 0, that is, when the estimation accuracy is ignored. ^This is the (m _i , m _j ) element of ^(k) and is as follows.

ただし、Ｃ_ｆｋ’はΔｆ_ｋ’ ^（Ｄ）に関するＲ_ｋ’ｋ’ ^０（ｋ）に対するテーパ行列（ＣＭＴ：Ｃｏｖａｒｉａｎｃｅ Ｍａｔｒｉｘ Ｔａｐｅｒ）、Ｃ_ｔｋ’はΔｔ_ｋ’に対するテーパ行列であり、第（ｍ_ｉ，ｍ_ｊ）要素が下記の式（３５）（３６）のように与えられる。

Therefore, from the equation (32), R _{k′k ′} ^(k) is expressed as the following equation (34) using R _{k′k ′} ^{0 (k)} .

Here, C _{fk ′} is a taper matrix (CMT: Covariance Matrix _Tape ⁾ for R _{k′k ′} ^{0 (k) with} respect to Δf _{k ′} ^(D ), C _{tk ′} is a taper matrix for Δt _{k ′} , and (m _i , M _j ) elements are given by the following equations (35) and (36).

That is, the correlation matrix R _{k′k ′} ^{0 (k)} according to the equation (33) is obtained from the estimated value estimated values t hat _{k ′} and f hat _{k ′} ^(D) , and the estimation accuracy Δt _{k ′} and Δf _{k ′} ^(D). If C _{fk ′} and C _{tk ′} are obtained from Equations (35) and (36), a correlation matrix R _{k′k ′} ^(k) is obtained by Hadamard product of these matrices as shown in Equation (34).
Thus, the k-th interference wave correlation matrix R _kk can be obtained from the equation (21).
Thereafter, eigenvalue / eigenvector analysis is performed to obtain a projection matrix using the eigenvector corresponding to the selected large eigenvalue. Further, an inverse matrix may be obtained by performing appropriate DL (Diagonal Loading) on the k-th interference wave correlation matrix R _kk without using the projection matrix. Both function as an interference wave suppression matrix.

式（３７）において、Ｉは単位行列である。 When the interference wave correlation matrix calculation unit 13 calculates the kth interference wave correlation matrix _Rkk , the interference wave suppression matrix calculation unit 14 performs eigenvalue / eigenvector analysis of the kth interference wave correlation matrix _Rkk . Since the k-th interference wave correlation matrix R _kk is a Hermitian matrix, the eigenvalue is a real number.
Next, the interference wave suppression matrix calculation unit 14 selects an eigenvalue having a relatively large value from the eigenvalues of the k-th interference wave correlation matrix _Rkk .
When the interference wave suppression matrix calculation unit 14 selects an eigenvalue having a relatively large value, a matrix that arranges eigenvectors corresponding to the eigenvalue as a column vector is set as E _kk , and a projection matrix P as shown in the following Expression (37) _kk is obtained, and its projection matrix P _kk is set as the k-th interference wave suppression matrix.

In Expression (37), I is a unit matrix.

ただし、ρはＤＬＬ（Ｄｉａｇｏｎａｌ Ｌｏａｄｉｎｇ Ｌｅｖｅｌ）である。 Here, an example in which the projection matrix P _kk is obtained as the k-th interference wave suppression matrix is shown, but the k-th interference wave suppression matrix can be obtained by a method other than the method of obtaining the projection matrix P _kk .
For example, an inverse matrix invR _kk as in the following equation (38) may be obtained as the k-th interference wave suppression matrix.

Here, ρ is a DLL (Diagonal Loading Level).

ここでは、干渉波抑圧部１５が、第ｋの干渉波抑圧行列Ｐ_ｋｋを受信信号ベクトルｘに乗算する例を示したが、干渉波抑圧行列算出部１４により算出された逆行列ｉｎｖＲ_ｋｋを受信信号ベクトルｘに乗算するようにしてもよい。 When the interference wave suppression matrix calculation unit 14 calculates the k-th interference wave suppression matrix P _kk , the interference wave suppression unit 15 receives the k-th interference wave suppression matrix P _kk and the received signal vector x calculated by the spectrum multiplication unit 9. By multiplying by, scattered waves other than the k-th scattered wave included in the received signal vector x are suppressed as interference waves.
When the received signal vector x is multiplied by the kth interference wave suppression matrix _Pkk , the received signal vector x is projected onto a subspace orthogonal to the subspace in which the kth interference wave exists. The k-th interference wave included in x is suppressed.
The received signal vector x _null after the k-th interference wave is suppressed by the interference wave suppressing unit 15 is expressed by the following equation (39).

Here, an example in which the interference wave suppression unit 15 multiplies the reception signal vector x by the k-th interference wave suppression matrix P _kk is shown, but the inverse matrix invR _kk calculated by the interference wave suppression matrix calculation unit 14 is received. The signal vector x may be multiplied.

Upon receiving the received signal vector x _null after the k-th interference wave is suppressed from the interference wave suppressing unit 15, the post-pulse-compression signal calculating unit 16 performs inverse discrete Fourier transform on the received signal vector x _null and converts the received signal vector x _null. The result is output as a received signal vector after pulse compression.

As is apparent from the above, according to the first embodiment, the KA type pulse compression units 10-1 to 10-K perform the pulse compression on the received signal vector x calculated by the spectrum multiplication unit 9, and Of the range side lobes of the digital received signal z (t _m ) formed by performing pulse compression using the scattered wave information, the range side lobes of the distance at which the main lobe of the scattered wave reflected by the interference source appears Therefore, it is possible to sufficiently suppress the interference of the range side lobe caused by the clutter or the existing target.

In the first embodiment, the description has been made assuming the K-1 interference wave, but more generally, the Kp wave (1 ≦ P ≦ K−1) interference wave is assumed. is there.
It is also possible to selectively suppress the K-1 interference wave by using the k-th interference wave suppression matrix P _kk while achieving a low side lobe of the received signal after pulse compression using the window function. It is.
In this case, in the inverse discrete Fourier transform of the received signal vector x _null after the k-th interference wave is suppressed, the received signal vector x _null may be multiplied by a predetermined window function.

  In the present invention, any constituent element of the embodiment can be modified or any constituent element of the embodiment can be omitted within the scope of the invention.

  DESCRIPTION OF SYMBOLS 1 Transmitter (pulse signal generation means), 2 Duplexer (transmission / reception means), 3 Antenna (transmission / reception means), 4 Receiver (transmission / reception means), 5 AD converter (transmission / reception means), 6 Pulse compressor (pulse compression means) 7 discrete Fourier transform unit (first frequency spectrum calculation means), 8 discrete Fourier transform unit (second frequency spectrum calculation means), 9 spectrum multiplication unit (spectrum multiplication means), 10-1 to 10-K KA type Pulse compression unit (first to K-th KA type pulse compression means), 11 sidelobe interference wave information extraction unit, 12 interference wave steering vector calculation unit, 13 interference wave correlation matrix calculation unit, 14 interference wave suppression matrix calculation unit, 15 interference wave suppression unit, 16 pulse-compressed signal calculation unit.

Claims (9)

Pulse signal generating means for generating a pulse signal using a predetermined frequency signal;
A transmission / reception means for radiating the pulse signal generated by the pulse signal generation means to a space in a predetermined beam directing direction while receiving a scattered wave of the pulse signal reflected and returned by a target or a clutter; and
A range side lobe of the received signal formed by performing pulse compression on the received signal of the transmitting / receiving means and using the scattered wave information on the scattered wave reflected by a known interference source. Pulse compression means for suppressing the range side lobe of the distance where the main lobe of the scattered wave reflected by the interference source appears ,
The pulse compression means has first to Kth pulse compression units,
The k-th (k = 1, 2,..., K) pulse compression unit is configured to reflect the first to K-th scattering reflected by the first to K-th interference sources included in the scattered wave information. Interference wave suppression matrix for suppressing the interference wave is calculated from the interference wave information using information on the scattered wave other than the k-th scattered wave among the information on the wave as interference wave information, and the interference wave suppression matrix is used. A radar apparatus that suppresses a scattered wave other than the k-th scattered wave included in the received signal as an interference wave. The pulse compression means includes
A first frequency spectrum calculating means for calculating a frequency spectrum of the received signal of the transmitting and receiving means,
And the second frequency spectrum calculating means for calculating a frequency spectrum of the pulse signal generated by said pulse signal generating means,
Spectrum multiplying means for calculating a received signal vector after spectrum multiplication by multiplying the frequency spectrum calculated by the first frequency spectrum calculating means by the frequency spectrum calculated by the second frequency spectrum calculating means; With
It said pulse compression means, with carrying out the pulse compression on the received signal vector calculated by the spectrum multiplying means, by using the scattered wave information, the range side of the reception signal formed by performing the pulse compression of lobe, radar apparatus of claim 1, wherein suppressing the range side lobe of the distance which the main lobe of the scattered wave reflected on the interference source appears. The kth pulse compression unit includes:
Among information on the scattered waves of the first to K contained in the scattered wave information, the interference to extract information about the scattered wave other than the scattered wave reflected on the interference source of the k as interference wave information of the k A wave information extraction unit;
An interference wave steering vector calculation unit that calculates a kth interference wave steering vector using the kth interference wave information extracted by the interference wave information extraction unit;
An interference wave correlation for calculating a k-th interference wave correlation matrix from the k-th interference wave information extracted by the interference wave information extraction unit and the k-th interference wave steering vector calculated by the interference wave steering vector calculation unit. A matrix calculator;
An interference wave suppression matrix calculator that calculates a k-th interference wave suppression matrix from the k-th interference wave correlation matrix calculated by the interference wave correlation matrix calculator;
The interference suppression matrix of the k calculated by the interference suppression matrix calculating section by multiplying the received signal vector calculated by the spectrum multiplication means, the scattered waves of the k contained in the received signal vector An interference wave suppression unit that suppresses other scattered waves as interference waves,
The radar apparatus according to claim 2, characterized in that it comprises a pulse compressor after signal calculation unit for calculating a received signal vector after the pulse compression from the received signal vector after the interference wave suppression by the interference suppression unit. Said pulse compression means, as the scattered wave information, and the predicted value and the predicted value of the velocity of the target distance to the target obtained by tracking a target is the interference source, the predicted value of the distance The radar apparatus according to claim 1, wherein the prediction accuracy and the prediction accuracy of the predicted value of the speed are acquired. It said pulse compression means, as the scattered wave data, from the clutter map, radar according to any one of claims 1 to 3, characterized in that to obtain the distance to the clutter is the source of interference apparatus. Said pulse compression means, as the scattered wave information, synthesized from aperture radar image, radar apparatus according to any one of claims 1 to 3, characterized in that to obtain the distance to the interferer . Said pulse compression means, as the scattered wave data, from an optical image, a radar apparatus according to any one of claims 1 to 3, characterized in that to obtain the distance to the interference source. It said pulse compression means, as the scattered wave information, from the map information, the radar apparatus according to any one of claims 1 to 3, characterized in that to obtain the distance to the interference source. The radar apparatus according to any one of claims 1 to 3, wherein the pulse compression means acquires scattered wave information set by a user as the scattered wave information.

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