JP5247068B2 - Radar equipment - Google Patents

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JP5247068B2
JP5247068B2 JP2007149551A JP2007149551A JP5247068B2 JP 5247068 B2 JP5247068 B2 JP 5247068B2 JP 2007149551 A JP2007149551 A JP 2007149551A JP 2007149551 A JP2007149551 A JP 2007149551A JP 5247068 B2 JP5247068 B2 JP 5247068B2
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doppler
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time delay
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JP2008304220A (en
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敬之 稲葉
冬樹 福島
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三菱電機株式会社
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Description

  The present invention relates to a radar apparatus that detects a target by transmitting and receiving radio waves.

  Conventionally, a radio wave transmitted from a transmission antenna and reflected by a target is received by the transmission antenna, and the received signal is processed by a signal processing unit including an FFT processing unit and a super-resolution processing unit to estimate a delay time. The type of radar apparatus is well known (for example, see Non-Patent Document 1).

FIG. 38 is a block diagram showing the radar apparatus disclosed in Non-Patent Document 1.
38, a conventional radar apparatus includes a transmitter 1 that generates a pulse-modulated radio wave, a transmission antenna 2 that transmits a radio wave (transmission wave) toward a target 3, and a radio wave reflected by the target 3. A reception antenna 4 that receives (received wave), a receiver 5 that performs band limitation and phase detection on the received wave, an A / D converter 6 that converts a received signal composed of an analog signal into a digital signal, and a digital signal And a super-resolution time delay estimation processing unit 10 for processing the converted received signals s (1),..., S (Nr).

  The super-resolution time delay estimation processing unit 10 performs FFT (Fast Fourier Transform: high speed) for obtaining the frequency spectrum y (1),..., Y (Nr) of the received signal s (1),. Fourier transform) processing means 17 and a divided signal obtained by dividing the frequency spectrum y (1),..., Y (Nr) of the received signal by the frequency spectrum Γ (1),. divide signal generation means 18 for generating x (1),..., x (Nr), and a super-resolution time delay that is estimated based on the divided signals x (1),. Resolution processing means 20.

Next, the operation of the conventional radar apparatus shown in FIG. 38 will be described with reference to FIGS. 39 and 40. FIG. 39 is a block diagram specifically showing the internal configuration of the super-resolution processing means 20 in FIG. 38, and FIG. 40 is an explanatory diagram showing the relationship between the transmission pulse train and the reception pulse train and the delay time between them.
39, the super-resolution processing means 20 includes a correlation matrix generation means 49 that generates a correlation matrix R, an eigenvector calculation means 50 that calculates eigenvectors e (K + 1),..., E (Md), and an eigenvector e ( K + 1),..., E (Md) and MUSIC (Multiple Signal Classification) processing means 51.

  In FIG. 38, when radio waves subjected to pulse modulation are transmitted from the transmitter 1 and the transmission antenna 2, the radio waves reflected by the target 3 are received via the reception antenna 4 and the receiver 5. At this time, as shown in FIG. 40, a received pulse train is obtained in the range of required maximum delay time Nr × T corresponding to the number of samples Nr and the sample interval T with respect to the transmitted pulse train.

  The received wave from the receiving antenna 4 is band-limited by the receiver 5 and phase-detected and input to the A / D converter 6. The A / D converter 6 samples the input signal and outputs received signals s (1),..., S (Nr). At this time, the received signals s (1),..., S (Nr) include the target signal component reflected and received by the target 3 and the noise component of the receiver 5.

  The received signals s (1),... S (Nr) are first input to the FFT processing means 17 in the super-resolution time delay estimation processing unit 10. The FFT processing means 17 performs FFT processing on the received signals s (1),..., S (Nr), and calculates frequency spectra y (1),. The frequency spectrum y (1),..., Y (Nr) is input to the division signal generation means 18. The division signal generation means 18 divides the frequency spectrum y (1),..., Y (Nr) by the transmission wave spectrum Γ (1),. ), Division signals x (1),..., X (Nr) are generated.

  The division signals x (1),..., X (Nr) obtained from the expression (1) by the division signal generation means 18 are input to the super-resolution processing means 20. The division signals x (1),..., X (Nr) input to the super-resolution processing means 20 are first input to the correlation matrix generation means 49 (see FIG. 39) in the super-resolution processing means 20. The correlation matrix generation unit 49 generates a correlation matrix R by the following equation (2).

In Equation (2), Md represents the number of dimensions of the correlation matrix R, and Xm H represents the conjugate transpose of the vector Xm.
The correlation matrix R obtained from the equation (2) by the correlation matrix generation unit 49 is input to the eigenvector calculation unit 50. The eigenvector calculating means 50 includes eigenvalues νe (1),..., Νe (Md) (νe (1)> νe (2)>...> Νe (Md)) of the correlation matrix R and each eigenvalue νe ( i) An eigenvector e (i) corresponding to (1 ≦ i ≦ Md) is obtained, and eigenvectors e (K + 1),..., e (Md) constituting the noise space are output.

  The eigenvectors e (K + 1),..., E (Md) are input to the MUSIC processing means 51. MUSIC processing means 51 performs MUSIC processing using eigenvectors e (K + 1),..., E (Md) as noise spaces. Specifically, the steering vector a (τ) is obtained by the following equation (3) using the sampling interval T of the A / D converter 6 and the kth target delay time τk.

  In addition, the MUSIC processing means 51 estimates the K types of steering vectors a (τk ^) (τk ^ is the delay time of the kth target signal, which are orthogonal to all of the eigenvectors e (K + 1), ..., e (Md). Value) to obtain the estimated delay time value τk ^.

Kikuma "Adaptive signal processing by array antenna"

  Since the conventional radar apparatus is configured as described above, when the phase rotation of the target signal occurs due to the Doppler effect resulting from the movement of the target 3, the time delay estimation by the super-resolution time delay estimation processing unit 10 is performed. There is a problem in that the accuracy deteriorates and the detection accuracy of the target 3 finally detected deteriorates.

  The present invention estimates the Doppler frequency due to the target movement, and corrects the phase rotation of the target signal from the Doppler frequency estimated value, thereby estimating the delay time without degrading the estimation accuracy and accurately targeting the target. An object is to obtain a radar device capable of detection.

  A radar apparatus according to the present invention includes a transmitter that generates radio waves, a transmission antenna that transmits radio waves, a reception antenna that receives a radio wave transmitted from the transmission antenna and reflected by a target as a reception wave, and band limitation of the reception wave And a receiver that generates a target signal corresponding to the target by performing phase detection, a Doppler estimation processing unit that estimates a Doppler frequency of the target signal due to a Doppler effect caused by the movement of the target, and obtains a Doppler frequency estimate, and a Doppler And a time delay estimation processing unit that corrects the phase rotation of the target signal due to the effect and estimates the time delay of the target signal.

  According to the present invention, the Doppler frequency resulting from the target movement is estimated, and the phase rotation of the target signal is corrected from the Doppler frequency estimated value, so that the delay time is estimated without degrading the estimation accuracy. The target can be detected.

Embodiment 1 FIG.
1 is a block diagram showing a radar apparatus according to Embodiment 1 of the present invention. In FIG. 1, the transmitter 1, the transmission antenna 2, the target 3, the reception antenna 4, the receiver 5, the A / D converter 6, and the super-resolution time delay estimation processing unit 10 are the same as those described above (see FIG. 38). The detailed description is omitted.

  In FIG. 1, as a difference from the conventional apparatus (FIG. 38), a plurality of pulse directions FFT7 (# 1) to FFT7 (#) are provided between the A / D converter 6 and the super-resolution time delay estimation processing unit 10. Nr), signal detection processing means 8 and Doppler correction means 9 are inserted.

  The pulse directions FFT7 (# 1),..., FFT7 (#Nr) are classified for each time delay and arranged in parallel on the output side of the A / D converter 6, and the time delay zero received signal,. For each of the time-delayed Nr chip reception signals, FFT processing is performed in the pulse direction.

The signal detection processing means 8 observes the Doppler frequency of the target signal corresponding to the target 3 based on the output signals in the pulse directions FFT7 (# 1) to FFT7 (#Nr), and the target signal Doppler in which the target signal exists. The bin (corresponding to the Doppler frequency observation value fd) and the received signal of the target signal Doppler bin are output.
The Doppler correction means 9 outputs a correction signal obtained by correcting the phase rotation due to the Doppler effect caused by the movement of the target 3 with respect to the target signal included in the output signal of the signal detection processing means 8. Enter 10.

Next, the operation according to the first embodiment of the present invention shown in FIG. 1 will be described with reference to FIGS.
FIG. 2 is an explanatory diagram showing the time relationship of each signal in a plurality of times (Nf times) of transmission and reception, and the horizontal axis corresponds to time. In Figure 2, the chip width T (here, are set to the same value and the sampling interval T) and transmission signal composed of pulses of the chip speed Ns, the reception signal s 1 (1), ···, s 1 , S Nf (1),..., S Nf (Nr) are shown as delay times (time delays).

  FIG. 3 is a block diagram showing a specific configuration of the signal detection processing means 8 in FIG. In FIG. 3, the signal detection processing unit 8 includes a time direction output unit 11 and a target Doppler bin detection processing unit 12. FIG. 4 is an explanatory diagram showing the relationship between the target signal (see gray block), the time delay bin and the Doppler bin, and corresponds to the processing of the target Doppler bin detection processing means 12 (see FIG. 3) in the signal detection processing means 8. doing.

  In FIG. 1, as described above, when a pulse-modulated radio wave is transmitted from the transmitter 1 and the transmission antenna 2 with a pulse chip width T, the radio wave reflected by the target 3 The signal is received via the receiver 5, and a reception signal is output from the receiver 5. At this time, the transmission / reception number is transmitted only in the first pulse repetition period PRF1. The received signal includes a target signal component and a noise component of the receiver 5.

The above transmission / reception of radio waves is repeated Nf times as shown by the time relationship in FIG. In FIG. 2, the time delay of the target signal in the first transmission / reception is τd.
Hereinafter, the received signal is band-limited by the receiver 5, phase-detected, and input to the A / D converter 6. The A / D converter 6 performs sampling at a sampling interval T (the same value as the chip width T of the transmission pulse). At this time, the A / D converter 6 receives only Nr chips (corresponding to the observation area length of the radar device) and receives signals (s 1 (1),..., S 1 (Nr) in the first time ). ) Is sampled and output.

Here, assuming that the received signal in the nf (1 ≦ nf ≦ Nf) transmission / reception is s nf (1),..., S nf (Nr), the received signals s nf (1) ,. (Nr) is classified for each time delay and input to the pulse direction FFT 7 (# 1),..., 7 (#Nr). That is, the nr-th received signal s 1 (nr),..., S Nf (nr) is input to the nr-th (1 ≦ nr ≦ Nr) pulse direction FFT 7 (#nr).

The nr-th pulse direction FFT 7 (#nr) performs FFT processing on the received signals s 1 (nr),..., s Nf (nr) to obtain a Doppler frequency component of the received signal. That is, the pulse direction FFT7 (#nr) is the received signal s 1 (nr), ···, s Nf (nr) Doppler frequency component z0 1 (nr) of, ···, z0 Nf (nr) signal detection Input to the processing means 8.

  Each output signal from the pulse direction FFT 7 (# 1) to 7 (#Nr) is a two-dimensional signal of time and Doppler frequency. The signal detection processing means 8 detects and detects the Doppler bin in which the target signal exists as shown in FIG. 4 from each output signal (two-dimensional signal) from the pulse directions FFT 7 (# 1) to 7 (#Nr). Output the target signal of the Doppler bin.

3, the input signal to the signal detection processing unit 8 z0 1 (1), ··· , z0 Nf (1), ···, z0 1 (Nr), ···, z0 Nf (Nr) is The signal is input to the time direction output means 11 in the signal detection processing means 8. Time direction output unit 11, the signal z0 1 input a sequence of Doppler direction (1), ···, z0 Nf (1), ···, z0 1 (Nr), ···, z0 Nf (Nr ) Are converted into signals z0 1 (1), ..., z0 1 (Nr), ..., z0 Nf (1), ..., z0 Nf (Nr) arranged in the time direction and output. . The output signal from the time direction output unit 11 is input to the target Doppler bin detection processing unit 12.

Here, for example, as shown in FIG. 4, the target signal exists in “nf Doppler bin”, the time delay of the target signal is τk (1 ≦ k ≦ K), and all the time delays τk are nf Doppler. A description will be given assuming a situation where the bins are included in the same time delay bin nτd .
The target Doppler bin detection processing means 12 first detects a target signal by threshold comparison or the like. At this time, as shown in FIG. 4, with respect to the time delay bin, the target signal is detected from n τd bin to n τd + Ns bin, and the time delay bin n τd is expressed by the following equation (4).

However, in the equation (4), floor [τk / T] is a function for rounding off the value of “τk / T”.
Subsequently, the target Doppler bin detection processing means 12 calculates the Doppler frequency observation value fd of the target signal by the following equation (5).

As a result, the signal detection processing means 8 outputs the Doppler frequency observation value fd corresponding to the nf Doppler bin and the received signals z0 Nf (1),..., Z0 Nf (Nr) of the Doppler frequency observation value fd. Is done. The Doppler frequency observation value fd and the received signals z0 nf (1),..., Z0 nf (Nr) are input to the Doppler correction means 9. The Doppler correction unit 9 corrects and corrects the phase rotation of the target signal included in the received signals z0 nf (1),..., Z0 nf (Nr) due to the Doppler effect by the following equation (6). Signal.

The Doppler correction signals z0 ′ nf (1),..., Z0 ′ nf (Nr) obtained from the equation (6) by the Doppler correction means 9 are input to the super-resolution time delay estimation processing unit 10. Thereafter, the super-resolution time delay estimation processing unit 10 operates in the same manner as described above to estimate the time delay of the target signal.

  As described above, the radar apparatus according to Embodiment 1 of the present invention includes the transmitter 1 that generates radio waves, the transmission antenna 2 that transmits radio waves, and the radio waves that are transmitted from the transmission antenna 2 and reflected by the target 3. A receiving antenna 4 that receives the signal as a received wave, a receiver 5 that performs band limitation and phase detection of the received wave to generate a target signal corresponding to the target, and a Doppler frequency of the target signal due to the Doppler effect resulting from the movement of the target 3 And a time delay estimation processing unit for correcting a phase rotation of the target signal due to the Doppler effect and estimating a time delay of the target signal.

  The Doppler estimation processing unit also converts the received signal obtained through the receiver 5 into a digital signal, and the received signal converted into the digital signal through the A / D converter 6. Detects Doppler bins where the target signal exists based on the output signals in the pulse direction FFT7 (# 1) to 7 (#Nr) and the pulse direction FFT7 (# 1) to 7 (#Nr) to perform FFT processing in the pulse hit direction And a Doppler correction unit 9 for correcting the phase rotation of the target signal due to the Doppler effect, and the time delay estimation processing unit performs a super-resolution time delay estimation process for super-resolution estimation of the time delay of the target signal. The unit 10 is configured.

  Accordingly, the signal detection processing unit 8 estimates the Doppler frequency, and the Doppler correction unit 9 corrects the phase rotation of the target signal from the Doppler frequency estimation value fd, so that the delay time is estimated without deteriorating the estimation accuracy. And the target 3 can be detected with high accuracy.

Embodiment 2. FIG.
In the first embodiment (see FIG. 1), the super-resolution time delay estimation processing unit 10 similar to that in the conventional apparatus (see FIG. 38) is used. However, as shown in FIG. The estimation processing unit 14 may be used.
5 is a block diagram showing a radar apparatus according to Embodiment 2 of the present invention. In FIG. 5, the transmitter 1, the transmission antenna 2, the target 3, the reception antenna 4, the receiver 5, the A / D converter 6, and the pulse direction FFT 7 (# 1) to 7 (#) similar to the above (see FIG. 1). Nr), the Doppler correction means 9 will be given the same reference numerals as those described above and will not be described in detail.

In this case, the signal detection processing unit 13 observes the Doppler frequency of the target signal, and inputs the Doppler bin signal in which the target signal exists as described above to the Doppler correction unit 9, and further, the time gate type super-resolution time. A time delay bin of the target signal is input to the delay estimation processing unit 14.
The time gate type super-resolution time delay estimation processing unit 14 estimates the time delay of the target signal to the super-resolution by limiting the time gates to short time gates.

  Next, the operation according to the second embodiment of the present invention will be described with reference to FIGS. 6 to 9 together with FIG. FIG. 6 is a block diagram showing the internal configuration of the signal detection processing means 13, and FIG. 7 is a block diagram showing the internal configuration of the time gate type super-resolution time delay estimation processing unit 14. FIG. 8 is an explanatory diagram showing the time relationship of the time gate, and corresponds to the processing of the time gate processing means 16 in FIG. FIG. 9 is a block diagram showing the internal configuration of the decimation processing means 19 in FIG.

In FIG. 6, the signal detection processing means 13 includes the time direction output means 11 described above (see FIG. 3) and the target Doppler bin detection processing means 15.
In FIG. 7, the time gate type super-resolution time delay estimation processing unit 14 includes a time gate processing unit 16, an FFT processing unit 17 and a division signal generation unit 18 described above (see FIG. 38), a decimation processing unit 19, and (See FIG. 38).

FIG. 8 shows reception states at time gates (1), (2),..., Ig,..., Nd / Ns of the same time width Nd in the Nr chip (observation region). In this example, a received pulse is detected at the ig-th time gate ig. In FIG. 8, the horizontal axis corresponds to time.
9, the decimation processing means 19 multiplies the division signals x ′ nf , ig (1),..., X ′ nf , ig (Nd + Ns) by respective weighting factors w (m) (1 ≦ m ≦ Ndeci). And an adder for adding the multiplication results for each block.

In FIG. 5, first, as described above, when radio waves subjected to pulse modulation are transmitted from the transmitter 1 and the transmission antenna 2, the radio waves reflected by the target 3 are received by the receiver 4 and the reception antenna 5. The Hereinafter, the received signal through the A / D converter 6 is input to the pulse direction FFT 7 (#nr) (1 ≦ nr ≦ Nr) and output from the pulse direction FFT 7 (#nr) (1 ≦ nr ≦ Nr). The signals z0 1 (nr),..., Z0 Nf (nr) are input to the signal detection processing means 13.

Next, in FIG. 6, the input signals z0 1 (nr),..., Z0 Nf (nr) to the signal detection processing unit 13 are input to the time direction output unit 11 as described above, and the time direction output unit 11, the signals z0 i (1),..., Z0 i (Nr) (1 ≦ i ≦ Nf) converted in the time delay direction are output and input to the target Doppler bin detection processing means 15. The

Subsequently, from the target Doppler bin detection processing means 15, similarly to the target Doppler bin detection processing means 12 described above (see FIG. 3), the Doppler frequency observation value fd of the target signal and the target signal Doppler bin (nf Doppler bin) of the received signal z0 nf (1), ···, z0 nf (Nr) and is output, is input to the Doppler compensation means 9. Hereinafter, the Doppler correction means 9 outputs Doppler correction signals z0 ′ nf (1),..., Z0 ′ nf (Nr) corrected for phase rotation due to the Doppler effect, and time gate type super-resolution time delay estimation. Input to the processing unit 14.
On the other hand, the target Doppler bin detection processing means 15 further outputs a time delay bin n τd of the target signal and inputs it to the time gate type super-resolution time delay estimation processing unit 14.

In FIG. 7, first, the time gate processing means 16 in the time gate type super-resolution time delay estimation processing unit 14 divides the observation area of Nr chip into time gates of Nd chip (see FIG. 8), and time delay bin n Check the number of the time gate including τd .

In FIG. 8, it is assumed that a received pulse train (gray shaded portion) is detected after a certain time delay with respect to a transmitted pulse train (shaded portion). At this time, the timing at which the received pulse train is detected is the time delay bin n τd , and the time delay bin n τd is included in the ig-th time gate ig.

Reflected waves from the K targets are superimposed on the time gate ig with a slight time delay difference. Therefore, the time gate processing means 16 is a time gate signal z nf ((ig−1) Nd + 1),..., Z nf (ig · Nd + Ns) obtained by adding the chip length (Ns chip) of the transmission pulse to the time gate ig. Is output. Here, the time gate signal z nf ((ig−1) Nd + i) (1 ≦ i ≦ Nd + Ns) is expressed as z nf , ig (i). Time gate signals z nf , ig (1),..., Z nf , ig (Nd + Ns) are input to the FFT processing means 17.

The FFT processing means 17 performs FFT processing on the time gate signals z nf , ig (1),..., Z nf , ig (Nd + Ns), and performs FFT processing signals y nf , ig (1) ,. nf , ig (Nd + Ns) is generated and input to the division signal generation means 18.
Similarly to the above, the division signal generation means 18 uses the transmission wave spectrum Γ (1),..., Γ (Nd + Ns), and the division signal x ′ nf , ig (1) according to the following equation (7). ,..., X ′ nf , ig (Nd + Ns) are generated.

Subsequently, the division signals x ′ nf , ig (1),..., X ′ nf , ig (Nd + Ns) are input to the decimation processing means 19.
In FIG. 9, the decimation processing means 19 uses a weighting coefficient w (m) (1 ≦ m ≦ Ndeci) that determines the characteristics of the decimation filter, and performs a decimation processing on the divided signals x nf , ig (1),. X nf , ig (Nd) is generated by the following equation (8).

In equation (8), floor [] represents a function that rounds off a real value and outputs an integer value.
The decimation processing signals x nf , ig (1),..., X nf , ig (Nd) are input to the super-resolution processing means 20. Thereafter, the super-resolution processing means 20 in the time-gated super-resolution time delay estimation processing unit 14 obtains the target signal delay time estimated value τk ^ (1 ≦ k ≦ K) corresponding to the target 3 as described above. It is done.

  As described above, the radar apparatus according to Embodiment 2 of the present invention includes a transmitter 1 that generates radio waves, a transmission antenna 2 that transmits radio waves, and a radio wave that is transmitted from the transmission antenna 2 and reflected by the target 3. A receiving antenna 4 that receives the received signal as a received wave, a receiver 5 that performs band limitation and phase detection of the received wave to generate a target signal corresponding to the target 3, and a Doppler of the target signal due to the Doppler effect resulting from the movement of the target 3 A Doppler estimation processing unit that estimates a frequency and obtains a Doppler frequency estimation value fd; and a time delay estimation processing unit that estimates a time delay of the target signal by correcting phase rotation of the target signal due to the Doppler effect. .

  The Doppler estimation processing unit also converts the received signal obtained through the receiver 5 into a digital signal, and the received signal converted into the digital signal through the A / D converter 6. Detects Doppler bins where the target signal exists based on the output signals in the pulse direction FFT7 (# 1) to 7 (#Nr) and the pulse direction FFT7 (# 1) to 7 (#Nr) to perform FFT processing in the pulse hit direction And a Doppler correction unit 9 that corrects the phase rotation of the target signal due to the Doppler effect, and the time delay estimation processing unit limits the time delay estimation range and superimposes the time delay of the target signal. The time gate type super-resolution time delay estimation processing unit 14 to be estimated is configured.

  As a result, by limiting the time delay estimation range to the time gate range in the time gate type super resolution time delay estimation processing unit 14, it is possible to reduce the super resolution time delay estimation processing load for Doppler correction.

Embodiment 3 FIG.
In the first embodiment (see FIG. 1), the signal detection processing means 8 that does not consider pulse compression is used. However, as shown in FIGS. 10 to 12, the pulse compression signal detection processing means 22 is used. Also good.
FIG. 10 is a block diagram showing a radar apparatus according to Embodiment 3 of the present invention, and FIG. 11 is a block diagram specifically showing the pulse compression type signal detection processing means 22 in FIG. FIG. 12 is a block diagram specifically showing the pulse compression means 23 (23 (# 1) to 23 (#Nr)) in FIG.

  In FIG. 10, the transmitter 1, the transmission antenna 2, the target 3, the reception antenna 4, the receiver 5, the A / D converter 6, and the pulse direction FFT 7 (# 1) to 7 (#) similar to the above (see FIG. 1). Nr), the Doppler correction means 9 and the super-resolution time delay estimation processing unit 10 are assigned the same reference numerals as those described above and will not be described in detail.

  The radar apparatus of FIG. 10 includes a memory circuit 21 associated with pulse direction FFTs 7 (# 1) to 7 (#Nr) and pulse compression type signal detection processing means 22. The memory circuit 21 stores the pulse direction FFT 7 (# 1) to 7 (#Nr) and the output signal of the pulse compression type signal detection processing means 22 and inputs them to the Doppler correction means 9.

  In this case, the pulse compression type signal detection processing unit 22 uses the target signal subjected to the pulse compression processing based on the output signals in the pulse directions FFT7 (# 1) to 7 (#Nr), and the target signal exists. Detect Doppler bins. That is, the pulse compression type signal detection processing means 22 detects the target signal after improving the S / N (signal power to noise power ratio) by the pulse compression processing based on the pulse direction FFT output, and detects the target signal Doppler bin. Output the received signal.

  In FIG. 11, the pulse compression type signal detection processing unit 22 includes a time direction output unit 11 similar to that described above, a plurality of pulse compression units 23 (# 1) to 23 (#Nr), and a pulse compression unit 23 (# 1). ) To 23 (#Nr), the target Doppler bin detection processing means 26 for outputting the Doppler frequency observation value fd is provided.

  In FIG. 12, the pulse compression means 23 includes an FFT processing means 17 similar to that described above (see FIG. 7) provided on the input side, a correlation processing means 24 comprising a plurality of multipliers, and an IFFT provided on the output side. (Inverse fast Fourier transform) processing means 25. Note that the pulse compression unit 23 integrally shows the pulse compression units 23 (# 1) to 23 (#Nr) in FIG. 11, and is represented by the pulse compression unit 23 (#if) (1 ≦ if ≦ Nf). It can also be expressed.

Next, the operation according to the third embodiment of the present invention shown in FIGS. 10 to 12 will be described.
First, in FIG. 10, when radio waves are transmitted and received in the same manner as described above, signals z0 1 (nr),..., Z0 Nf (nr) are obtained from the pulse direction FFT7 (#nr) (1 ≦ nr ≦ Nr). It is output and input to the pulse compression type signal detection processing means 22 and the memory circuit 21.

Subsequently, in FIG. 11, the input signals z0 1 (nr),..., Z0 Nf (nr) to the pulse compression signal processing means 22 are inputted to the time direction output means 11 and described above (FIG. 3, FIG. Similarly to 6), the converted signals z0 if (1),..., Z0 if (Nr) (1 ≦ if ≦ Nf) in the time delay direction are output. The time direction output means output signals z0 if (1),..., Z0 if (Nr) are input to the pulse compression means 23 (#if).

Next, in FIG. 12, input signals z0 if (1),..., Z0 if (Nr) to the pulse compression means 23 (#if) are input to the FFT processing means 17. The FFT processing means 17 performs FFT processing on the input signal z0 if (1),..., Z0 if (Nr), and the frequency spectrum z0f if (1),..., Z0f if (Nr) of the input signal. ) Is output. The frequency spectrum z0f if (1),..., Z0f if (Nr) is input to the correlation processing unit 24.

The correlation processing unit 24 multiplies the frequency spectrum z0f if (1),..., Z0f if (Nr) by the transmission wave spectrum Γ ′ (1),. Signals Γ ′ * (1) × z0f if (1),..., Γ ′ * (Nr) × z0f if (Nr) are output. Here, Γ ′ * (nr) represents a complex conjugate of the transmission wave spectrum Γ ′ (nr). The multiplication signals Γ ′ * (1) × z0f if (1),..., Γ ′ * (Nr) × z0f if (Nr) are input to the IFFT processing means 25.

The IFFT processing means 25 performs IFFT processing on the input signal Γ ′ * (1) × z0f if (1),..., Γ ′ * (Nr) × z0f if (Nr), and generates a pulse compression signal z1. If (1),..., z1 if (Nr) is output. The pulse compression signals z1 if (1),..., Z1 if (Nr) are input to the target Doppler bin detection processing means 26 (see FIG. 11) in the pulse compression type signal detection processing means 22.

At this time, the received signal of the target signal Doppler bin (nf Doppler bin) and the component z1 nf (n τd ) of the time delay bin n τd are integrated with the target signal component by pulse compression, and the power is particularly large. .
The target Doppler bin detection processing means 26 detects the target signal component z1 nf (n τd ) by threshold comparison or the like, and outputs the Doppler frequency observation value fd of the target signal. The Doppler frequency observation value fd is input to the Doppler correction means 9 and the memory circuit 21 (see FIG. 10).

The memory circuit 21 outputs nf Doppler bin reception signals z0 nf (1),..., Z0 nf (Nr) corresponding to the Doppler frequency observation value fd. The nf Doppler bin received signals z0 nf (1),..., z0 nf (Nr) are input to the Doppler correction means 9.
Thereafter, based on the correction signal from the Doppler correction means 9, the super-resolution time delay estimation processing unit 10 obtains the time delay estimated value τk ^ (1 ≦ k ≦ K) of the target signal as described above.

  As described above, the radar apparatus according to Embodiment 3 of the present invention includes the transmitter 1 that generates radio waves, the transmission antenna 2 that transmits radio waves, and the radio waves that are transmitted from the transmission antenna 2 and reflected by the target 3. A receiving antenna 4 that receives the signal as a received wave, a receiver 5 that performs band limitation and phase detection of the received wave to generate a target signal corresponding to the target, and a Doppler frequency of the target signal due to the Doppler effect resulting from the movement of the target 3 And a Doppler estimation processing unit for obtaining the Doppler frequency estimation value fd, and a time delay estimation processing unit for correcting the phase rotation of the target signal due to the Doppler effect to estimate the time delay of the target signal.

  The Doppler estimation processing unit also converts the received signal obtained through the receiver 5 into a digital signal, and the received signal converted into the digital signal through the A / D converter 6. Doppler bins in which a target signal exists using pulse direction FFTs 7 (# 1) to 7 (#Nr) for performing FFT processing in the pulse hit direction and a target signal subjected to pulse compression processing based on an output signal in the pulse direction FFT And a Doppler correction unit 9 that corrects the phase rotation of the target signal due to the Doppler effect, and the time delay estimation processing unit is a super-resolution estimation unit that estimates the time delay of the target signal. The resolution time delay estimation processing unit 10 is configured.

  Thus, since the S / N is improved by the pulse compression in the pulse compression type signal detection processing means 22, the detection performance of the target detection process performed in the super-resolution time delay estimation processing unit 10 can be improved.

Embodiment 4 FIG.
In the third embodiment (see FIGS. 10 and 11), the pulse compression type signal detection processing unit 22 that outputs the Doppler frequency observation value fd and the super-resolution time delay estimation processing unit 10 are used. 13 and FIG. 14, the pulse compression type signal detection processing means 27 for outputting the target signal Doppler bin (nf Doppler bin) and the time delay bin n τd corresponding to the Doppler frequency observation value, and the time gate type super resolution time The delay estimation processing unit 14 may be used.

13 is a block diagram showing a radar apparatus according to Embodiment 4 of the present invention, and FIG. 14 is a block diagram specifically showing the pulse compression type signal detection processing means 27 in FIG.
In FIG. 13, the transmitter 1, the transmission antenna 2, the target 3, the reception antenna 4, the receiver 5, the A / D converter 6, the pulse direction FFT7 (# 1) to the same as described above (see FIGS. 5 and 10) 7 (#Nr), Doppler correction means 9, time gate type super-resolution time delay estimation processing unit 14, and memory circuit 21 are denoted by the same reference numerals as those described above, and detailed description thereof is omitted.

In FIG. 14, the pulse compression type signal detection processing means 27 is provided on the output side with the time direction output means 11 and the pulse compression means 23 (#if) (1 ≦ if ≦ Nf) similar to those described above (see FIG. 11). The target Doppler bin detection processing means 28 is configured.
In this case, the pulse compression type signal detection processing means 27 detects the target signal after improving the S / N by the pulse compression processing, and outputs the target signal Doppler bin (Doppler frequency observation value) and the time delay bin nτd . .

Next, the operation according to the fourth embodiment of the present invention shown in FIGS. 13 and 14 will be described.
First, in FIG. 13, when radio waves are transmitted and received in the same manner as described above, signals z0 1 (nr),..., Z0 Nf (nr) are output from the pulse direction FFT7 (#nr) (1 ≦ nr ≦ Nr). And input to the pulse compression signal detection processing means 27 and the memory circuit 21.

Subsequently, in FIG. 14, input signals z0 1 (nr),..., Z0 Nf (nr) (1 ≦ nr ≦ Nr) to the pulse compression signal detection processing unit 27 are transmitted via the time direction output unit 11. is input to the pulse compression means 23 (#if) Te in the same manner as described above, the pulse compressed signal z1 if (1), ···, are converted into z1 if (Nr) (1 ≦ if ≦ Nf), the target doppler Input to the bin detection means 28.

The target Doppler bin detection means 28 detects the target signal component z1 nf (n τd ) as described above, and outputs the nf Doppler bin corresponding to the Doppler frequency observation value fd of the target signal and the time delay bin n τd. To do. The Doppler frequency observation value fd (nf Doppler bin) is input to the Doppler correction means 9 and the memory circuit 21, and the time delay bin n τd is input to the time gate type super-resolution time delay estimation processing unit 14.

The memory circuit 21 inputs the nf Doppler bin received signals z0 nf (1),..., Z0 nf (Nr) corresponding to the Doppler frequency observation value fd to the Doppler correction means 9. Thereafter, based on the correction signal from the Doppler correction means 9, the time gate type super-resolution time delay estimation processing unit 14 obtains the time delay estimated value τk ^ (1 ≦ k ≦ K) of the target signal as described above. It is done.

As described above, according to the radar apparatus according to Embodiment 4 of the present invention, by limiting the time delay estimation range to the time gate range, the Doppler correction means 9 and the time gate super-resolution time delay estimation processing unit 14 The processing load can be reduced.
Further, since the S / N is improved by the pulse compression in the pulse compression type signal detection processing means 27, the detection performance of the target detection process performed in the super-resolution time delay estimation process can be improved.

Embodiment 5 FIG.
In the first embodiment (see FIGS. 1 and 3), the signal detection processing means 8 having the time direction output means 11 and the target Doppler bin detection processing means 12 is used. However, as shown in FIG. Only the direction output means 29 may be used.
15 is a block diagram showing a radar apparatus according to Embodiment 5 of the present invention. In FIG. 15, the transmitter 1, the transmission antenna 2, the target 3, the reception antenna 4, the same as those described above (see FIG. 1), The receiver 5, the A / D converter 6, and the pulse direction FFTs 7 (# 1) to 7 (#Nr) are denoted by the same reference numerals as those described above, and detailed description thereof is omitted.

In FIG. 15, on the output side of the time direction output means 29, a plurality of Doppler correction means 9 (# 1) to 9 (#Nr) and a plurality of super-resolution time delay estimation processing units 10 (# 1) to 10 (# 1 to 10) #Nr) is provided.
The Doppler direction output means 29 converts the signals arranged in the pulse direction in the time delay direction, and outputs the Doppler frequency observation value fd of the target signal and the received signal of the target signal Doppler bin (nf Doppler bin).

Next, the operation according to the fifth embodiment of the present invention shown in FIG. 15 will be described.
First, as described above, when radio waves are transmitted and received, signals z0 1 (nr),..., Z0 Nf (nr) (1 ≦ nr ≦ Nr) are input to the time direction output unit 29.
The time direction output means 29 converts the signals z0 1 (nr),..., Z0 Nf (nr), which are input in the Doppler direction, in the time direction, and converts the signals z0 if (1),. Z0 if (Nr) (1 ≦ if ≦ Nf) is output, and the Doppler frequency fd (if) corresponding to the Doppler bin if is calculated by the following equation (9) and output.

The output signals z0 if (1),..., Z0 if (Nr) and the Doppler frequency fd (if) of the time direction output means 29 are input to the Doppler correction means 9 (#if).
Thereafter, based on the correction signal from the Doppler correction means 9 (#if), the super-resolution time delay estimation processing unit 10 (#if) similarly performs the super-resolution time delay related to the nf Doppler bin in which the target signal exists. A time delay estimated value τk ^ (1 ≦ k ≦ K) is obtained from the estimation processing unit 10 (#nf).

  As described above, according to the fifth embodiment of the present invention, all Doppler bins are processed by the super-resolution time delay estimation processing unit 10 (# 1) that performs Doppler correction. Therefore, the target 3 can be reliably detected.

Embodiment 6 FIG.
In the first embodiment (see FIG. 1), the switching of the transmission pulse is not considered. However, as shown in FIG. 16, the H / L-PRF generator 30 is provided to switch the transmission pulse. Good.
FIG. 16 is a block diagram showing a radar apparatus according to Embodiment 6 of the present invention, and FIG. 17 is a block diagram specifically showing the signal detection processing means 32 in FIG.

  In FIG. 16, the same transmitter 1, transmitting antenna 2, target 3, receiving antenna 4, receiver 5, A / D converter 6, pulse direction FFT 7 (# 1) to 7 (# 1) as described above (see FIG. 1) Nr), the Doppler correction means 9 and the super-resolution time delay estimation processing unit 10 are assigned the same reference numerals as those described above and will not be described in detail.

  In FIG. 16, an H / L-PRF generator 30, a changeover switch 30a, and a Doppler frequency estimation means 31 are added. The Doppler frequency estimation means 31 outputs a Doppler frequency estimated value fd ^, and a signal detection processing means. 32 outputs only the received signal of the target signal Doppler bin.

The H / L-PRF generator 30 inputs a transmission switching instruction to the transmitter 1, and transmits a high pulse repetition frequency (HPRF) transmission pulse and a low pulse repetition frequency (LPRF) from the transmitter 1 and the transmission antenna 2. The transmission pulse is switched and transmitted.
In addition, when the HPRF transmission pulse is selected, the H / L-PRF generator 30 switches the changeover switch 30a from the illustrated normal state to the Doppler frequency estimation means 31 side.

  The Doppler frequency estimation means 31 constitutes a Doppler estimation processing section together with the signal detection processing means 32 and the Doppler correction means 9, detects a target signal by transmission / reception based on a transmission pulse of HPRF, estimates a Doppler frequency, The frequency estimated value fd ^ (the Doppler bin received signal in which the target signal exists) is input to the Doppler correction means 9.

In FIG. 17, the signal detection processing means 32 is the same as the time direction output means 11 as described above (see FIG. 3), and only the received signals z0 nf (1),..., Z0 nf (Nr) of the target signal Doppler bin. And a target Doppler bin detection processing means 33 for outputting.

Next, the operation according to the sixth embodiment of the present invention shown in FIGS. 16 and 17 will be described.
First, in FIG. 16, the transmitter 1 and the transmission antenna 2 transmit HPRF pulses in response to an instruction from the H / L-PRF generator 30, and the reception antenna 4 and the receiver 5 Receive reflected waves. At this time, the changeover switch 30a is switched from the state shown in the figure and is conducted to the Doppler frequency estimation means 31 side.
Thus, the received signals s ′ 1 ,..., S ′ Nf output from the A / D converter 6 at the time of HPRF pulse transmission / reception are input to the Doppler frequency estimation means 31.

Doppler frequency estimation unit 31, the received signal s '1, ···, s' performs FFT processing on Nf, the Doppler signal sf '1, ···, sf' generates Nf, due threshold comparison, the target The signal Doppler signal sf ′ nf is detected. Further, from the nf Doppler bin, the Doppler frequency estimated value fd ^ is obtained by the following equation (10).

However, in Formula (10), T HPRF represents the pulse repetition frequency of HPRF.

Next, in response to an instruction from the H / L-PRF generator 30, the transmitter 1 and the transmitting antenna 2 transmit the radio waves subjected to pulse modulation as LPRF pulses. Further, the H / L-PRF generator 30 returns the changeover switch 30a to the illustrated state.
In the same manner as described above, signals z0 1 (nr),..., Z0 Nf (nr) (1 ≦ nr ≦ Nr) are input to the signal detection processing unit 32.

That is, in FIG. 17, signals z0 1 (nr),..., Z0 Nf (nr) are input to the time direction output means 11, and converted signals z0 if (1),. , Z0 if (Nr) (1 ≦ if ≦ Nf) is input to the target Doppler bin detection processing means 33.

The target Doppler bin detection processing unit 33 detects the target signal in the same manner as described above, outputs the nf Doppler bin reception signals z0 nf (1),..., Z0 nf (Nr), and inputs them to the Doppler correction unit 9. To do.
The Doppler correction means 9 receives not only the received signal of the target signal Doppler bin but also the Doppler frequency estimation value fd ^ from the Doppler frequency estimation means 31. Thereafter, based on the correction signal from the Doppler correction means 9, the super-resolution time delay estimation processing unit 10 determines the time delay estimated value τk ^ (1 ≦ k ≦ K) in the same manner as described above.

  As described above, the radar apparatus according to Embodiment 6 of the present invention includes a transmitter 1 that generates radio waves, a transmission antenna 2 that transmits radio waves, and a radio wave that is transmitted from the transmission antenna 2 and reflected by the target 3. A receiving antenna 4 that receives the received signal as a received wave, a receiver 5 that performs band limitation and phase detection of the received wave to generate a target signal corresponding to the target, and a Doppler frequency of the target signal due to the Doppler effect resulting from the movement of the target. A Doppler estimation processing unit that estimates and calculates a Doppler frequency estimated value fd ^, a time delay estimation processing unit that corrects phase rotation of the target signal due to the Doppler effect and estimates a time delay of the target signal, and transmits to the transmitter 1 An H / L-PRF generator 30 is provided for inputting a switching instruction and switching the transmission pulse of HPRF or LPRF from the transmitter 1 for transmission. .

The Doppler estimation processing unit includes Doppler frequency estimation means 31 that estimates the Doppler frequency of the target signal by transmission and reception based on the transmission pulse of HPRF.
In this way, the Doppler frequency estimation means 31 can estimate the Doppler frequency without the ambiguity by widening the Doppler frequency band and estimating the Doppler frequency using HPRF. The accuracy of correction can be improved.

Embodiment 7 FIG.
In the sixth embodiment (see FIGS. 16 and 17), the signal detection processing unit 32 that outputs only the received signal of the target signal Doppler bin and the super-resolution time delay estimation processing unit 10 are used. As shown in FIG. 18 and FIG. 19, a signal detection processing means 34 for outputting the received signal and time delay of the target signal Doppler bin and a time gate type super resolution time delay estimation processing unit 14 may be used.

18 is a block diagram showing a radar apparatus according to Embodiment 7 of the present invention, and FIG. 19 is a block diagram specifically showing the signal detection processing means 32 in FIG.
In FIG. 18, the same transmitter 1, transmission antenna 2, target 3, reception antenna 4, receiver 5, A / D converter 6, pulse direction FFT 7 (# 1) to 7 (#) as described above (see FIG. 16). Nr), the Doppler correction means 9, the H / L-PRF generator 30, and the Doppler frequency estimation means 31 are assigned the same reference numerals as those described above and will not be described in detail. Further, the time gate type super resolution time delay estimation processing unit 14 is the same as that in the above-described fourth embodiment (see FIG. 13).

In FIG. 19, the signal detection processing means 34 is constituted by the time direction output means 11 and the target Doppler bin detection processing means 35 similar to those described above, detects the target signal, and receives the received signal of the target signal Doppler bin. z0 nf (1), ···, and outputs the z0 nf and (Nr) time delay bin n τd.

Next, the operation according to the seventh embodiment of the present invention shown in FIGS. 18 and 19 will be described.
First, similarly to the above, in FIG. 18, the transmitter 1 and the transmission antenna 2 transmit HPRF pulses in response to an instruction from the H / L-PRF generator 30, and the reception antenna 4 and the receiver 5 The reflected wave from the target 3 is received. The reception signals s ′ 1 ,..., S ′ Nf output from the A / D converter 6 are input to the Doppler frequency estimation unit 31, and the Doppler frequency sf ′ nf of the target signal is obtained by the Doppler frequency estimation unit 31. The Doppler frequency estimated value fd ^ is detected from the target signal Doppler bin (nf Doppler bin) and input to the Doppler correction means 9.

Next, in response to an instruction from the H / L-PRF generator 30, the pulse-modulated LPRF is transmitted and received, and signals z0 1 (nr),..., Z0 Nf (nr) ( 1 ≦ nr ≦ Nr) is input from the A / D converter 6 to the signal detection processing means 34 via the pulse direction FFT.

Subsequently, in FIG. 19, signals z0 if (1),..., Z0 if (Nr) (1 ≦ if ≦ Nf) converted in the time delay direction by the time direction output unit 11 in the signal detection processing unit. Is input to the target Doppler bin detection processing means 35.
Target Doppler bin detection processing unit 35 detects the target signal, the received signal z0 nf (1) of the target signal Doppler bins, ..., and outputs the z0 nf (Nr), and outputs the time delay bin n .tau.d . Thereafter, based on the correction signal from the Doppler correction means 9, the time gate type super-resolution time delay estimation processing unit 14 determines the time delay estimated value τk ^ (1 ≦ k ≦ K) in the same manner as described above.

  As described above, according to the seventh embodiment of the present invention, the super-resolution time delay estimation process in which Doppler correction is performed in consideration of ambiguity by limiting the time delay estimation range to the time gate range. The load can be reduced.

Embodiment 8 FIG.
In the sixth embodiment (see FIGS. 16 and 17), the signal detection processing means 32 that outputs the received signal of the target signal Doppler bin without considering the pulse compression is used. However, as shown in FIG. The same pulse compression type signal detection processing means 22 as that of the third embodiment (see FIGS. 10 and 11) may be used.

  20 is a block diagram showing a radar apparatus according to Embodiment 8 of the present invention. The same transmitter 1, transmission antenna 2, target 3, reception antenna 4, receiver 5 as those described above (see FIG. 16), A / D converter 6, pulse direction FFT 7 (# 1) to 7 (#Nr), Doppler correction means 9, super-resolution time delay estimation processing section 10, H / L-PRF generator 30, Doppler frequency estimation means 31 Are denoted by the same reference numerals as those described above and will not be described in detail. The memory circuit 21 is the same as that in the third embodiment (see FIG. 10).

Next, the operation according to the eighth embodiment of the present invention shown in FIG. 20 will be described with reference to FIG.
First, in the same manner as described above, in response to an instruction from the H / L-PRF generator 30, HPRF pulses are transmitted and received to obtain the Doppler frequency estimated value fd ^, and then LPRF pulses are transmitted and received. As a result, the output signal in the pulse direction FFT is input to the pulse compression type signal detection processing means 22 and the memory circuit 21.

Subsequently, the Doppler frequency observation value fd (see FIG. 11) observed by the pulse compression signal detection processing means 22 is input to the memory circuit 21, and the received signal of the nf Doppler bin corresponding to the Doppler frequency observation value fd is stored in the memory circuit. 21 to the Doppler correction means 9.
Thereafter, similarly to the above-described sixth embodiment, based on the correction signal from the Doppler correction means 9, the super-resolution time delay estimation processing unit 10 performs the time delay estimated value τk ^ (1 ≦ k ≦ K) of the target signal. Is required.

  As described above, according to the eighth embodiment of the present invention, since the S / N is improved by pulse compression, the detection performance of the target detection process performed by the super-resolution time delay estimation processing unit 10 can be improved. it can.

Embodiment 9 FIG.
In the eighth embodiment (see FIG. 20), the pulse compression type signal detection processing unit 22 that outputs the Doppler frequency observation value fd and the super-resolution time delay estimation processing unit 10 are used. As described above, the pulse compression type signal detection processing means 27 for outputting the target signal Doppler bin (the Doppler frequency observation value) and the time delay bin n τd similar to those in the fourth embodiment (see FIGS. 13 and 14), and the time A gate-type super-resolution time delay estimation processing unit 14 may be used.

  FIG. 21 is a block diagram showing a radar apparatus according to Embodiment 9 of the present invention. The transmitter 1, the transmission antenna 2, the target 3, the reception antenna 4, the receiver 5, and the like (see FIG. 20), A / D converter 6, pulse direction FFT 7 (# 1) to 7 (#Nr), Doppler correction means 9, time gate type super-resolution time delay estimation processing unit 14, H / L-PRF generator 30, Doppler frequency estimation About the means 31, the same code | symbol as above-mentioned is attached | subjected and detailed description is abbreviate | omitted.

Next, the operation according to the ninth embodiment of the present invention shown in FIG. 21 will be described with reference to FIG.
First, in the same manner as described above, HPRF and LPRF pulse transmission / reception is performed according to an instruction from the H / L-PRF generator 30, and the Doppler frequency estimation value fd ^ is input to the Doppler correction means 9, and the pulse direction FFT is also transmitted. The output signal is input to the pulse compression signal detection processing means 27 and the memory circuit 21.

Subsequently, a target signal Doppler bin (nf Doppler bin) and a time delay bin n τd (see FIG. 14) corresponding to the Doppler frequency observation value fd are output from the pulse compression signal detection processing means 27, and the target signal Doppler bin is output. Is input to the memory circuit 21, and the time delay bin n τd (see FIG. 14) is input to the time gate type super-resolution time delay estimation processing unit 14.
Also, a received signal of a target signal Doppler bin (nf Doppler bin) corresponding to the Doppler frequency observation value fd is input from the memory circuit 21 to the Doppler correction means 9.

Thereafter, as described above, based on the correction signal from the Doppler correction means 9, the time gate type super-resolution time delay estimation processing unit 14 determines the time delay estimated value τk ^ (1 ≦ k ≦ K) of the target signal. .
As described above, according to the ninth embodiment of the present invention, the processing load of super-resolution time delay estimation for Doppler correction can be reduced by limiting the time delay estimation range to the time gate range.
Further, since the S / N is improved by the pulse compression, it is possible to improve the detection performance of the target detection process performed in the super-resolution time delay estimation process.

Embodiment 10 FIG.
In the sixth embodiment (see FIG. 16), the H / L-PRF generator 30, the changeover switch 30a, and the Doppler frequency estimation means 31 are provided in the configuration of the first embodiment (see FIG. 1). The signal detection processing means 32 that outputs only the Doppler bin reception signal is used. As shown in FIG. 22, the multi-PRF switching means 36 and the Doppler ambiguity correction means 37 are added to the configuration described above (FIG. 1). It may be provided.

  FIG. 22 is a block diagram showing a radar apparatus according to Embodiment 10 of the present invention. The same transmitter 1, transmission antenna 2, target 3, reception antenna 4, receiver 5, A as those described above (FIG. 1). / D converter 6, pulse direction FFT 7 (# 1) to 7 (#Nr), signal detection processing means 8, Doppler correction means 9, and super-resolution time delay estimation processing unit 10 are assigned the same reference numerals as described above. Detailed description is omitted.

  In FIG. 22, the multi-PRF switching means 36 inputs a transmission switching instruction to the transmitter 1, and transmits the transmission pulse of the first pulse repetition period PRF 1 and the first pulse repetition period from the transmitter 1 and the transmission antenna 2. The transmission pulse of the second pulse repetition period PRF2 that is relatively prime to PRF1 is switched and transmitted. As a result, transmission / reception is performed by switching to two pulse repetition periods PRF1 and PRF2 that are in a prime relationship.

The Doppler ambiguity correction means 37 is connected to the multi-PRF switching means 36 and is inserted between the signal detection processing means 8 and the Doppler correction means 9 and together with the signal detection processing means 8 and the Doppler correction means 9, Doppler estimation. The processing unit is configured.
The Doppler ambiguity correction means 37 estimates the Doppler frequency by solving the ambiguity of the Doppler frequency, outputs the received signal of nf Doppler bin including the target signal, and the Doppler frequency estimated value fd ^, and performs the Doppler correction. Input to means 9.

  Specifically, when obtaining the Doppler frequency estimated value fd ^, the Doppler ambiguity correcting unit 37 receives the target signal obtained from the signal detection processing unit 8 when transmitting / receiving with the transmission pulse of the first pulse repetition period PRF1. The first Doppler frequency observation value fd1 and the second Doppler frequency observation value fd2 of the target signal obtained from the signal detection processing means 8 when transmission / reception is performed with a transmission pulse having the second pulse repetition period PRF2, the Doppler frequency To solve the ambiguity.

Next, the operation according to the tenth embodiment of the present invention shown in FIG. 22 will be described.
First, in response to an instruction from the multi-PRF switching means 36, the transmitter 1 and the transmission / reception antenna 2 transmit a radio wave subjected to pulse modulation at the first pulse repetition frequency f PRF1 . At this time, it is assumed that the first pulse repetition frequency f PRF1 is expressed by the following equation (11) with respect to a certain integer N PRF1 .

Hereinafter, the first Doppler frequency observation value fd1 of the target signal output from the signal detection processing means 8, and the first received signals z0 nf1 (1),..., Z0 nf1 (Nr) of the target signal Doppler bin. Is input to the Doppler ambiguity correcting means 37. Further, the first pulse repetition frequency f PRF1 is also input to the Doppler ambiguity correcting means 37.

Next, in response to an instruction from the multi-PRF switching unit 36, the transmitter 1 and the transmission / reception antenna 2 transmit the radio waves subjected to pulse modulation at the second pulse repetition frequency f PRF2 . At this time, it is assumed that the second pulse repetition frequency f PRF2 is expressed by the following equation (12) with respect to the integer N PRF2 that is relatively prime to the integer N PRF1 .

Hereinafter, the second Doppler frequency observation value fd2 of the target signal output from the signal detection processing unit 8, and the second received signals z0 nf2 (1),..., Z0 nf2 (Nr) of the target signal Doppler bin Is input to the Doppler ambiguity correcting means 37. The second pulse repetition frequency f PRF2 is also input from the multi-PRF switching unit 36 to the Doppler ambiguity correcting unit 37.
The Doppler ambiguity correcting means 37 first calculates the Doppler bin estimated value n τd ^ by the following equation (13).

However, mod ( nτd1 · N PRF2 · N ′ PRF2 + n τd2 · N PRF1 · N ′ PRF1, N PRF1 · N PRF2 ) in the expression (13) is an integer “n τd1 · N PRF2 · N ′ PRF2 + n τd2 · N PRF1 · N ′ PRF1 ”is a function that outputs a“ remainder ”when the integer“ N PRF1 · N PRF2 ”is divided.
Further, n τd1 represents a Doppler bin corresponding to the first Doppler frequency observation value fd1, and n τd2 represents a Doppler bin corresponding to the second Doppler frequency observation value fd2. N ′ PRF1 is an integer that satisfies the following expression (14).

Similarly, N ′ PRF2 is an integer that satisfies the following formula (15).

  Next, the Doppler ambiguity correcting unit 37 calculates the Doppler frequency estimated value fd ^ by the following equation (16).

Subsequently, the Doppler ambiguity correcting unit 37 receives the first received signal z0 nf1 (1),..., Z0 nf1 (Nr) corresponding to the target signal Doppler bin, or the second received signal z0 nf2 ( 1),..., Z0 nf2 (Nr) are output as output signals z0 nf (1),..., Z0 nf (Nr). The Doppler ambiguity correcting means 37 also outputs a Doppler frequency estimated value fd ^.

The received signals z0 nf (1),..., Z0 nf (Nr) and the Doppler frequency estimation value fd ^ are input to the Doppler correction means 9 and input to the super-resolution time delay estimation processing unit 10 as a correction signal. The Thereafter, the super-resolution time delay estimation processing unit 10 determines the time delay estimated value τk ^ (1 ≦ k ≦ K) of the target signal in the same manner as described above.

  As described above, the radar apparatus according to Embodiment 10 of the present invention includes a transmitter 1 that generates radio waves, a transmission antenna 2 that transmits radio waves, and a radio wave that is transmitted from the transmission antenna 2 and reflected by the target 3. A receiving antenna 4 that receives the signal as a received wave, a receiver 5 that performs band limitation and phase detection of the received wave to generate a target signal corresponding to the target, and a Doppler frequency of the target signal due to the Doppler effect resulting from the movement of the target 3 The Doppler estimation processing unit for estimating the Doppler frequency estimation value fd ^, the time delay estimation processing unit for correcting the phase rotation of the target signal due to the Doppler effect and estimating the time delay of the target signal, and the transmitter 1 When a transmission switching instruction is input, the transmitter 1 transmits the transmission pulse of the first pulse repetition period PRF1 and the first pulse repetition period PRF1 to each other. Of a transmission pulse of the second pulse repetition period PRF2 in a relationship, and a multi-PRF switching means 36 for transmitting by switching.

  The Doppler estimation processing unit includes Doppler ambiguity correction means 37. The Doppler ambiguity correction means 37 receives the first target signal obtained when transmission / reception is performed with a transmission pulse having the first pulse repetition period PRF1. By solving the Doppler frequency ambiguity from the Doppler frequency observation value fd1 and the second Doppler frequency observation value fd2 of the target signal obtained when transmission / reception is performed with a transmission pulse having the second pulse repetition period PRF2, Doppler frequency is obtained. The frequency estimated value fd ^ is obtained.

  As a result, according to the tenth embodiment of the present invention, Doppler having no ambiguity is obtained by estimating the Doppler frequency using the first and second pulse repetition periods PRF1 and PRF2 that are in a prime relationship with each other. The frequency can be estimated, and the accuracy of Doppler correction of the target signal can be improved.

Embodiment 11 FIG.
In the tenth embodiment (see FIG. 22), the same signal detection processing means 8 and super-resolution time delay estimation processing section 10 as those in the first embodiment (see FIGS. 1 and 3) are used. As shown in FIG. 23, the same signal detection processing means 13 and time gate type super-resolution time delay estimation processing unit 14 as in the second embodiment (see FIGS. 5 and 6) may be used.

  FIG. 23 is a block diagram showing a radar apparatus according to Embodiment 11 of the present invention. The same transmitter 1, transmitting antenna 2, target 3, receiving antenna 4, receiver 5, A as those described above (FIG. 22). The / D converter 6, the pulse direction FFTs 7 (# 1) to 7 (#Nr), the Doppler correction unit 9, and the multi-PRF switching unit 36 are assigned the same reference numerals as those described above and will not be described in detail.

In FIG. 23, the Doppler ambiguity correcting means 38 takes in the three types of signals from the signal detection processing means 13, estimates the Doppler frequency by solving the ambiguity of the Doppler frequency, and receives the Doppler bin including the target signal. signal z0 nf (1), ···, z0 nf and (Nr), the Doppler frequency estimate fd ^, and outputs the time delay bin n .tau.d target signal.

Next, the operation according to the eleventh embodiment of the present invention shown in FIG. 23 will be described.
First, similarly to the above (FIG. 22), when a radio wave subjected to pulse modulation is transmitted at the first pulse repetition frequency f PRF1 , the signal detection processing unit 13 outputs the first Doppler frequency observation value of the target signal. fd1, the first received signal z0 nf1 (1),..., z0 nf1 (Nr) of the target signal Doppler bin and the time delay bin n τd1 of the target signal are input to the Doppler ambiguity correcting means 38. Is done. The first pulse repetition frequency f PRF1 is also input from the multi-PRF switching unit 36 to the Doppler ambiguity correcting unit 38.

Subsequently, when the pulse-modulated radio wave is transmitted at the second pulse repetition frequency fPRF2, the signal detection processing unit 13 outputs the second Doppler frequency observation value fd2 of the target signal and the first of the target signal Doppler bins. 2 received signals z0 nf2 (1),..., Z0 nf2 (Nr) and the time delay bin n τd2 of the target signal are input to the Doppler ambiguity correcting means 38. The second pulse repetition frequency f PRF2 is also input from the multi-PRF switching unit 36 to the Doppler ambiguity correcting unit 38.

Subsequently, the Doppler ambiguity correction means 38 receives the signal output from z0 nf (1), ···, z0 nf and (Nr), and time delay bin n .tau.d, of the Doppler frequency estimate fd ^, The Doppler frequency estimated value fd ^ and the received signals z0 nf (1),..., Z0 nf (Nr) are input to the Doppler correction means 9, and the time delay bin n τd is a time gate type super resolution time delay estimation process. Input to the unit 14. Thereafter, based on the correction signal from the Doppler correction means 9, the time gate type super resolution time delay estimation processing unit 14 obtains the target signal delay time estimated value τk ^ (1 ≦ k ≦ K) in the same manner as described above. It is done.

  As described above, according to the eleventh embodiment of the present invention, by limiting the time delay estimation range to the time gate range, the processing load of super-resolution time delay estimation that performs Doppler correction in consideration of ambiguity is reduced. Can be made.

Embodiment 12 FIG.
In the tenth embodiment (see FIG. 22), the same signal detection processing means 8 as in the first embodiment (see FIGS. 1 and 3) is used. However, as shown in FIG. The same pulse compression type signal detection processing means 22 as that of the third embodiment (see FIGS. 10 and 11) may be used, and a memory circuit 21 may be provided.

  FIG. 24 is a block diagram showing a radar apparatus according to Embodiment 12 of the present invention. The same transmitter 1, transmitting antenna 2, target 3, receiving antenna 4, receiver 5, A as those described above (FIG. 22). / D converter 6, pulse direction FFT 7 (# 1) to 7 (#Nr), Doppler correction means 9, super-resolution time delay estimation processing unit 10, multi-PRF switching means 36, Doppler ambiguity correction means 37 The same reference numerals as those described above are attached and detailed description is omitted.

Next, the operation according to the twelfth embodiment of the present invention shown in FIG. 24 will be described.
First, when a pulse-modulated radio wave is transmitted at the first pulse repetition frequency f PRF1 , the first Doppler frequency observation value fd1 of the target signal is obtained from the pulse compression type signal detection processing means 22 as a Doppler ambiguity. Input to the tee correction means 37. The first pulse repetition frequency f PRF1 is input to the Doppler ambiguity correction unit 37 from the multi-PRF switching unit 36, and the first received signal z0 nf1 (1) of the target signal Doppler bin is output from the memory circuit 21. ,..., Z0 nf1 (Nr) is input.

Subsequently, when the pulse-modulated radio wave is transmitted at the second pulse repetition frequency f PRF2 , the second Doppler frequency observation value fd2 of the target signal is obtained from the pulse compression type signal detection processing unit 22. It is input to the Guiti correction means 37. Further, the second pulse repetition frequency f PRF2 is input from the multi-PRF switching unit 36 to the Doppler ambiguity correcting unit 37, and the second received signal z0 of the Doppler bin in which the target signal is present from the memory circuit 21. nf2 (1),..., z0 nf2 (Nr) is input.

As a result, the Doppler ambiguity correcting unit 37 obtains the Doppler frequency estimated value fd ^ of the target signal, and the Doppler frequency estimated value fd ^ and the received signals z0 nf (1), ..., z0 nf (Nr) are obtained. Input to the Doppler correction means 9. Thereafter, based on the correction signal from the Doppler correction means 9, the super-resolution time delay estimation processing unit 10 obtains the time delay estimated value τk ^ (1 ≦ k ≦ K) of the target signal as described above.

  As described above, according to the twelfth embodiment of the present invention, since the S / N is improved by pulse compression, the detection performance of the target detection process performed in the super-resolution time delay estimation process can be improved.

Embodiment 13 FIG.
In the twelfth embodiment (see FIG. 24), the same pulse compression type signal detection processing means 22 as in the third embodiment (see FIGS. 10 and 11) and the tenth embodiment (see FIG. 22). The same Doppler ambiguity correcting means 37 and super-resolution time delay estimation processing unit 10 as in FIG. 25 are used. As shown in FIG. 25, the same as in the above-described fourth embodiment (see FIGS. 13 and 14). The pulse compression type signal detection processing unit 27 of the above, the Doppler ambiguity correction unit 38 and the time gate type super resolution time delay estimation processing unit 14 similar to those of the above-described eleventh embodiment (see FIG. 23) may be used. .

  FIG. 25 is a block diagram showing a radar apparatus according to Embodiment 13 of the present invention. The same transmitter 1, transmitting antenna 2, target 3, receiving antenna 4, receiver 5, A as those described above (FIG. 24). The / D converter 6, the pulse direction FFTs 7 (# 1) to 7 (#Nr), the Doppler correction means 9, the memory circuit 21, and the multi-PRF switching means 36 are assigned the same reference numerals as those described above and will not be described in detail. .

Next, the operation according to the thirteenth embodiment of the present invention shown in FIG. 25 will be described.
First, when a pulse-modulated radio wave is transmitted at the first pulse repetition frequency f PRF1 , the pulse compression type signal detection processing unit 27 outputs the first Doppler frequency observation value fd1 of the target signal and the first time. The delay bin n τd1 is input to the Doppler ambiguity correcting means 38. The Doppler ambiguity correcting means 38 receives the first pulse repetition frequency f PRF1 from the multi-PRF switching means 36 and the first received signal (1) of the target signal Doppler bin from the memory circuit 21. ..., z0 nf1 (Nr) is input.

Subsequently, when the pulse-modulated radio wave is transmitted at the second pulse repetition frequency f PRF2 , the pulse compression type signal detection processing unit 27 sends the second Doppler frequency observation value fd2 of the target signal and the second The time delay bin n τd2 is input to the Doppler ambiguity correcting means 38. The second pulse repetition frequency f PRF2 is input from the multi-PRF switching unit 36 to the Doppler ambiguity correcting unit 38, and the second received signal z0 nf2 (1) of the target signal Doppler bin from the memory circuit 21. ,..., Z0 nf2 (Nr) is input.

As a result, the Doppler ambiguity correcting unit 38 obtains the Doppler frequency estimated value fd ^ of the target signal, and the Doppler frequency estimated value fd ^ and the received signals z0 nf (1), ..., z0 nf (Nr) are obtained. Input to the Doppler correction means 9. Thereafter, based on the correction signal from the Doppler correction means 9, the time gate type super-resolution time delay estimation processing unit 14 obtains the time delay estimated value τk ^ (1 ≦ k ≦ K) of the target signal as described above. It is done.

As described above, according to the thirteenth embodiment of the present invention, by limiting the time delay estimation range to the time gate range, the processing load of super-resolution time delay estimation that performs Doppler correction in consideration of ambiguity is reduced. Can be made.
Further, since the S / N is improved by the pulse compression, it is possible to improve the detection performance of the target detection process performed in the super-resolution time delay estimation process.

Embodiment 14 FIG.
In the tenth embodiment (see FIG. 22), the multi-PRF switching means 36 and the Doppler ambiguity correcting means 37 are provided in the configuration of the first embodiment (see FIG. 1). As shown, the transmission frequency switching means 39 and the Doppler ambiguity correction means 40 may be provided in the configuration described above (FIG. 1).

  26 is a block diagram showing a radar apparatus according to Embodiment 14 of the present invention. The same transmitter 1, transmitting antenna 2, target 3, receiving antenna 4, receiver 5, A as those described above (FIG. 1). / D converter 6, pulse direction FFT 7 (# 1) to 7 (#Nr), signal detection processing means 8, Doppler correction means 9, and super-resolution time delay estimation processing unit 10 are assigned the same reference numerals as described above. Detailed description is omitted.

  In FIG. 26, the transmission frequency switching means 39 inputs a transmission switching instruction to the transmitter 1, and receives two transmission frequencies (first and second) whose wavelengths are coprime from the transmitter 1 and the transmission antenna 2. Are switched and transmitted. Here, the two transmission frequencies are represented by c / λ1 and c / λ2 using the first and second wavelengths λ1 and λ2 and the speed of light c.

The Doppler ambiguity correction means 40 is connected to the transmission frequency switching means 39 and is inserted between the signal detection processing means 8 and the Doppler correction means 9 and together with the signal detection processing means 8 and the Doppler correction means 9, Doppler estimation. The processing unit is configured.
The Doppler ambiguity correcting means 40 solves the Doppler frequency ambiguity, estimates the Doppler frequency, and outputs the received signal of the target signal Doppler bin and the Doppler frequency estimated value fd ^.

  Specifically, the Doppler ambiguity correcting unit 40 transmits the first Doppler frequency observation value fd1 of the target signal obtained when transmission / reception is performed with the transmission pulse of the first wavelength λ1, and the transmission pulse of the second wavelength λ2. The Doppler frequency estimation value fd ^ is obtained by solving the ambiguity of the Doppler frequency from the second Doppler frequency observation value fd2 of the target signal obtained when transmitting and receiving at. Here, for the sake of convenience, the first and second Doppler frequency observation values fd1 and fd2 are denoted by the same reference numerals as those in the above-described tenth to thirteenth embodiments. And

Next, the operation according to the fourteenth embodiment of the present invention shown in FIG. 26 will be described.
First, in response to an instruction from the transmission frequency switching means 39, a radio wave subjected to pulse modulation is transmitted from the transmitter 1 and the transmission antenna 2 at the first transmission frequency c / λ1 (first wavelength λ1). Is done. Here, it is assumed that the first wavelength λ1 is expressed by the following equation (17) using a certain unit δλ and an integer Nλ1.

Subsequently, the first detection signal fd1 of the target signal and the first received signal z0 nλ1 (1) of the target signal Doppler bin based on the first wavelength λ1 from the signal detection processing means 8. z0 nλ1 (Nr) is input to the Doppler ambiguity correction means 40. Here, the first Doppler frequency observations fd1 is the following equation (18) is determined as the first target speed observation value n .lambda.1 corresponding to the unit [delta] [lambda].

  However, floor [] in equation (18) represents a function that rounds off the value in [].

  Next, in response to the instruction from the transmission frequency switching means 39, the radio waves subjected to pulse modulation are transmitted from the transmitter 1 and the transmission antenna 2 at the second transmission frequency c / λ2 (second wavelength λ2). Sent. Here, it is assumed that the second wavelength λ2 is expressed by the following equation (19) using the unit δλ and the integer Nλ2.

However, in the equation (19), the integer Nλ2 and the integer Nλ1 are relatively prime.
Subsequently, the second detection signal fd2 of the target signal and the second received signal z0 nλ2 (1) of the target signal Doppler bin based on the second wavelength λ2 from the signal detection processing means 8. z0 nλ2 (Nr) is input to the Doppler ambiguity correcting means 40. The second Doppler frequency observations fd2 is the following equation (20), is determined as a second target speed observation value n .lambda.2 corresponding to the unit [delta] [lambda].

Next, based on the input signal, the Doppler ambiguity correcting unit 40 first calculates a target speed estimated value n λ ^ as a unit δλ by the following equation (21).

  However, N′λ1 in the equation (21) is an integer satisfying the following equation (22).

  Also, N′λ2 in the equation (21) is an integer that satisfies the following equation (23).

  Next, the Doppler ambiguity correction unit 40 calculates the Doppler frequency estimated value fd ^ by the following equation (24).

Then, first received signals z0 nf1 (1),..., Z0 nf1 (Nr) corresponding to the target signal Doppler bin are output and input to the Doppler correction means 9. Here, the output signal of the Doppler ambiguity correcting means 40 is expressed as z0 nf (1),..., Z0 nf (Nr).
Further, the Doppler ambiguity correction unit 40 outputs the Doppler frequency estimated value fd ^ and inputs it to the Doppler correction unit 9.

Hereinafter, the Doppler correction means 9 outputs a correction signal based on the input signals (received signal z0 nf (1),..., Z0 nf (Nr) and Doppler frequency estimated value fd ^) to estimate the super-resolution time delay. The super-resolution time delay estimation processing unit 10 inputs to the processing unit 10 and obtains the time delay estimation value of the target signal as in the first embodiment.

The Doppler ambiguity correcting unit 40 obtains the Doppler frequency estimated value fd ^ based on the first wavelength λ1 by the above equation (24), but the second wavelength λ2 is obtained by the following equation (25). Based on this, the Doppler frequency estimated value fd ^ may be obtained and output together with the first received signals z0 nf1 (1), ..., z0 nf1 (Nr).

  As described above, the radar apparatus according to Embodiment 14 of the present invention includes the transmission frequency switching means 39 in addition to the configuration described above (see FIG. 1). To switch the transmission pulse of the first wavelength λ1 from the transmitter 1 to the transmission pulse of the second wavelength λ2 that is relatively prime to the first wavelength λ1. It is supposed to be sent.

  The Doppler estimation processing unit includes Doppler ambiguity correction means 40. The Doppler ambiguity correction means 40 is a first Doppler frequency of a target signal obtained when transmission / reception is performed with a transmission pulse of the first wavelength λ1. The Doppler frequency estimated value fd is obtained by solving the ambiguity of the Doppler frequency from the observed value fd1 and the second Doppler frequency observed value fd2 of the target signal obtained when transmission / reception is performed with the transmission pulse of the second wavelength λ2. Seek ^.

  Therefore, according to the fourteenth embodiment of the present invention, it is possible to estimate the Doppler frequency without ambiguity by estimating the Doppler frequency using the two types of wavelengths λ1 and λ2 that are relatively prime to each other. The Doppler correction accuracy of the target signal can be improved.

Embodiment 15 FIG.
In the fourteenth embodiment (see FIG. 26), the transmission frequency switching means 39 and the Doppler ambiguity correcting means 40 are provided in the configuration of FIG. 1, but as shown in FIG. The transmission frequency switching means 39 and the Doppler ambiguity correction means 41 may be provided in the configuration (see FIG. 5).

  FIG. 27 is a block diagram showing a radar apparatus according to Embodiment 15 of the present invention. The same transmitter 1, transmitting antenna 2, target 3, receiving antenna 4, receiver 5, A as those described above (FIG. 5). / D converter 6, pulse direction FFT 7 (# 1) to 7 (#Nr), signal detection processing means 13, Doppler correction means 9, and time gated super-resolution time delay estimation processing unit 14 have the same symbols as above. A detailed description will be omitted. The transmission frequency switching means 39 is the same as that described above (FIG. 26).

  In FIG. 27, the Doppler ambiguity correcting means 41 estimates the Doppler frequency by solving the ambiguity of the Doppler frequency, the received signal of the target signal Doppler bin, the Doppler frequency estimated value fd ^, and the time delay of the target signal. Bin and output. The received signal of the target signal Doppler bin and the Doppler frequency estimation value fd ^ are input to the Doppler correction means 9, and the time delay bin of the target signal is input to the time gated super-resolution time delay estimation processing unit 14.

Next, the operation according to Embodiment 15 of the present invention shown in FIG. 27 will be described.
First, in the same manner as described above, in response to an instruction from the transmission frequency switching means 39, a radio wave subjected to pulse modulation of the first transmission frequency c / λ1 is transmitted from the transmitter 1 and the transmission antenna 2.

As a result, the signal detection processing unit 13 gives the Doppler ambiguity correction unit 41 the first Doppler frequency observation value fd1 of the target signal and the received signal z0 nλ1 (1) of the target signal Doppler bin. , Z0 nλ1 (Nr) and the time delay bin n λ1 of the target signal are input. The transmission wavelength setting value (first wavelength λ1) is input to the Doppler ambiguity correction unit 41 from the transmission frequency switching unit 39.

Subsequently, in response to an instruction from the transmission frequency switching means 39, a radio wave subjected to pulse modulation of the second transmission frequency c / λ2 is transmitted from the transmitter 1 and the transmission antenna 2.
As a result, the signal detection processing means 13 gives the Doppler ambiguity correction means 41 the second Doppler frequency observation value fd2 of the target signal and the received signal z0 nλ2 (1) of the target signal Doppler bin. , Z0 nλ2 (Nr) and the time delay bin n λ2 of the target signal are input. The transmission wavelength setting value (first wavelength λ2) is input from the transmission frequency switching means 39 to the Doppler ambiguity correction means 41.

The Doppler ambiguity correcting means 41 is configured to calculate the first received signal z0 nλ1 (1),..., Z0 nλ1 (Nr) corresponding to the target signal Doppler bin and the first time delay bin n λ1 of the target signal. Either a combination or a combination of the second received signal z0 nλ2 (1),..., Z0 nλ2 (Nr) corresponding to the target signal Doppler bin and the second time delay bin n λ2 of the target signal Output one combination.

At this time, from the Doppler ambiguity correcting unit 41, the received signal z0 nλ (1) of the target signal Doppler bins, · · ·, z0 n [lambda and (Nr), and time delay bin n .tau.d, Doppler frequency estimate fd ^ , And the Doppler frequency estimation value fd ^ and the received signals z0 nf (1),..., Z0 nf (Nr) are input to the Doppler correction means 9, and the time delay bin n τd is a time gated super-resolution time. Input to the delay estimation processing unit 14.

Thereafter, as in the second embodiment (see FIG. 5) described above, the target signal delay time estimated value τk ^ is obtained.
As described above, according to 15 as described above, by limiting the time delay estimation range to the time gate range, it is possible to reduce the processing load of super-resolution time delay estimation that performs Doppler correction in consideration of ambiguity. it can.

Embodiment 16 FIG.
In the fourteenth embodiment (see FIG. 26), the transmission frequency switching means 39 and the Doppler ambiguity correcting means 40 are provided in the configuration shown in FIG. 1, but as shown in FIG. The transmission frequency switching means 39 and the Doppler ambiguity correction means 40 may be provided in the configuration (see FIG. 10).

  FIG. 28 is a block diagram showing a radar apparatus according to Embodiment 16 of the present invention. The same transmitter 1, transmission antenna 2, target 3, reception antenna 4, receiver 5, A as those described above (FIG. 10). The / D converter 6, the pulse direction FFT 7 (# 1) to 7 (#Nr), the Doppler correction unit 9, the super-resolution time delay estimation processing unit 10, the memory circuit 21, and the pulse compression signal detection processing unit 22 are described above. The same reference numerals are assigned and detailed description is omitted. The transmission frequency switching means 39 and the Doppler ambiguity correction means 40 are the same as those described above (FIG. 26).

Next, the operation according to the sixteenth embodiment of the present invention shown in FIG. 28 will be described.
First, a radio wave subjected to pulse modulation of the first transmission frequency c / λ1 is transmitted from the transmitter 1 and the transmission antenna 2, and the Doppler ambiguity correction unit 40 is transmitted from the pulse compression signal detection processing unit 22 to the Doppler ambiguity correction unit 40. Then, the first Doppler frequency observation value fd1 of the target signal and the received signals z0 nλ1 (1),..., Z0 nλ1 (Nr) of the target signal Doppler bin are input. At this time, the transmission wavelength setting value (first wavelength λ1) from the transmission frequency switching means 39 is also input to the Doppler ambiguity correction means 40.

Subsequently, the transmitter 1 and the transmission antenna 2 transmit a radio wave subjected to pulse modulation of the second transmission frequency c / λ2, and the pulse compression type signal detection processing means 22 sends it to the Doppler ambiguity correction means 40. On the other hand, the second Doppler frequency observation value fd2 of the target signal and the received signals z0 nλ2 (1),..., Z0 nλ2 (Nr) of the target signal Doppler bin are input. At this time, the transmission wavelength setting value (second wavelength λ2) from the transmission frequency switching means 39 is also input to the Doppler ambiguity correction means 40.

Thereafter, as described above, the time delay estimated value τk ^ of the target signal is obtained, and the Doppler frequency estimated value fd ^ and the received signals z0 (1),..., Z0 (Nr) are supplied to the Doppler correction means 9. The super-resolution time delay estimation processing unit 10 receives the delay time estimated value τk ^ of the target signal.
As described above, according to the sixteenth embodiment of the present invention, since the S / N is improved by the pulse compression processing in the pulse compression type signal detection processing means 22, the super resolution time delay estimation processing unit 10 executes The detection performance of the target detection process to be performed can be improved.

Embodiment 17. FIG.
In the fifteenth embodiment (see FIG. 27), the transmission frequency switching means 39 and the Doppler ambiguity correcting means 41 are provided in the configuration of the second embodiment (FIG. 5), but as shown in FIG. In addition, the transmission frequency switching means 39 and the Doppler ambiguity correction means 41 may be provided in the configuration of the above-described fourth embodiment (see FIG. 13).

  29 is a block diagram showing a radar apparatus according to Embodiment 17 of the present invention. The same transmitter 1, transmitting antenna 2, target 3, receiving antenna 4, receiver 5, A as those described above (FIG. 13). / D converter 6, pulse direction FFT 7 (# 1) to 7 (#Nr), Doppler correction means 9, time gate type super-resolution time delay estimation processing section 14, memory circuit 21, pulse compression type signal detection processing means 27 Are denoted by the same reference numerals as those described above and will not be described in detail. The transmission frequency switching means 39 and the Doppler ambiguity correction means 41 are the same as those described above (FIG. 27).

Next, the operation according to Embodiment 17 of the present invention shown in FIG. 29 will be described.
First, in the same manner as described above, based on transmission / reception of a pulse-modulated radio wave having the first transmission frequency c / λ1, the first signal of the target signal is transmitted from the pulse compression signal detection processing unit 27 to the Doppler ambiguity correction unit 41. The Doppler frequency observation value fd1 and the time delay bin n τd1 are input. The transmission wavelength setting value (first wavelength λ1) from the transmission frequency switching means 39 is input to the Doppler ambiguity correction means 41.

Subsequently, the second Doppler frequency observation of the target signal is performed from the pulse compression type signal detection processing unit 27 to the Doppler ambiguity correction unit 41 based on transmission / reception of the pulse modulated radio wave having the second transmission frequency c / λ2. The value fd2 and the time delay bin n τd2 are input. Further, the transmission wavelength setting value (second wavelength λ2) from the transmission frequency switching means 39 is input to the Doppler ambiguity correction means 41.

Hereinafter, similarly to the above-described fifteenth embodiment (FIG. 27), the Doppler ambiguity correction means 41 receives received signals z0 (1),..., Z0 (Nr) corresponding to the target signal Doppler bin. The time delay bin n τd and the Doppler frequency estimate fd ^ are output.
Among these, the Doppler frequency estimated value fd ^ and the received signals z0 nf (1),..., Z0 nf (Nr) are input to the Doppler correction means 9, and the time delay bin n τd is a time gated super-resolution time delay. Input to the estimation processing unit 14.
Thereafter, the delay time estimated value τk ^ of the target signal is obtained in the same manner as in the above-described fourth embodiment (FIG. 13).

As described above, according to the seventeenth embodiment of the present invention, by limiting the time delay estimation range to the time gate range, the processing load of super-resolution time delay estimation that performs Doppler correction in consideration of ambiguity is reduced. Can be made.
Further, since the S / N is improved by pulse compression, the detection performance of the target detection process performed in the super-resolution time delay estimation process can be improved.

Embodiment 18 FIG.
In the first embodiment (see FIG. 1), the evaluation of the time delay estimated value τk ^ and the ambiguity correction based on the evaluation result are not considered, but the time delay estimated value evaluation is performed as shown in FIG. Means 43 may be provided, and an ambiguity correction instruction signal may be input to the Doppler correction means 42 according to the evaluation result.

  30 is a block diagram showing a radar apparatus according to Embodiment 18 of the present invention. The same transmitter 1, transmitting antenna 2, target 3, receiving antenna 4, receiver 5, A as those described above (FIG. 1). The / D converter 6, the pulse direction FFTs 7 (# 1) to 7 (#Nr), the signal detection processing unit 8, and the super-resolution time delay estimation processing unit 10 are assigned the same reference numerals as those described above and will not be described in detail. .

  In FIG. 30, instead of the Doppler correction means 9 described above, a Doppler correction means 42 considering ambiguity is provided, and an ambiguity estimation section is included on the output side of the super-resolution time delay estimation processing section 10. Time delay estimated value evaluation means 43 is provided.

Time delay estimation value evaluating means 43, super-resolution time delay input from the estimation processing unit 10 delay estimates .tau.k ^, to assess the accuracy of n amb, relative Doppler compensation means 42, ambiguity correction instruction signal Enter.

Further, the ambiguity estimation unit in the time delay estimation value evaluation unit 43 uses the Doppler frequency ambiguity as an unknown parameter, and calculates the peak value of the MUSIC spectrum calculated in the process of the super-resolution time delay estimation processing unit 10. The ambiguity is estimated as the evaluation function value for the unknown parameter.
The Doppler correction unit 42 corrects the phase rotation of the target signal due to the Doppler effect in consideration of the ambiguity based on the ambiguity correction instruction signal.

Next, the operation according to the eighteenth embodiment of the present invention shown in FIG. 30 will be described.
First, as described above, when radio waves subjected to pulse modulation are transmitted from the transmitter 1 and the transmission antenna 2, the signal detection processing means 8 receives the Doppler frequency observation value fd of the target signal and the received signal of the target signal Doppler bin. z0 nf (1),..., z0 nf (Nr) are output and input to the Doppler correction means 42.

Subsequently, the Doppler correction means 42 initially sets the ambiguity number n amb to an initial value (= 0), and receives the received signal z0 nf (1),..., Z0 nf according to the following equation (26). The phase rotation of the target signal component included in (Nr) is corrected.

Note that the value of the ambiguity number n amb changes according to the ambiguity correction instruction signal from the time delay estimated value evaluation means 43 thereafter.
Ambiguity Doppler compensation signal z0 in tee number n amb 'nf, namb (1 ), ···, z0' nf, namb (Nr) is input to the super-resolution time delay estimation processing unit 10 and the time delay estimation evaluating means 43 Is done.

Thereafter, in the super-resolution time delay estimation processing unit 10, the time delay estimated value τk ^, n amb (1 ≦ k ≦ K) in the ambiguity number n amb is calculated as in the first embodiment. .

Time delay estimation value evaluating means 43, the Doppler correction signal z0 in ambiguity number n amb input from the Doppler correction means 42 'nf, namb (1) , ···, z0' nf, and nAmb (Nr), Based on the estimated time delay value τk ^, n amb (1 ≦ k ≦ K) in the ambiguity number n amb input from the super-resolution time delay estimation processing unit 10, first, the correlation matrix is expressed by the above equation (2). R is calculated.

Further, the time delay estimated value evaluation means 43 performs the time delay estimated value τk ^, n amb (1 ≦ 1) at the ambiguity number n amb with respect to the steering vector a (τ) represented by the above-described equation (3). Substituting k ≦ K), a (τ1 ^, n amb ),..., a (τK ^, n amb ) are obtained. Further, the time delay estimated value evaluation means 43 generates a matrix Rnoise from the eigenvectors e (K + 1),..., E (Md) corresponding to the noise eigenvalues according to the following equation (27).

Further, the time delay estimated value evaluating means 43 is an evaluation function value I (n amb ) representing the orthogonality between the steering vector a (τ1 ^, n amb ),..., A (τK ^, n amb ) and the matrix Rnoise. ) Is calculated by the following equation (28).

Next, the time delay estimated value evaluating means 43 compares the ambiguity number n amb with the assumed total ambiguity number Namb, and if n amb <Namb, the ambiguity number n amb is expressed as “ It is incremented to “n amb ← n amb +1” and input to the Doppler correction means 42. In response to this, the Doppler correction means 42 performs phase rotation of the target signal component included in the received signals z0 nf (1),..., Z0 nf (Nr) based on the input ambiguity number n amb. Then, signals z0 ′ nf (1),..., Z0 ′ nf (Nr) are generated.

In the same manner, the received signals z0 ′ nf (1),..., Z0 ′ nf (Nr) and the time delay estimated value τk ^ (1 ≦ k ≦ K) are input to the time delay estimated value evaluating means 43. The time delay estimated value evaluation means 43 calculates the evaluation function value I (n amb ) again by the equation (28).
By repeatedly executing the above processing, the evaluation function values I (1), I (2),..., I (Namb) are calculated, and the evaluation function values I (1), I (2),. • The ambiguity number n amb that maximizes I (Namb) is examined.

When the number of ambiguities obtained as the processing result is expressed as "n amb 0" super-resolution time delay estimation processor 10, the time delay estimates τk in ambiguity number n amb 0 ^, n amb 0 ( 1 ≦ k ≦ K) is output as the final time delay estimated value τk ^ (1 ≦ k ≦ K) as in the following equation (29).

As described above, the radar apparatus according to Embodiment 18 of the present invention evaluates the accuracy of the time delay estimation value τk ^, n amb input from the super-resolution time delay estimation processing unit 10 (time delay estimation processing unit). A time delay estimation value evaluation unit 43 including an ambiguity estimation unit for performing the estimation, and the ambiguity estimation unit in the time delay estimation value evaluation unit 43 uses the ambiguity of the Doppler frequency as an unknown parameter and estimates the time delay. The ambiguity is estimated by using the peak value of the MUSIC spectrum calculated in the process of the processing unit as the evaluation function value for the unknown parameter.
Thereby, the ambiguity of the Doppler frequency can be searched brute force, and the Doppler frequency can be estimated without being affected by the ambiguity.

Embodiment 19. FIG.
In the eighteenth embodiment (see FIG. 30), the Doppler correction means 42 and the time delay estimated value evaluation means 43 are applied to the configuration of the first embodiment (FIG. 1), but as shown in FIG. The Doppler correction unit 42 and the time delay estimated value evaluation unit 43 may be applied to the configuration of the above-described second embodiment (see FIG. 5).

  FIG. 31 is a block diagram showing a radar apparatus according to Embodiment 19 of the present invention. The same transmitter 1, transmission antenna 2, target 3, reception antenna 4, receiver 5, A as those described above (FIG. 5). The / D converter 6, the pulse direction FFT 7 (# 1) to 7 (#Nr), the signal detection processing unit 13, and the time gate type super-resolution time delay estimation processing unit 14 are described in detail with the same reference numerals. Is omitted. Further, the Doppler correction means 42 and the time delay estimated value evaluation means 43 are the same as those described above (FIG. 30).

Next, the operation of the nineteenth embodiment of the present invention shown in FIG. 31 will be described.
First, similarly to the above (FIG. 5), when radio waves subjected to pulse modulation are transmitted from the transmitter 1 and the transmission antenna 2, the radio waves reflected by the target 3 are transmitted via the reception antenna 4 and the receiver 5. The received signal of the target signal Doppler bin, the Doppler frequency observation value, and the time delay bin of the target signal are output from the signal detection processing means 13.

Thereafter, as described above (FIG. 30), the evaluation and correction processing by the Doppler correction means 42 and the time delay estimated value evaluation means 43 is repeated, and the time gated super-resolution time delay difference estimation processing unit 14 performs the Doppler correction signal. Then, based on the time delay bin, the final target signal time delay estimate fd ^ is output.
As described above, according to the nineteenth embodiment of the present invention, in addition to the operational effects of the eighteenth embodiment described above, the time delay estimation range is limited to the time gate range, so that ambiguity is taken into consideration. The processing load of super-resolution time delay estimation for Doppler correction can be reduced.

Embodiment 20. FIG.
In the nineteenth embodiment (see FIG. 31), the Doppler correction means 42 and the time delay estimated value evaluation means 43 are applied to the configuration of the above-described second embodiment (FIG. 5), but as shown in FIG. The Doppler correction unit 42 and the time delay estimated value evaluation unit 43 may be applied to the configuration of the above-described third embodiment (see FIG. 10).

  FIG. 32 is a block diagram showing a radar apparatus according to Embodiment 20 of the present invention. The same transmitter 1, transmitting antenna 2, target 3, receiving antenna 4, receiver 5, A as those described above (FIG. 10). / D converter 6, pulse direction FFT 7 (# 1) to 7 (#Nr), super resolution time delay estimation processing unit 10, memory circuit 21, and pulse compression type signal detection processing means 22 are assigned the same reference numerals as described above. Detailed description is omitted. The Doppler correction means 42 and the time delay estimated value evaluation means 43 are the same as those described above (FIGS. 30 and 31).

Next, the operation of the twentieth embodiment of the present invention shown in FIG. 32 will be described.
First, similarly to the above (FIG. 10), when a pulse is transmitted from the transmitter 1 and the transmission antenna 2, the radio wave reflected by the target 3 is received by the reception antenna 4 and the receiver 5, and the pulse compression signal The detection processing means 22 outputs a Doppler frequency observation value, and the memory circuit 21 outputs a reception signal of the target signal Doppler bin.

Thereafter, as described above (FIG. 30), the evaluation and correction processing by the Doppler correction means 42 and the time delay estimation value evaluation means 43 are repeated, and the super-resolution time delay estimation processing unit 10 performs the time delay of the final target signal. An estimated value fd ^ is determined.
As described above, according to the twentieth embodiment of the present invention, since the S / N is improved by the pulse compression in addition to the function and effect of the eighteenth embodiment described above, the super-resolution time delay estimation process is performed. The detection performance of the target detection process can be improved.

Embodiment 21. FIG.
In Embodiment 20 (see FIG. 32), Doppler correction means 42 and time delay estimated value evaluation means 43 are applied to the configuration of Embodiment 3 (FIG. 10), but as shown in FIG. The Doppler correction unit 42 and the time delay estimated value evaluation unit 43 may be applied to the configuration of the above-described fourth embodiment (see FIG. 13).

  33 is a block diagram showing a radar apparatus according to Embodiment 21 of the present invention. The same transmitter 1, transmitting antenna 2, target 3, receiving antenna 4, receiver 5, A as those described above (FIG. 13). The / D converter 6, the pulse direction FFT 7 (# 1) to 7 (#Nr), the time gate type super resolution time delay estimation processing unit 14, the memory circuit 21, and the pulse compression type signal detection processing means 27 are the same as described above. A detailed description is omitted with reference numerals. The Doppler correction means 42 and the time delay estimated value evaluation means 43 are the same as those described above (FIGS. 30 to 32).

Next, the operation according to the twenty-first embodiment of the present invention shown in FIG. 33 will be described.
First, as described above, when a pulse is transmitted from the transmitter 1 and the transmission antenna 2, the radio wave reflected by the target 3 is received by the reception antenna 4 and the receiver 5, and the pulse compression signal detection processing means 27 is received. Output a Doppler frequency observation value and a time delay bin of the target signal, and output a received signal of the target signal Doppler bin from the memory circuit 21.

Thereafter, as described above (FIG. 30), the evaluation and correction processing by the Doppler correction means 42 and the time delay estimation value evaluation means 43 is repeated, and the final target signal is obtained by the time gate type super-resolution time delay estimation processing unit 14. The estimated time delay value fd ^ is obtained.
As described above, according to the twenty-first embodiment of the present invention, in addition to the operational effects of the eighteenth embodiment described above, the time delay estimation range is limited to the time gate range, so that ambiguity is taken into consideration. The processing load of super-resolution time delay estimation for Doppler correction can be reduced.
Further, since the S / N is improved by the pulse compression, it is possible to improve the detection performance of the target detection process performed in the super-resolution time delay estimation process.

Embodiment 22. FIG.
In the second embodiment (see FIG. 5), the Doppler correction means 9 that simply outputs only the Doppler correction signal is used. However, as shown in FIGS. 34 and 35, the Doppler based on the pulse correlation signal and the pulse compression is used. You may use the Doppler correction means 44 which outputs a correction signal and a time delay bin.

  FIG. 34 is a block diagram showing a radar apparatus according to Embodiment 22 of the present invention. The same transmitter 1, transmitting antenna 2, target 3, receiving antenna 4, receiver 5, A as those described above (FIG. 5). The / D converter 6, the pulse direction FFT 7 (# 1) to 7 (#Nr), the signal detection processing unit 13, and the time gate type super-resolution time delay estimation processing unit 14 are described in detail with the same reference numerals. Is omitted.

FIG. 35 is a block diagram specifically showing the Doppler correction means 44 in FIG.
In FIG. 35, Doppler correction means 44 includes pulse correlation processing means 45, Doppler ambiguity limiting means 46, ambiguity limited Doppler correction means 47, pulse compression means 23, and Doppler correction accuracy evaluation means 48. It is comprised by.

  The pulse correlation processing unit 45 in the Doppler correction unit 44 detects the received pulse with respect to the Doppler bin in which the target signal exists, using the time delay bin and the received signal from the signal detection processing unit 13 as input information, and transmits the received pulse. A pulse correlation signal is generated by complex multiplication of the pulse.

  The Doppler ambiguity limiting means 46 performs FFT processing on the pulse correlation signal using the Doppler frequency observation value from the signal detection processing means 13 and the pulse correlation signal from the pulse correlation processing means 45 as input information, and performs Doppler. Estimate the frequency and limit the Doppler ambiguity range.

The ambiguity limiting type Doppler correction unit 47 receives the received signal from the signal detection processing unit 13 and the Doppler ambiguity range limited by the Doppler ambiguity limiting unit 46 as input information. Ambiguity correction is performed, and the received signal after Doppler ambiguity correction is output to the pulse compression means 23 as a correction signal.
As described above (see FIG. 12), the pulse compression unit 23 includes the FFT processing unit 17 and the IFFT processing unit 25, and outputs the correction signal subjected to the pulse compression processing to the Doppler correction accuracy evaluation unit 48.

  The Doppler correction accuracy evaluation unit 48 includes an ambiguity estimation unit, and based on the correction signal subjected to the pulse compression process input from the pulse compression unit 23, the ambiguity correction instruction signal is converted into an ambiguity limited type Doppler. The correction signal is input to the correction unit 47, and the Doppler correction signal by the ambiguity when the peak value is the highest among the correction signals from the pulse compression unit 23 is output together with the time delay bin.

  The ambiguity estimation unit in the Doppler correction accuracy evaluation means 48 estimates the ambiguity using the ambiguity as an unknown parameter and the peak value after the pulse compression processing by the pulse compression means 23 as the evaluation function value for the unknown parameter. To do.

  The ambiguity estimation unit in the Doppler correction accuracy evaluation unit 48 includes an ambiguity search unit. The ambiguity search unit includes a transmission pulse transmitted from the transmitter 1 and the transmission antenna 2, and a reception antenna 4. Further, the range of the Doppler frequency is specified from the phase change of the correlation signal with the received pulse received by the receiver 5, and the ambiguity is searched by limiting to the range of the Doppler frequency.

Next, the operation according to the twenty-second embodiment of the present invention shown in FIGS. 34 and 35 will be described with reference to FIG.
FIG. 36 is an explanatory diagram showing the processing of the Doppler correction means 44 including the pulse correlation processing means 45, and a plurality of target signals based on the received signals from the signal detection processing means 13 within the resolution range of the Doppler frequency of the pulse correlation signals. Component candidates are shown for each Doppler frequency field of view Bd1.

  In FIG. 36, the horizontal axis represents the Doppler frequency, the vertical axis represents the input signal intensity, and the Doppler frequency field of view Bd1 of each region # 1 to # (Nr / Ns) corresponds to each pulse direction FFT within the resolution range of the Doppler frequency. doing. Of the target signal component candidates (integrated values) for each Doppler frequency field of view Bd1, only one candidate that has a peak value (the phase of the Doppler frequency coincides with the actual movement of the target 3) corresponds to the true target signal.

  In other words, within the resolution range of the Doppler frequency of the pulse correlation signal, Nr / Ns target signal component candidates fd1- [Nr / 2Ns] Bd1, fd1-{[Nr / 2Ns] -1} Bd1,..., Fd1 + Among [Nr / 2Ns] Bd1, one component candidate indicating the peak value corresponds to the Doppler frequency of the target signal.

34, first, when a radio wave subjected to pulse modulation is transmitted from the transmitter 1 and the transmission antenna 2, and the radio wave reflected by the target 3 is received by the reception antenna 4 and the receiver 5, signal detection processing means 13, a Doppler frequency observations fd2 target signal, the received signal of the target signal Doppler bins z0 nf (1), ···, z0 nf and (Nr), and a time delay bin n .tau.d target signal, the Doppler correction Input to means 44.

  Here, the Doppler frequency observation value fd2 output from the signal detection processing means 13 and the Doppler frequency fd1 observed at the chip width (sampling interval) T from the peak value of the pulse correlation signal in the Doppler correction means 44 are used. Although fd1 and fd2 are used for distinction, it goes without saying that they are different from fd1 and fd2 used in the above-described embodiment.

In Figure 35, time delay bin n .tau.d and received signals z0 nf (1), ···, z0 nf (Nr) is input to the pulse correlation processing unit 45 in the Doppler compensation unit 44.
Pulse correlation processing unit 45, the received signal z0 nf (1), ···, z0 nf (Nr) and based on the time delay bin n .tau.d, pulses received are reflected at the target 3 z0 nf (n τd), ..., and generates a z0 nf (n τd + Ns- 1), transmission pulse waveform z0 (1), ..., with z0 (Ns), the following equation (30), the pulse correlation signal z1 (1), ..., z1 (Ns) is generated.

However, in the equation (30), z0 * nf ( i) denotes the complex conjugate of z0 nf (i).
The pulse correlation signals z1 (1),..., Z1 (Ns) are subjected to FFT, and the Doppler frequency fd1 when observed with the chip length is obtained from the peak value of the FFT processing result. Here, when the peak is present in the i z1 (1 ≦ i z1 ≦ Ns) bin, the Doppler frequency fd1 when observed with the chip length is expressed by the following equation (31).

  The resolution of the Doppler frequency of the pulse correlation signal is expressed by the following equation (32).

  Further, the final Doppler frequency fd (1) corresponding to the actual movement of the target 3 is expressed by the following equation (33), and the lower limit value and the upper limit value based on the resolution obtained by the equation (32). Exists in the range between.

Here, the integer (minimum value) n min when the multiplication value n · Bd1 of the integer n and the Doppler frequency field of view Bd1 becomes larger than the lower limit value for the first time, and the multiplication value n · Bd1 becomes larger than the upper limit value for the first time. A candidate Dodler frequency fd (n amb ) is set by the following equation (34) using the integer (maximum value) n max of the hour .

The Doppler frequency candidate fd (n amb ) is input to the ambiguity limited Doppler correction means 47. First, the ambiguity limiting type Doppler correction means 47 sets the ambiguity number n amb = 0 and performs Doppler correction of the input signal by the following equation (35).

  The output signal of the ambiguity limited Doppler correction means 47 is input to the pulse compression means 23, and the pulse compression means 23 inputs the signal subjected to the pulse compression processing to the Doppler correction accuracy evaluation means 48. The Doppler correction accuracy evaluation means 48 checks the value of the peak that is the target signal component of the pulse compression signal.

When the evaluation result of the ambiguity number n amb is n amb <Ns / Nd, the Doppler correction accuracy evaluation means 48 increments the ambiguity number n amb “n amb ← n amb +1” and An instruction signal for performing Doppler correction by the Guyty restriction type Doppler correction means 47 is input.

Further, when the evaluation result of the ambiguity number n amb is n amb = Ns / Nd, the Doppler correction accuracy evaluation means 48 checks the ambiguity number n amb 0 when the peak value is the largest, The correction signal at that time is output.
Thereafter, in the same manner as described above, the time gate type super-resolution time delay estimation processing unit 14 obtains the time delay estimated value of the target signal.

  As described above, according to the twenty-second embodiment (see FIGS. 34 and 35) of the present invention, the Doppler estimation processing unit includes the Doppler correction unit 44, and the Doppler correction unit 44 includes the FFT processing unit 17 and the IFFT processing. Pulse compression means 23 including means 25 (see FIG. 12) and Doppler correction accuracy evaluation means 48 including an ambiguity estimation unit are included.

  The ambiguity estimation unit in the Doppler correction accuracy evaluation means 48 uses the ambiguity as an unknown parameter, and uses the peak value after the pulse compression processing by the pulse compression means 23 as an evaluation function value for the unknown parameter. Is estimated.

  The ambiguity estimation unit in the Doppler correction accuracy evaluation unit 48 includes an ambiguity search unit. The ambiguity search unit includes a transmission pulse transmitted from the transmitter 1 and the transmission antenna 2, and a reception antenna 4. Further, the range of the Doppler frequency is specified from the phase change of the correlation signal with the received pulse received by the receiver 5, and the ambiguity is searched by limiting to the range of the Doppler frequency.

  Thereby, the search range of the ambiguity of the Doppler frequency can be limited to the resolution range of the Doppler frequency in the pulse correlation signal, and the load of the ambiguity search can be reduced.

Embodiment 23. FIG.
In the twenty-second embodiment (see FIG. 34), the Doppler correction means 44 is applied to the configuration of the above-described second embodiment (see FIG. 5). However, as shown in FIG. You may apply the Doppler correction means 44 to the structure of (refer FIG. 13).

  FIG. 37 is a block diagram showing a radar apparatus according to Embodiment 23 of the present invention. The same transmitter 1, transmission antenna 2, target 3, reception antenna 4, receiver 5, A as those described above (FIG. 13). The / D converter 6, the pulse direction FFT 7 (# 1) to 7 (#Nr), the time gate type super resolution time delay estimation processing unit 14, the memory circuit 21, and the pulse compression type signal detection processing means 27 are the same as described above. A detailed description is omitted with reference numerals. The Doppler correction means 44 is the same as that shown in FIGS.

Next, the operation according to the twenty-third embodiment of the present invention shown in FIG. 37 will be described.
As described above, when a pulse is transmitted from the transmitter 1 and the transmission antenna 2 and the radio wave reflected by the target 3 is received by the reception antenna 4 and the receiver 5, the pulse compression type signal detection processing unit 27 outputs the Doppler signal. The frequency observation value and the time delay bin of the target signal are output and input to the Doppler correction means 44.
Further, the received signal of the target signal Doppler bin is input from the memory circuit 21 to the Doppler correction unit 44. Thereafter, in the same manner as described above, the time gate type super-resolution time delay estimation processing unit 14 obtains the time delay estimated value of the target signal.

  As described above, according to the twenty-third embodiment of the present invention, in addition to the effect of the twenty-second embodiment described above, the S / N is improved by the pulse compression in the pulse compression type signal detection processing means 27. The detection performance of the target detection process performed during the super-resolution time delay estimation process can be further improved.

In the case of a radar apparatus using the super resolution time delay estimation processing unit 10 instead of the time gate type super resolution time delay estimation processing unit 14 in FIG. It is also possible to apply pulse compression processing as the time delay estimation processing unit.
In addition, a time gate that limits the time delay estimation range is provided, and pulse compression processing limited to the time gate and super-resolution time delay estimation processing are performed while sliding the time gate, and target signals in the entire delay time range are detected. It is also possible to adopt a method to do this.

It is a block block diagram which shows the radar apparatus which concerns on Embodiment 1 of this invention. It is explanatory drawing which shows the time relationship of each signal in the transmission / reception of multiple times by Embodiment 1 of this invention. It is a block diagram which shows the specific structure of the signal detection process means based on Embodiment 1 of this invention. It is explanatory drawing which shows the relationship between the target signal by Embodiment 1 of this invention, a time delay bin, and a Doppler bin. It is a block block diagram which shows the radar apparatus which concerns on Embodiment 2 of this invention. It is a block diagram which shows the internal structure of the signal detection process means based on Embodiment 2 of this invention. It is a block diagram which shows the internal structure of the time gate type | mold super-resolution time delay estimation process part which concerns on Embodiment 2 of this invention. It is explanatory drawing which shows the time relationship of the time gate in Embodiment 2 of this invention. It is a block diagram which shows the internal structure of the decimation processing means which concerns on Embodiment 2 of this invention. It is a block block diagram which shows the radar apparatus which concerns on Embodiment 3 of this invention. It is a block diagram which shows concretely the pulse compression type signal detection processing means concerning Embodiment 3 of this invention. It is a block diagram which shows concretely the pulse compression means which concerns on Embodiment 3 of this invention. It is a block block diagram which shows the radar apparatus which concerns on Embodiment 4 of this invention. It is a block diagram which shows concretely the pulse compression type signal detection processing means concerning Embodiment 4 of this invention. It is a block block diagram which shows the radar apparatus concerning Embodiment 5 of this invention. It is a block block diagram which shows the radar apparatus which concerns on Embodiment 6 of this invention. It is a block diagram which shows concretely the signal detection processing means concerning Embodiment 6 of this invention. It is a block block diagram which shows the radar apparatus which concerns on Embodiment 7 of this invention. It is a block diagram which shows concretely the signal detection processing means concerning Embodiment 7 of this invention. It is a block block diagram which shows the radar apparatus based on Embodiment 8 of this invention. It is a block block diagram which shows the radar apparatus which concerns on Embodiment 9 of this invention. It is a block block diagram which shows the radar apparatus based on Embodiment 10 of this invention. It is a block block diagram which shows the radar apparatus based on Embodiment 11 of this invention. It is a block block diagram which shows the radar apparatus based on Embodiment 12 of this invention. It is a block block diagram which shows the radar apparatus based on Embodiment 13 of this invention. It is a block block diagram which shows the radar apparatus based on Embodiment 14 of this invention. It is a block block diagram which shows the radar apparatus based on Embodiment 15 of this invention. It is a block block diagram which shows the radar apparatus based on Embodiment 16 of this invention. It is a block block diagram which shows the radar apparatus based on Embodiment 17 of this invention. It is a block configuration diagram showing a radar apparatus according to Embodiment 18 of the present invention. It is a block configuration diagram showing a radar apparatus according to Embodiment 19 of the present invention. It is a block block diagram which shows the radar apparatus based on Embodiment 20 of this invention. It is a block block diagram which shows the radar apparatus based on Embodiment 21 of this invention. It is a block block diagram which shows the radar apparatus based on Embodiment 22 of this invention. It is a block diagram which shows concretely the Doppler correction means based on Embodiment 22 of this invention. It is explanatory drawing which shows the process of the Doppler correction means based on Embodiment 22 of this invention. It is a block block diagram which shows the radar apparatus based on Embodiment 23 of this invention. It is a block block diagram which shows the conventional radar apparatus. FIG. 39 is a block diagram specifically showing the internal configuration of the super-resolution processing means in FIG. 38. It is explanatory drawing which shows the relationship between the transmission pulse train and reception pulse train in the conventional radar apparatus, and the delay time between both.

Explanation of symbols

  1 transmitter, 2 transmit antenna, 3 target, 4 receive antenna, 5 receiver, 6 A / D converter, 7 (# 1) to 7 (#Nr) pulse direction FFT, 8, 13, 32, 34 signal detection Processing means 9, 9 (# 1) to 9 (#Nf), 42, 44 Doppler correction means, 10 super-resolution time delay estimation processing section, 11 time direction output means, 12, 15, 26, 28, 33, 35 Target Doppler bin detection processing means, 14-hour gate type super-resolution time delay estimation processing section, 16-time gate processing means, 17 FFT processing means, 18 division signal generation means, 19 decimation processing means, 20 super-resolution processing means, 21 memory circuit 22, 27 Pulse compression type signal detection processing means, 23 Pulse compression means, 24 Correlation processing means, 25 IFFT processing means, 29 Time direction output means, 0 H / L-PRF generator, 31 Doppler frequency estimation means, 36 Multi-PRF switching means, 37, 38, 40, 41 Doppler ambiguity correction means, 39 Transmission frequency switching means, 43 Time delay estimated value evaluation means, 45 Pulse correlation processing means, 46 Doppler ambiguity limiting means, 47 ambiguity limited Doppler correction means, 48 Doppler correction accuracy evaluation means.

Claims (10)

  1. A transmitter that generates radio waves,
    A transmitting antenna that transmits radio waves,
    A receiving antenna that receives a radio wave transmitted from the transmitting antenna and reflected by a target as a received wave;
    A receiver that performs band limitation and phase detection of the received wave to generate a target signal corresponding to the target;
    Estimating a Doppler frequency of the target signal due to a Doppler effect resulting from the movement of the target, and obtaining a Doppler frequency estimate, and
    A radar apparatus comprising: a time delay estimation processing unit configured to correct a phase rotation of the target signal due to the Doppler effect and estimate a time delay of the target signal.
  2. The Doppler estimation processing unit
    An A / D converter that converts a received signal obtained via the receiver into a digital signal;
    A pulse direction FFT for performing FFT processing on a received signal converted into a digital signal via the A / D converter in a pulse hit direction;
    And a signal detection processing means for detecting the Doppler bins presence of the target signal based on the output signal of the pulse direction FFT,
    The radar apparatus according to claim 1, wherein the time delay estimation processing unit includes a super resolution time delay estimation processing unit that performs super resolution estimation of a time delay of the target signal.
  3. The Doppler estimation processing unit
    An A / D converter that converts a received signal obtained via the receiver into a digital signal;
    A pulse direction FFT for performing FFT processing on a received signal converted into a digital signal via the A / D converter in a pulse hit direction;
    And a signal detection processing means for detecting the Doppler bins presence of the target signal based on the output signal of the pulse direction FFT,
    2. The time delay estimation processing unit includes a time gate type super resolution time delay estimation processing unit that limits a time delay estimation range and estimates a time delay of the target signal in a super resolution. Radar device.
  4. The Doppler estimation processing unit
    An A / D converter that converts a received signal obtained via the receiver into a digital signal;
    A pulse direction FFT for performing FFT processing on a received signal converted into a digital signal via the A / D converter in a pulse hit direction;
    Using the target signal subjected to the pulse compression processing based on the output signal of the pulse direction FFT, and a pulse compression-type signal detection processing means for detecting the Doppler bins presence of the target signal,
    The radar apparatus according to claim 1, wherein the time delay estimation processing unit includes a super resolution time delay estimation processing unit that performs super resolution estimation of a time delay of the target signal.
  5. An H / L-PRF generator for inputting a transmission switching instruction to the transmitter and switching and transmitting a transmission pulse of HPRF or LPRF from the transmitter,
    The radar apparatus according to claim 1, wherein the Doppler estimation processing unit includes Doppler frequency estimation means for estimating a Doppler frequency of the target signal by transmission / reception based on a transmission pulse of the HPRF.
  6. When a transmission switching instruction is input to the transmitter, a transmission pulse having a first pulse repetition period PRF1 and a second pulse having a prime relationship with respect to the first pulse repetition period PRF1 are transmitted from the transmitter. A multi-PRF switching means for switching and transmitting the transmission pulse of the repetition period PRF2;
    The Doppler estimation processing unit includes Doppler ambiguity correction means,
    The Doppler ambiguity correcting means is
    A first Doppler frequency observation value of a target signal obtained when transmitting and receiving with a transmission pulse of the first pulse repetition period PRF1,
    From the second Doppler frequency observation value of the target signal obtained when transmitting and receiving with the transmission pulse of the second pulse repetition period PRF2,
    The radar apparatus according to claim 1, wherein the Doppler frequency estimation value is obtained by solving the ambiguity of the Doppler frequency.
  7. When a transmission switching instruction is input to the transmitter, a transmission pulse of the first wavelength λ1 from the transmitter and a transmission pulse of the second wavelength λ2 that are relatively prime to the first wavelength λ1 Including transmission frequency switching means for switching and transmitting,
    The Doppler estimation processing unit includes Doppler ambiguity correction means,
    The Doppler ambiguity correcting means is
    A first Doppler frequency observation value of a target signal obtained when transmitting and receiving with a transmission pulse of the first wavelength λ1,
    From the second Doppler frequency observation value of the target signal obtained when transmitting and receiving with the transmission pulse of the second wavelength λ2,
    The radar apparatus according to claim 1, wherein the Doppler frequency estimation value is obtained by solving the ambiguity of the Doppler frequency.
  8. A time delay estimation value evaluation unit including an ambiguity estimation unit for evaluating the accuracy of the time delay estimation value input from the time delay estimation processing unit;
    The ambiguity estimation unit is
    Let the ambiguity of the Doppler frequency be an unknown parameter,
    The peak value of the MUSIC spectrum calculated in the process of the time delay estimation processing unit as the evaluation function value for the unknown parameter,
    The radar apparatus according to claim 1, wherein the ambiguity is estimated.
  9. The Doppler estimation processing unit includes Doppler correction means,
    The Doppler correction unit includes a pulse compression unit including an FFT processing unit and an IFFT processing unit, and a Doppler correction accuracy evaluation unit including an ambiguity estimation unit,
    The ambiguity estimation unit is
    Let ambiguity be an unknown parameter,
    The peak value after the pulse compression processing by the pulse compression means, as an evaluation function value for the unknown parameter,
    The radar apparatus according to claim 1, wherein the ambiguity is estimated.
  10. The ambiguity estimation unit includes ambiguity search means,
    The ambiguity search means is:
    From the phase change of the correlation signal between the transmission pulse transmitted from the transmitter and the transmission antenna and the reception pulse received by the reception antenna and the receiver,
    10. The radar apparatus according to claim 8, wherein a range of the Doppler frequency is specified, and ambiguity is searched by limiting to the range of the Doppler frequency. 11.
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