JPS63282666A - Current-voltage transducer - Google Patents

Abstract

(57) [Summary] This bulletin contains application data before electronic filing, so abstract data is not recorded.

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to an electromagnetic contactor, and more particularly to an apparatus for sensing current in a contactor.

Electromagnetic contactors are well known, and one example is U.S. Patent '33,3.
No. 39.161. Magnetic contactors are particularly useful devices for applications such as motor starting, lighting, and switching. A motor starting contactor with an overload relay system is called a motor controller. Contactors and motor controllers sense the current flowing through the closed contacts and utilize information regarding this current for a variety of useful purposes. Current is often sensed with a known current transformer that outputs a current approximately proportional to its input terminal. Traditional current transformer systems have many drawbacks. Part 1
For one thing, conventional current transformers can only be used for a relatively narrow range of input power. To measure input currents outside this range, the current meter must be replaced with a larger or smaller one. Another disadvantage is that current transformers tend to be relatively large and relatively heavy. This is extremely disadvantageous if it is desired to downsize the motor controller or contactor. Yet another drawback is that conventional current transformers are configured to operate in the saturation range of the magnetization curve. This means that the output voltage has a nonlinear relationship over a wide range of input currents. It would be beneficial if a device could be developed that outputs a linear output voltage proportional to the input current so that the voltage can be used directly in the voltage sensing control circuit, which typically uses a current transformer to obtain an output current proportional to the input terminal. be. An air-core current transformer, sometimes called a linear coupler, is one type of transducer that outputs a voltage proportional to the derivative of an input current. However, linear couplers are only used for relatively low input powers. A relatively large input t over a wide range? It is desired to develop this type of device that can be used for a.

Therefore, it would be desirable to implement a motor controller or contactor that uses current transformers that are relatively compact yet available for a wide range of input currents. However, conventionally, it has been difficult to develop a current transformer that satisfies the above two conditions. For example, U.S. Patent 4,
No. 454,557 discloses a current transformer that can be used with a relatively wide range of input currents, but achieves its purpose by providing an intentionally nonlinear output. Current transformers are also utilized in various saturation stages associated with relatively heavy current transformers with high losses. Another current transformer of interest as a background to the present invention is that disclosed in U.S. Pat. No. 3,197,721.

The invention comprises a magnetic core and a primary winding that electromagnetically interacts with the magnetic core to generate a magnetic flux in the magnetic core that is approximately proportional to the amount of current I flowing through the primary winding. A voltage transducer, wherein the magnetic core has a discontinuously variable air gap, and the discontinuously variable air gap is such that the magnetic core The magnetic core has a first magnetic resistance that prevents saturation, and the magnetic core is provided with a secondary winding that outputs a voltage from an output terminal that is approximately proportional to the first derivative of the magnetic flux in the non-saturated magnetic core. We propose a current-voltage transducer characterized by:

Embodiments of the present invention will be described below along with the accompanying drawings.

1 and 2 illustrate a three-phase contactor or controller 10. FIG. For convenience, the configuration of only one of the three poles will be explained, but the other two poles are also exactly the same. The contactor 10 has a housing 1 made of a suitable electrically insulating material such as a glass/nylon composition.
2 and controlled by the contactor 10;
Electrical load terminal 1 for connection with a circuit or system
4.16 are arranged in the housing 12. An example of such a system is schematically illustrated in FIG. Terminal 14.16
may each be configured to form a part of the three-phase terminal. Terminals 14 and 16 are spaced apart from each other and are connected to the housing 1.
It connects internally with a conductor 20.24 extending into the center of 2.

Inside the housing, the ends of the conductors 2°, 24 form suitably fixed contacts 22, 26, respectively. Contact 22.2
8 are connected to each other, the terminals 14 and 16 are closed, and the contactor 10 becomes conductive. A separately manufactured coil control board 28 (as shown in Figures 8.9 and 10) is secured within the housing 12 in a manner to be described below. A coil or solenoid assembly 30 including a coil or solenoid 31 as a part is attached to this coil control panel 28 . A spring seat 32 is provided at a distance from the coil control panel 28 and forms one end of a coil assembly 30, and one end of a kickout spring 34 is fixed to this spring seat 32. kickout spring 3
At the other end of 4, the support member 42 moves as will be described later, and its lower part 42A picks up the spring 34 and holds it against the seat 32.
until the portion 12. of the housing 12 is pressed into contact with the housing 12. It is in contact with A. The pressure contact occurs in a plane outside the plane of FIG. The spring 34 surrounds the armature 40 and the support member lower part 4
2A and is picked up by the lower portion 42A. The dimension of member 42 in front of the plane of FIG. 2 is larger than the diameter of spring 34. A fixed magnet or magnetic material slug 36 is placed in a suitable manner within a passageway 38 in radial alignment with the solenoid or coil 31 of the coil assembly 30 and axially offset from the fixed magnet 36 in the same passageway 38. , a magnetic armature or magnetic flux conducting member 40 is provided which is movable in the longitudinal direction (axial direction) in the passage 38 with respect to the fixed magnet 36. An electrically insulating contact support member 42 that projects in the longitudinal direction is provided at the end of the armature 40 opposite to the fixed magnet 36, and an electrically conductive contact bridge 44 is attached to this. A contact 46 is mounted on one radial arm of the contact bridge 44, and a contact 48 is mounted on the other radial arm. It goes without saying that all three pairs of these contacts have the same configuration in a three-pole contactor. When the internal circuit is completed between the terminals 14 and 16 as the contactor 10 is closed, the contact 46 comes into contact with the contact 22 (22-46), and the contact 48 comes into contact with the contact 26 (26-46). 48). Conversely, when contact 22 separates from contact 46 and contact 26 separates from contact 48, the internal circuit between terminals 14, 18 opens. Such an open circuit state is shown in FIG. The provision of an arc box 50 surrounding the contact bridge 44 and the terminals 22, 26, 46.48 creates a partially enclosed space inside the housing in which the current flowing between the terminals 14.16 can be safely interrupted. ing. A recess 52 is provided in the center of the arc box 50, and the cross spar 54 of the contact support member 42 is inserted into this recess and fixed so as not to move in the lateral direction (radial direction) as shown in FIG. S! in the longitudinal direction (axial direction) of the center line 38A of the passage 38! I] or to be able to slide. Contact bridge 44 is held to support member 42 by contact springs 56 . Contact 22-46.26
Contact spring 56 compresses so that contact support member 42 continues to move toward slug 36 even after -48 is in abutment or "closed" condition. Further compression of the contact spring 56 significantly increases the pressure on the closing contacts 22-46 and 26-48, increasing the current carrying capacity of the internal circuit between the terminals 14.16 and ensuring that the contacts remain in contact even after significant contact wear. Provides an automatic adjustment feature that allows the user to reach the closed or “closed” position. The longitudinal region between magnet 36 and movable armature 40 defines an air gap 58 in which magnetic flux is generated when coil 31 is energized.

The externally accessible terminals in the terminal block J1 are arranged on the coil control board 28 in such a way that they can be connected to the coil or solenoid 31 via a printed circuit bus or other conductor on the coil control board 28, in particular. Another terminal block JX (shown in FIG. 32) may also be provided on the printed circuit board 28 with another purpose. When the coil or solenoid 31 is energized via the externally accessible terminal in the terminal block J1, the stationary magnet or slug 36 is activated, for example in response to the generation of a contact closure signal in said externally accessible terminal block J1. , the air gap 58 and the armature 4o. As is well known, in this condition the armature 40 moves longitudinally within the passageway 38 to shorten the air gap 58 and eventually abut the magnet or slug 36. This movement is initially resisted by the compression force of the kickout spring 34, and at the later stage of the movement of the armature 40, the contacts 22-46, 26-4
After contact 8 comes into contact, new resistance is applied due to the compressive force of the contact spring 56.

Also provided within the housing 12 of the contactor 10 (also shown in Figures 8, 9 and 10) is an overload relay printed circuit board or card 60 (of which only one 62B is shown in Figure 2). )
Embodiments in which a current-voltage transducer 62 is provided are also possible. In embodiments of the invention that utilize overload relay board 60, conductor 24 is connected to annular opening 62T of current-to-voltage transducer 62B such that current flowing through conductor 24 can be sensed by current-to-voltage transducer 62B. It should be configured so that it passes through. By using the detected information in a manner described later, circuit information necessary for the contactor 1o can be obtained.

It is also possible to provide a selector switch 64 accessible from the outside of the housing 12 at one end of the overload m electrical panel 60. Another embodiment of the present invention is shown in FIGS. 30 and 31, and its structure and operation will be described later.

FIG. 3 depicts four intersecting curves to illustrate the current technology: a magnetic solenoid, such as that shown at 31 in FIG. 2; a kick-out spring, such as that shown at 34; The relationship between force and distance is shown for a contact spring as shown at 56, and the relationship between instantaneous velocity and distance is shown for an armature as shown at 40 (curve 92).
). In both curves, the independent variable is distance, but
Time, which is closely related to distance in the curve of FIG. 3, can also be an independent variable. For convenience of explanation, the components of the contactor 10 shown in FIG. 2 will be described as an example, but this does not mean that the components shown in FIG. 2 as a whole are included in the known technology. A first curve 70 shows the distance (or time) versus force relationship for a kickout spring (eg 34) as it begins to be compressed from point 72. 74 is spring 3
4 and the spring 34 resists compression with increasing force until a point 78 on the distance axis is reached.

Point 722 Point 749 Curve 701 Point 76, Point 78 and Point 72
The area bounded by the line connecting the armature 40 is the total amount required to compress the kickout spring and close the air gap 58 between the armature 40 and the fixed magnet 36 by movement of the armature 40 as the armature 40 is accelerated. Represents the amount of energy. This force resists movement of armature 40. At point 80 on the distance axis, for example, the second
When the contacts 22-42, 26-48 of the figures are brought into abutment and the armature 40 continues to move, the contact spring 56 is compressed, exerting an even greater force on the already abutting contacts for the reasons discussed above. Let it work.

Curve 79 represents the total amount of force acting on movable armature 40 that is accelerated in the direction of closing air gap 58.

When the contacts 22-42, 26-48 make contact, the force increases in a step function between points 81 and 82. This force gradually increases until the combination of kickout spring 34 and contact spring 56 exerts a maximum force on the moving armature 40 at point 78. contact spring 56
The amount of supplementary energy that the moving armature must supply to overcome the resistance of point 81.82,
It is represented by an area surrounded by a line connecting curve 791 point 84.76, curve 76A, and point 81.

Therefore, in the process of accelerating the armature 40 from the non-operating position 72 to the abutting position 78 with the magnet 36, a small amount of
At least the coil or solenoid 31 has points 72, 74,
81.82, 84.78 and 72 must be supplied with the amount of energy represented by the lines. The positive slope of curve 70 indicates that armature 40 will decrease when coil energy is removed.
must be made as small as possible so that the contactor is driven in the opposite direction and the contactor is opened again. The initial force that armature 4o has to overcome in the first phase of its movement is the force threshold represented by the difference between points 72, 74.

The armature must therefore provide at least a force corresponding to this force at this point. Therefore,
For purposes of explanation, assume that electromagnetic coil 31 provides the force 88 (FIG. 3) required by armature 40 at point 72. Also, the force provided by the coil or solenoid 31 at point 8 in FIG.
It must be greater than the force expressed by the distance between 0.82 and 0.82, otherwise the accelerating armature air 40 will stall midway and the contact between the contacts 22-46 and 28-48 will be extremely weak. This is a condition in which the contacts are susceptible to welding and shunting, which is undesirable; therefore, the force supplied by coil 31 in accelerating armature 40 is reduced at point 80 to point 82.
The force must be greater than the force shown in The magnetic attraction curve for a solenoid and its associated movable armature will vary depending on a variety of factors such as armature weight, magnetic field strength, air gap size, etc. Such a curve will have a relatively predictable shape. It is shown at 86 in Figure 3. The relative shape of the curve 86 and the constraints up to the point 80, i.e. the coil 3 at points 72 and 80 on the distance axis in FIG.
1 determines the entire magnetic attraction curve of the armature 40 and coil 31 shown in FIG. This curve ends at a force of 90. Note that the magnetic attraction curve is characterized in that the magnetic force increases significantly as the moving armature 40 approaches the fixed magnet 36 and the air gap 58 narrows. Therefore, force 90 is exhibited at point 78. It is at this point 78 that the armature 40 first abuts or contacts the stationary magnet 36. However, as a result, two inconvenient situations occur. First, as is clear from the drawing, points 72, 88 . The total energy supplied to the magnetic system by the coil 31, represented by the line connecting curve 869 point 90.78 and point 72, is much greater than the amount of energy required to overcome the various spring resistances; this energy difference is Point 74.88. Curve 869 points 90, 84, 82° 81
and is again represented by an area surrounded by a line connecting to point 74. This energy is wasted or unnecessary energy, and it would be extremely advantageous if this energy did not have to be generated. A second disadvantageous property or event is that the acceleration of armature 40 is at its maximum just before it abuts magnet 36, generating most of its kinetic energy. As shown in FIG. 3, velocity curve 92 beginning at point 72 and ending at point 94 represents the velocity of armature 40 as it accelerates along the axial motion path.

Note the change in shape at point 80 where it engages kickout spring 34. Armature 40 is magnet 3
Immediately before contact with 6, the velocity 1 reaches its maximum value. This is extremely inconvenient because the speed at the moment when the armature 40 and the magnet 36 collide or abut each other is high, so a high kinetic energy is transmitted. This energy must be instantaneously dissipated or absorbed by other elements of the system. Typically, an instantaneous drop in energy is required to instantaneously reduce the armature speed to zero at point 78. This kinetic energy is converted into impact sound, heat, bounce, vibration, mechanical wear, etc. An armature 40 is loosely connected to contacts 46-48 on a contact bridge 44 by contact springs 56. If it bounces, the mechanical system consisting of these elements will vibrate, resulting in contact structure 2
2-42.26-48 is likely to open and close rapidly and repeatedly. This is a highly disadvantageous property in electrical circuits. Therefore, the kickout spring 34 and the contact spring 5
The energy delivered to coil 31 is carefully monitored and selected in such a way that only the exact amount of energy (or a value close to this) required to overcome the resistance of FIG. It is desirable to use the container 10. It is also desirable to significantly reduce the speed of the armature 40 when it abuts the magnet 36 to effectively reduce the possibility of "bounce." For example, the solutions to the above problems can be found in Sections 4.5 and 6.
This is achieved by the present invention as shown graphically in the figure.

In the following margin, explanations will be given in accordance with Figures 2.3 and 4. A family of curves related to the present invention, similar to the curves in FIG. 3 related to the prior art, is shown in FIG. In this case, the spring force curves 70.79 associated with kickout spring 34 and contact spring 56, respectively, are the same as in FIG. It is. In this embodiment of the invention, the magnetic attraction curve 86' representing the force provided by the coil 31 begins at a point or force level 95 to overcome the critical force of the kickout spring, and the point or force that appears at distance M96. Continues until level 97. Armature 4 by coil 31
The electrical energy supplied to 0 disappears at a distance 96 corresponding to a force level 97. That is, armature 4o
disappears before reaching the contact position with the fixed magnet 36. The maximum velocity Vm reached by armature 40 at this point is shown at point 98 on velocity curve 92'. This is the maximum speed that the armature reaches while moving into contact with the magnet 36. In other words, when the supply of electrical energy from the coil 31 is cut off, the armature stops accelerating and begins to decelerate. The deceleration curve 100 in FIG. 4 spans the range from point 98 to point 78, and the slope changes at the position where the kickout spring is engaged. This is accomplished by cutting off the flow of electrical energy to coil 31 early when distance 96 is reached. Only the amount of energy necessary to overcome the spring force is provided before the armature 40 completes its movement into abutment position with the fixed magnet 36, providing an energy efficient system. Points 96, 99 . Curve 701 points 81.82.
This is an area surrounded by a line connecting curve 792 points 84 and 78 and point 96 again. This energy is applied at point 7 of the time when electrical energy is supplied to the armature coil 31.
4,95. Curve 86゛ is surrounded by a line connecting point 97.99 and point 74 again (although not necessarily to scale)
It is supplied over a portion represented by region Z. Such an energy balance is selected by any appropriate method, such as empirical analysis of the energy level determined by experiment.

The energy represented by the area Z° is utilized to compress the kickout spring 34 during the initial movement phase of the armature, but is not utilized during the subsequent eight strokes. As discussed below, a microprocessor may be used to determine the amount of energy to be delivered. The joint M8 movement of armature 40 during the deceleration phase represented by curve 100 is equal to the kinetic energy level E reached by armature 40 at point 96 where electrical energy to coil 31 is cut off.
determined by This energy E is equal to the armature mass (M) multiplied by the square of the velocity (Vm) at point 98. In a system with perfect energy balance, the decelerating armature 40 abuts the stationary magnet 36 at zero speed at point 78 so that no bouncing occurs and excess energy in the form of noise, wear, heat, etc. is absorbed. There's no need. It goes without saying that it is difficult to realize the ideal as shown in FIG. 4, and in fact there is no need to manufacture such a highly efficient system. Therefore, FIG. 4 shows an ideal system for explaining the present invention in detail, and it is extremely difficult to bring the armature 40 into contact with the magnet 36 at point 78 at exactly zero velocity. Small residual velocities are acceptable, especially when compared to the velocity 94 in known systems as shown in FIG.

2.4 and 5, which relates to a system in which the contact spring 56 is relatively strong and therefore the forces that the armature 40 has to overcome are also large. A group of curves similar to those shown in FIG. 4 was shown. In addition to the features of the embodiment described above, other features are also presented in FIG. For example, since the coil is powered for a longer time than in the embodiments described above, the speed of the movable armature 40 can reach higher values.

Higher velocity values are required because the spring force of the contact spring 56 is greater than in the embodiment shown in FIG. 4, and to overcome this it is necessary to increase the kinetic energy. The same reference symbols in FIGS. 4 and 5 represent corresponding points on the curves in both figures. In the embodiment of the invention shown in FIG. 5, kick act spring 34 and contact spring 56
The total energy required to compress points 82, 102.
Curve 79° 1 point 104, 84. Curve 79 and point 82 again
increases by the amount U represented by the area surrounded by the curve or line connecting . The remaining area, namely points 72, 74.
Curve 70. Points 81, 82. The area surrounded by the line connecting curve 79, points 84, 78 and 72 again is the same as the corresponding area in FIG. In order to obtain a larger energy U, a magnetic attraction curve 86" different from that shown in FIG. 4 is formed. This magnetic attraction curve has a slightly larger average slope;
The connection spans the time interval represented by the distance difference between points 96 and 100, resulting in an incremental increase in energy U.

The new magnetic attraction curve 86'' is at point 9 as in Figure 4.
5 and ends at point 97', which is represented by a distance of 100. This attraction curve produces a steeper and longer velocity curve 92'' for movable armature 40 than in FIG. 4. A peak velocity v2 is reached at point 98' of velocity curve 92''. At this point, the kinetic energy (E2) of armature 40 is equal to % of MV2 squared. The instantaneous velocity then decreases, forming a curve of 100° with a clear breakpoint at velocity 1. This breakpoint represents the first contact between the armature and contact spring 56.

The increased velocity v2 and therefore the increased energy E2
Since a portion of the energy is rapidly absorbed by the above-mentioned increase in energy due to the strong, i.e., high resistance, contact spring, the curve 100'' theoretically becomes zero at the point 78 when the movable armature 40 abuts the fixed magnet 36. This will now be explained with reference to Figures 2, 4 and 6. Figure 6 shows the voltage and current curves for the coil 31 and the relationship between these curves and the force curve of Figure 4. In a preferred embodiment of the invention, the coil current and voltage are controlled in the following four steps in the manner described in connection with the embodiment of FIG. 7: (1) Armature 40;
ACCELERATION stage to accelerate (2
) Co for adjusting the armature speed at the latter stage of armature movement before contact with the fixed magnet 36
AST stage, (3) GRAB stage in which the armature 40 is brought into close contact with the fixed magnet 36 to damp vibration and bounce immediately after contact, and (4) HOLD stage to hold the armature. Please refer to Table 1 for supplementary explanation above and below. The information from Table 1 is stored in the microprocessor's memory as a menu, as described below. During the ACCELERATION phase, electrical energy is supplied to the coil or solenoid 31 at time 72° associated with point 72 on the distance axis of FIG. 4, and at time 9 associated with point 96 on the distance axis of FIG.
At 6° the supply is cut off.

The energy represented by regions 2 and Z' in FIG. 4 is obtained by appropriate selection of the voltage across the coil 31 and the current flowing through the coil.

Apparatus and methods for controlling the voltages and currents are described in detail below with respect to FIG. Appropriate waveforms are shown in FIG. 6 for convenience, and a device for providing these waveforms will be described later. In a preferred embodiment of the invention, the voltage applied across the terminals of coil 31 may be an unfiltered full wave rectified AC voltage represented by waveform 106 having peak amplitude 110.

The current flowing through coil 31 is full-wave rectified, unfiltered,
An AC current pulse 108 with conduction angle control, which current flows through the coil 31 according to Table 1. The voltage may be applied to the coil 31 as shown at 106A, 106B, 106C and 106D in FIG. In one embodiment of the invention, the total power supplied to the coil 31 over the time period from time 72' to time 96' is such that the combination of current and voltage that constitutes the time is (72''-96°). over voltage wave 106 such that the amplitude of the fully conducting current waveform is equal to the mechanical energy required to close the contacts as described above.
is obtained by adjusting in relation to the peak amplitude 110 of . However, in other embodiments of the invention, as shown in Table 1, a gated device such as a triac may be connected in series with coil 31 as described in more detail below with respect to FIG. The predetermined portion α1 of the wave current pulse 108.
The total amount of power supplied to the coil 31 over a period of time (72'-96°) with the coil generally in a non-conducting state over α2, etc., i.e. with the coil generally in a conductive state over parts β1, β2, etc. can be adjusted. Some coil current flows during the conduction interval because the energy magnetically stored during the previous conduction interval is released. In a preferred embodiment of the invention, the number of current conduction angle control pulses is determined by the length of time that the coil 31 has to supply magnetic energy in the manner already described. Embodiments of the invention may also be configured to suitably adjust the pulse 108 before the point 96° and still provide a continuous electrical energy supply to the coil 31 for accelerating the armature 40 in the manner described above. It is possible. In other embodiments of the invention, sufficient energy cannot be obtained by simply adjusting the current conduction cycle at the appropriate time, and the necessary adjustments are made as described below.

Note that, for example, the smooth curves or waves 106° and 108 are just ideal waveforms, and are not actually as shown. In the ideal situation shown in FIG. 6, at time 96' the armature 40 is accelerated to an energy level E sufficient to continue to compress kickout spring 34 and contact spring 56, after which the armature decelerates and at time 78' According to curve 100, armature 40 gently contacts magnet 36 at zero speed as shown in FIG. However, in reality, it is difficult to achieve such conditions. For example, the appropriate time (72'-96')
Within one moment, the amount of electrical energy provided by the combination of voltage waveform 106 and conduction control current waveform 108 is insufficient to provide armature 40 with the kinetic energy necessary to complete a contact closing cycle. This state is represented by a speed curve 100A in FIG. 4, for example. That is,
Armature 40 stops before contacting fixed magnet 36. That is, zero velocity is reached. In this case, the combination spring 34- of the contact spring 56 and the kick-out spring 34
The contactor 40 acts to accelerate in the opposite direction of the armature 40 until the contactor 56 relaxes, thereby preventing the contact mechanically connected to the armature 40 from closing, thereby rendering the contactor 10 incapable of closing. Although this condition is inconvenient, the condition in which the armature 40 is about to come into contact with the fixed magnet 36 is even more inconvenient. This is because there is a risk that an arc will occur between the contacts and contact welding will significantly increase. Since sufficient energy is not available to accelerate the armature within a reasonable time frame, "mid-stream" modifications are required to "fine tune" the velocity curve of armature 40 based on new information.

This modification is made in the CoAST portion of FIG.

In a preferred embodiment of the present invention, the armature 40 is connected to the fixed magnet 3 at a relatively low, if not zero, speed.
6, the armature deceleration curve is deflected from curve 100 to curve 100B in FIG.
The armature 40 is re-accelerated by supplying a regulating current pulse 116 at 18". This regulating pulse 116 sets the triac firing control angle α3, for example, much larger than the angles α1 and α2. A preferred embodiment of the invention In this case, it is assumed that the angle α1=α2, but this condition is not necessarily restricted, and is selected depending on the control system used for the current conduction bus to the coil 31. 36, the contactor 10 is in the "closed" state.Since factors such as vibration can cause extremely undesirable bouncing, the control circuit for the current in the coil 31 is constructed in a known manner as described below. Operation generates a number of "5eal in" or GRAB pulses that act on the abutting armature 40 and fixed magnet 36. At least in theory, the armature 4
Since the advance of 0 has already been stopped by contact with the magnet 36, or is about to stop, the introduction of the contact pulse will not cause acceleration of the armature, i.e. the bus of the armature is fixed magnet 36. 36
This is because it is physically blocked by the presence of Rather than causing acceleration, all vibrations are damped,
The contacts are firmly attached. In a preferred embodiment of the invention,
For example, a close contact or GRAB pulse 120 is generated by passing a coil current over a portion of a current half-wave represented by conduction angles β4, β5, and β6, and a close contact or GRAB pulse 120 is generated.
B-stage control is performed. ACCELERAT
ION.

The COAST and GRAB control operations are performed based on the principle of feedforward voltage control. Final control stage HOL
At D, the mechanical system is almost stationary, but some magnetism is required to keep the armature 40 in contact with the stationary magnet 36 and keep the contacts closed. There, to prevent the kickout spring 34 from accelerating the armature 40 in the opposite direction and opening the contact, the relatively small current flow is applied once every half-cycle for the period of time that the contact must remain closed. , repeating the variable hold pulse 124. The amount of electrical energy required to hold armature 40 in contact with magnet 36 is sufficient to accelerate armature 40 toward magnet 36 to overcome the forces of kickout spring 34 and contact spring 56 during the closing operation. much smaller than needed. Pulse 124 is obtained by significantly increasing the phase back, retardation or firing angle, for example C7. The angle α7 can be changed by the current pulse. That is, the next phase delay angle α8 may be larger or smaller than the angle α7. This is achieved by closed loop current control. That is, the current flowing through coil 31 is sensed and readjusted as necessary, as described below in connection with FIG.

FIGS. 7A to 7D are block diagrams of control circuits of the present invention. The coil control card 28 of FIGS. 2.8.9 and 10 is provided with a terminal board or strip J1 for connection with external control elements, such as the one shown in FIG. 11, for example. The terminal board J1 has terminals 1 to 5, respectively labeled with reference symbols, and terminal "2" has a resistive element R1.
One end of the resistance element R2, and the first AC input terminal of the full-wave bridge rectifier BRI are connected. The other end of resistance element R1 is connected to one end of capacitive element C1 and one end of resistance element R16. Connect one end of resistance element R16 to 120 V.
The other end of resistive element R2 is indicated by "AC" of the bipolar linear custom analog integrated circuit module U1.
LINE" input terminal, the function of which will be described later. The "LINE" input terminal is also connected to the 840 terminal of the microprocessor U2 and one side of the capacitive element CX, and the other side of the capacitive element CX. is grounded.The microprocessor U2 is a μPD75CG33E or μPD7533 manufactured by Nippon Electric Power Co., Ltd.
can be adopted. The second of the bridge rectifier BRI
One side of resistance element R6 and TRIA are connected to the AC input terminal.
The anode of the gate control device Q1, such as C, is connected, and the other side of the resistive element R6 is grounded. The other end of the capacitive element C1 is connected to the anode of the diode CRI, the cathode of the diode CR2, and the adjustment terminal of the Zener diode ZNI. The cathode of the diode CR1 is connected to one side of the capacitive element C2 and the "+■" terminal of the integrated circuit U1, and the other side of the capacitive element C2 is grounded. The "+■" terminal of integrated circuit U1 represents the supply voltage VY, which in the preferred embodiment of the invention is +10 VDC.

The anode of diode CR2 is connected to one side of capacitive element C7, and the other side of element C7 is grounded. The other terminal of Zener diode ZN1 is connected to the non-regulated terminal of another Zener diode ZN2. The other side or adjustment terminal of Zener diode ZN2 is grounded. At the connection between the device CR2 and the anode of the capacitive element C7 appears the supply voltage x, which in the preferred embodiment of the invention is -7 VDC.

Input terminal "1" on terminal board Jl is grounded.

Input terminal "3" on terminal board Jl is connected to one side of resistive element R3, and the other side of element R3 is connected to capacitive element C4.
is connected to the "RUN" input terminal of the linear integrated circuit U1 and the 841 terminal of the microprocessor U2.

The other side of capacitive element C4 is grounded. Terminal "4" of terminal board J1 is connected to one side of resistive element R4,
The other side of element R4 is connected to one side of capacitive element C5, the 5TART" input terminal of linear circuit U1 and the B42 terminal of microprocessor U2. The other side of capacitive element C5 is connected to the ground terminal. Input terminal “5” of board J1
” is connected to one side of resistive element R5, and the other side of element R5 is connected to one side of capacitive element C6, linear integrated circuit U1.
“RESET” input terminal of and microprocessor U2
Connect to the 843 terminal of The other side of capacitive element C6 is grounded. Resistive element/capacitive element combination R3-C
4, R4-C5, and R5-CB represent filter circuits respectively associated with input terminals "3", "4", and "5" of the terminal board J1.

These filters are connected to the input “RUN” of the linear integrated circuit U1.
'', 5TART'' and ``RESET'', respectively.

Between the DC or output terminals of the full-wave bridge rectifier BRI,
Connect the solenoid coil 31 which has already been described and is used in the manner described in more detail below. The other main conductive terminal or cathode of gate control device Q1, such as a silicon controlled rectifier, is connected to one side of resistive element R7 and to the "CCI" terminal of device U1. The other side of resistive element R7 is grounded. The gate of the gate control device Q1, such as a silicon controlled rectifier, is connected to a linear integrated circuit U.
Connect to the “GATE” output terminal of No.1.

The linear integrated circuit U1 is denoted by the reference symbol vz and is connected to the REF input input of the microprocessor U2 by "+5".
V'' power supply terminal, and a resistive potentiometer element R8 for adjustment.The integrated circuit module U1 has a VDD input terminal of the microprocessor U2, a capacitive element C1.
has an output terminal "VDD" connected to one side of capacitive element C9 and one side of resistive element R15, and the other side of element R15 connects to one side of capacitive element C9 and the "VDDS" of linear analog module U1. ”Connect to the input terminal. The other side of capacitive elements C9 and C16 is grounded. The linear integrated circuit module U1 also includes a ground terminal "GND" for connection to a common system or ground. Integrated circuit U1 has a terminal "RS" that provides a "RES" signal to the RES input terminal of microprocessor U2. Linear integrated circuit module or chip U1 has a capacitive element C8 on one side and a resistive element R14 on one side. Terminal “DM” to connect with
(DEADMAN). The other side of resistive element R14 is connected to the 022 terminal of microprocessor U2.

The other side of capacitive element C8 is grounded. The chip or circuit U1 has a "TRIG" input terminal which is supplied with the signal "TRIG" from the B524 child of the microprocessor U2. The integrated circuit U1 is the IN of the microprocessor U2.
It has a "■OK" output terminal that supplies the signal "VDDOK" to the TO terminal. Finally, the integrated circuit U1 sends the signal "Co I LCU" to the AN2 input terminal of the microprocessor U2.
It has a “CCO” output terminal that supplies the signal “C”.
0ILCUR'' indicates the amount of coil current flowing through the coil 31.The internal operation of the bipolar linear integrated circuit U1 and the operation of various outputs will be described later.

The other side of the resistive element R16 is connected to the anode of the diode CR4, and the cathode of the diode CR4 is connected to one side of the capacitive element C13, one side of the resistive element R17, and the AN3 input terminal of the microprocessor U2. Connecting. AN
The 3 input terminal receives a signal “LV” indicating the line voltage of the system under control.
The other side of the capacitive element C13 and the other side of the resistive element R17 are grounded.

The coil control panel 28 has a signal or function “GND” (
(ground), “MCUR” (input), “DELAY” (
A connector or terminal block J2 having terminals to which input), "+5■" (power supply), "+10■" (power supply) and "-7■" (power supply) are supplied is separately provided. Control signals Z, A, B, C and SW are also formed here.

Terminals GND and AGND of microprocessor U2 are grounded. Terminal AN2 of microprocessor U2 is connected to the "MCUR" terminal of terminal board J2, terminal CL2 of microprocessor U2 is connected to one side of crystal Y1, and the other side of crystal Y1 is connected to terminal CLI of microprocessor U2. Connecting. Terminal CL2 is also connected to one side of capacitive element C14. Also, the terminal CLI
is also connected to one side of the capacitive element C15. The other side of capacitive elements C14 and C15 is connected to the system ground. The terminal DVL of the microprocessor U2 is connected to the "+5■" terminal of the terminal board J2.

The linear analog circuit U1 has a built-in regulated power supply RP5,
Its input is the “+■” input terminal, and its output is “+5■”
In the preferred embodiment of the present invention, the unregulated 10 volt value VY is converted into a highly regulated 5 volt signal vZ or +5■ in the regulated power supply RPS. Also, in the preferred embodiment of the invention, the internal output source COMPO of the regulated power supply RPS, which is set at 3.2 volts, is connected to the reference (-) of the comparator COMP.

■DDS for one input (+) of comparator COMP
A signal is provided. The output of comparator COMP is VO
It is represented by K. Input terminal “LlNE”, “RUN”
, "5TART" and "RESET" are connected to the clip/clamp circuit CLA in the linear integrated circuit U1, and in the preferred embodiment of the invention, the micro Processor U
The range of the signal supplied to 2 is from +4.6 volts to -0
, 4 volts. The linear circuit U1 is “TRI”
It has a built-in gate amplifier circuit GA that receives the DEADMAN signal "DM" input and supplies the GATE output.
If the DEADMAN/reset circuit DMC receives the DEADMAN/reset circuit DMC and supplies the reset signal RES in “R3”,
An inhibit signal is also supplied to the gate amplifier GA in the "work" so that the gate amplifier GA does not output the gate signal GATE when the ADMAN function is performed. Furthermore, a coil current amplifier CCA is provided which receives a coil current signal from terminal "CCI" and outputs from terminal CCO an output signal Co I LCUR for use by microprocessor U2 in a manner to be described below.

The functions provided by microprocessor U2 at the various input/output terminals will be discussed below.

Connector J101 and connector J that are connected to the coil current control board 28 via the cable 64 and have a complementary relationship therewith.
An overload relay board 60 including 102 is also provided. Above current -
Voltage transducer or transformer 62 may be represented by three transformers 62A, 62B, 62C for a three phase electrical system controlled by overload relay board 60. One side of each secondary winding of these current-voltage transducers 62A, 62B, 62C is grounded, and the other side is connected to a resistive element R101, R1, respectively.
02, connect to one side of R103. Resistance element R1o
t, R102, F? , 103, respectively.

A triple two-channel analog multiplexer/demultiplexer or transmission gate U101 with cOR is also provided. ay of gate U101.

The by and CX terminals are grounded. Terminals ax, bx, and CX of gate UIO1 are electrically combined and connected to one side of integrating capacitor C101 and the anode of rectifier CR101. The other side of the capacitor C101 is connected to the cathode of the rectifier CR102, and the anode of CR102 is connected to the cathode of the rectifier CR101 and the differential amplifier U10.
3 and the bOR1 child of the second triplex two-channel analog multiplexer/demultiplexer U102. The other side of the integrating capacitor C101 is also connected to the positive input terminal of a buffer amplifier containing a gain U105 and to the cOR output terminal of the second analog multiplexer/demultiplexer or transmission gate U102. The collective terminals ax, bx, and cx of the transmission gate U101 are also connected to the ay and CX terminals of the transmission gate U102. The aX terminal of transmission gate or analog multiplexer/demultiplexer U102 is grounded. The aOR terminal of the device U102 is connected to one side of the capacitive element ClO2,
The other side of element ClO2 is the bxl element of multiplexer/demultiplexer U102 and the above differential amplifier U103.
Connect to the negative input terminal of The positive input terminal of the differential amplifier U103 is grounded. The negative input terminal of differential amplifier U105 is connected to the wiper of potentiometer P101, one main terminal of potentiometer P101 is grounded,
The other main terminal is connected to the terminal board J102 with “MCIJR,”
The output signal is supplied from one side of resistive element R103,
The other side of the resistive element R103 is connected to the output of the differential amplifier U105, the anode of the diode CR104, and the diode C.
Connected to the cathode of R105. Diode CR1
The anode of the diode CRI O4 is grounded, and the cathode of the diode CRI O4 is connected to the +5■ power supply terminal VZ. Device UIO1
, U102, and U2O5 are supplied with power from the -7 power supply. +t
The OV power supply voltage is the same as that of the gain amplifier U105 and resistance element 1.
04, and the other side of the resistive element 104 is connected to the power supply, the transmission gates U101, U2O5 and the anodes of the diode CRI O6,
The cathode of IO6 is connected to +5■ power supply voltage. The +5V power level Z of terminal board J102 is also supplied to one side of the filter capacitive element ClO3, whose other side is grounded, and to one main terminal of potentiometer P102, the other main terminal of potentiometer P102 is grounded. ing. The wiper of potentiometer P102 is connected to the microblower set "1-U2 (7) via the terminal board J101.
) provides a 'DELAY' output signal to child ANO.control terminal A of the analog multiplexer/demultiplexer unit U101.

B and C are parallel-serial 8-bit static shift registers U10.
Connect to the A, B, and C signal terminals of 4 respectively. signal A,
B and C are terminals 032.031 of the microprocessor 42
, 030, respectively.

An 8-pole switch 5WIOI is provided having poles AM, Go, C1, SP, HO, H1, H2, H3. One side of each switch pole is a parallel-series 8-bit static shift register U1.
5 volt power supply VZ via the PO to P7 input terminals of 04
The "C0M" output terminal of the register U104 receives the "SW" signal from the terminal board J101 and the terminal 110 of the microprocessor U2.
O'' through H3'' indicate that the device controlled by overload relay board 60 is of the "heater" class.

By suitably manipulating some or all of the four poles HO to H3 in switch SWI O1, a heater class device protected by overload relay board 60 can be represented.

The configuration of the printed circuit board used for manufacturing the coil control board 28 and overload relay board 60 will be explained with reference to FIGS. 2.8.9 and 10. Specifically, in addition to the terminal block J1, a coil assembly 30 is arranged on the coil control panel 28, and the coil of the coil assembly 30 is omitted from the drawing. The coil assembly 30 includes a spring seat 32 and a coil seat 31A. The coil control panel 28 is also provided with a connector J2, and one end of the flat cable 64 is inserted into the connector J2 by soldering or the like. The other end of the flat cable 64 reaches connectors J102 and J102 of the overload IT board 60. Figure 8 shows the three-phase current generator 62, assuming that the three phases are Wenzhou.
are shown as 62A, 62B, and 62C on the overload relay board 60. An 8-pole dip switch is provided as the switch 5W101. Also illustrated are potentiometers Plot and P102 used for factory calibration and delay adjustment, respectively.

In a preferred embodiment of the invention, the coil control board 28 and overload relay board 60 are formed on preformed, soldered, and connected wheel-piece printed circuit board material. The single piece printed circuit board material is then separated in the region 100, for example by folding the neck portion 102, to form overload relay boards 60 which are hinged to each other at right angles, as is particularly apparent from FIGS. 2 and 10. and a coil control panel 28.

Next, an embodiment of a control circuit configuration using devices and electrical elements of the coil control panel 28 and overload relay panel 60 will be described with reference to FIGS. 2 and 11. Specifically, three main power supply lines Ll, L2. L3 is provided to provide three-phase AC power from a suitable three-phase power supply. Each of these feeder lines is a contactor MA.

Power is supplied via MB and MC. Terminal block J1 includes terminals "C", "E", -P", "3", and "R", whose reference symbols indicate the function or connection "C", respectively.
0MM0N",'ACPOWER".

“RUN PERMIT/5TOP”, “5TART
-REQUEST", and RESET", for example, 348, 9.10. As already apparent from FIG. The overload relay board 60 includes a switch 5W101 that performs the above-mentioned functions, and has current transformers 62A to 6
A 2C secondary winding is connected to an overload relay board 60. Further, the secondary windings of the current transformers 62A to 62C are connected to the overload relay board 60. Current transformers 62A to 62C are connected to terminal T.
1. T2. T3 to line L1. L2. The line L1. which is supplied to the motor is connected to L3. Instantaneous line current iL1. flowing through L2, L3. iL2. Monitor iL3. Power is supplied to the coil control board 28 and the overload relay board 60, for example, via a current transformer CPT whose primary winding is connected between lines Ll and L2. The secondary winding of current transformer CPT connects to the "C" and "E" terminals of terminal block J1. Current transformer C
One side of the PT secondary winding can be connected to one side of a normally closed 5TOP pushbutton and one side of a normally open RESET pushbutton. The other side of the 5TOP pushbutton connects to the "P" input terminal of terminal block J1 and one side of the normally open 5TART pushbutton. The other side of the normally open 5TART pushbutton is connected to the “3” input terminal of terminal block J1, and the R
The other side of the ESET pushbutton connects to the reset terminal R of terminal block J1. By operating the push button in a known manner, the coil control panel 28 and the overload relay panel 6 are
0 can be supplied with control information.

The configuration and operation of various current transformers 62 of the present invention will be discussed with reference to FIGS. 2, 7C, and 12 to 18. Conventional current sensing transformers create a secondary winding current that is proportional to the primary winding current. The output current signal from this type of current transformer is fed into a resistive current shunt, and the shunt voltage is set at the overload relay board 6.
There is a proportional relationship between the input and output when supplied to voltage sensing electronics such as those incorporated in 0.0. An air core transformer, also called a linear coupler, can be used for current sensing by providing a secondary winding voltage that is proportional to the derivative of the current flowing through the primary winding. Traditional iron core current transformers and linear couplers have several drawbacks. Disadvantage 1
In the current transformer of the present invention, the current transformer's magnetic core The rate of change over time of the magnetic flux appearing in is proportional to the current flowing through the primary winding in the absence of magnetic flux saturation in the magnetic core. Since an output voltage proportional to the derivative of the current flowing through the primary winding is generated, and the ratio between the output voltage and current changes easily, it can be applied to various current detections. Iron core current transformers are relatively large, but the current transformer of the present invention can be made smaller.

As is particularly clear from FIG. 12, the current transformer 62 of the present invention
X includes an annular magnetic core 110 having a substantially discontinuous air gap 111. Since the primary current iL1, that is, the current to be detected, passes through the center of the magnetic core 110, a JIL winding input primary winding corresponding to the line L1 is formed. Current transformer 62X 2
The secondary winding 112 includes multiple turns having a number of N2 turns for purposes of illustration; the secondary winding 112 has a number of turns sufficient to output a voltage level sufficient to drive the electronic circuitry that monitors the current transformer. . For convenience of explanation, the circumferential length of the magnetic core 110 is set to 11, and the length of the air gap 111 is set to 12.

Let the cross-sectional area of the magnetic core be AI, and the cross-sectional area of the air gap be A2. The output voltage of the current transformer is varied by varying the effective length of the air gap 12.

This can be achieved either by inserting metal shims into the air gap 111 as shown in Figures 15 and 16, or by moving separate sections of the current transformer core structure to close the air gap as shown in Figure 17. 111 may be made smaller or larger. Once the length of the air gap 111 is set,
A relatively compact current sensing current transformer is formed which outputs an output voltage eo(t) approximately proportional to the derivative of the input current iL1 flowing through the input winding of the current transformer. One advantage of this configuration is that it does not necessarily require the use of sinusoidal or periodic input currents. For example, the output voltage eO(t) from the secondary winding of current transformer 62X shown in FIG. 12 is given by equation (1).

N1・N2 d e 0(t)” (ILL
sin (11t)11 J22 dt □+□ ·-(1) μIA1 μ2A2 μm and μ2 are the magnetic permeability of the magnetic core 110 and the air gap 111, respectively. ω (omega) is the instantaneous current iL1
and ILI is equal to the peak amplitude of the instantaneous current iL1, and if all parameters remain unchanged except the length of the air gap I12 and the frequency ω, equation (1) can be simplified to equation (2). Become.

N 1 ・ N2 eO(t)x [ωI Llco
s (11t) (2) 2 fL in kl+ However, the term in parentheses is equivalent to the derivative part of equation (1).

The voltage eo(t) in equation (2) is 113 as shown in FIG.
In a preferred embodiment of the present invention, an integrator circuit such as
The output of 3 is expressed by the following equation (3).

N 1 ・ N2 e'0(t) = ILlsin ωt
(3)kl+kH! 2 When the length of the air gap 111, I22, changes, the output voltage e'o(t), which is directly proportional to the input current iL1, changes inversely to the length of the air gap 111, ft2.

FIG. 14 shows changes in the length 12 of the air gap 111,
It is a graph showing the relationship between the output voltage e'o(t) divided by the input current (for example, i Ll). Primary station wave number ω
In the special case where is constant or is assumed to be constant, there is no need to use the integrating circuit 113 of FIG. can.

eO(t)w ILlcos ωt
(4) A part of 2λ2 constant frequency term ω is formed in kl+. In this case, the output 5O(t) from the current transformer secondary winding 112 is the input current IL
I and inversely proportional to the length fL2 of the air gap 111.

With particular reference to Figure 15.16.17, if it is desired to sense several current ranges using the same current transformer, the output can be changed by effectively varying the length j22 of the air gap 111. The voltage eO(t) can be varied. For this purpose, a shim of a predetermined width may be inserted into the air gap of the current transformer 62Y depending on the range of the required output voltage eO(t). Alternatively, a semi-core 119 may be inserted into the air gap 111 of the current transformer 62Z in a wedge shape.

Furthermore, a similar result can be achieved in current transformer 62U of FIG. 17 by dividing its magnetic core into two portions 116A, 116B and forming two complementary air gaps IIIA, 111B. Figures 12-17 illustrate a current-to-voltage transducer in which a primary winding is arranged on a magnetic core such that a magnetic flux is generated in the magnetic core that is approximately proportional to the amount of current flowing through the primary winding. The magnetic core has a discontinuous, but variable, air gap that has a first air gap that prevents magnetic saturation in the magnetic core at current values equal to or less than the value 11. Has magnetic resistance. Further, a secondary winding is arranged on the magnetic core so that a voltage V approximately proportional to the magnetic flux in the magnetic core appears at the output terminal. The voltage 2 is equal to or less than the voltage 2 for the first reluctance and a value of the current I equal to or less than 11.

A variable, but non-continuous air gap prevents magnetic saturation in the magnetic core for values of current I equal to or less than I2 greater than 11. It can be varied to obtain a higher magnetoresistance value than the resistance. For the second air gap magnetic resistance value and the current value less than or equal to 2, the voltage V maintains a value of 1 or less.

In particular, as can be seen from FIG. 18, - there is no apparently wide discontinuous air gap 111, but the air gaps 124 are evenly distributed between the fine-grained magnetic core material 122, e.g. A homogeneous magnetic core 120 made of sintered or pressed powder metal such as ferrite is connected to a current transformer 62S.
It can also be used for. Said air gap 124 has the same effect as a discontinuous air gap such as 111 shown in FIG. 12, but reduces the effects of stray magnetic fields, making it possible to realize a highly reliable compact current transformer. do. This type of current transformer is made by compressing powder metal.
22 and a magnetic core with microscopic and uniformly distributed air gaps 124 around the metal grains. A magnetic core constructed in this manner does not require saturation and produces an output voltage proportional to the derivative of the excitation current. One embodiment of the invention places a non-magnetic insulating material in the air gap.

Next, the operation mode of the system will be explained with reference to FIGS. 7A to 7D and FIGS. 11, 19, 20, and 21. The system line voltage (e.g. VA E5 in Figure 11) is LIN which is used to synchronize the microprocessor U2 with the AC line voltage.
Represented by the E signal. This generates various supply voltages, for example VX, VY, ■Z. Similarly, power
Detmann circuit D used as an on-reset circuit
MC first provides a 5 volt, 10 millisecond reset signal RES to microprocessor U2.

This signal initializes microprocessor U2 by setting the output of microprocessor U2 to a high impedance level and setting the internal program to memory location O. Switch inputs are read via inputs B41-843. The algorithm is as shown in FIG. Under normal conditions, terminal B41. B42. B43 is an input terminal of the microprocessor U2, but is also configured as an output terminal serving as a discharge bus for the capacitor for discharging. The reason is as follows. That is, when the input pushbutton is opened, C4, C5, and C6 may become charged as described above or due to leakage current from the microprocessor, which may be incorrectly interpreted as logic 1. Charge the capacitor to a dangerous voltage level. Therefore, it is necessary to periodically discharge the capacitive elements C4, C5, and C6. Logic in Figure 19
The "READSWITCHES" algorithm at block 152 asks the following questions: "Is the line voltage read from the line signal LINE at the 840 input terminal of the microprocessor U2 a positive half cycle?" If the answer to this question is "yes", then the respective input terminal B41
．． B42. A logic block 154 is utilized that checks whether the "5TART", RUN, and "RESET" signals at B43 are a digital 1 or a digital 0. Regardless of the answer, when the above question is asked, function block 1
In the next step of the algorithm shown in 56, the instruction "
DISC:HARGE CAPACITOR5'' is issued. At this point, terminals B41-B43 of microprocessor U2 are internally set to zero, discharging the capacitor as described above. This occurs during the positive half cycle of the line voltage. If the answer to the question posed in function block 152 is "no", then the line voltage is in the negative half cycle and it is in this half cycle that input terminals B41-B43 are released from capacitor discharge mode. , I explained about the motor control device, but
The invention can also be applied to devices that detect the presence of AC voltage signals.

After the initialization is performed, the microprocessor U2 checks the input terminal INTO to monitor the state of the VOK output signal from the linear integrated circuit U1. If the voltage of the random access memory RAM contained in the microprocessor U2 is high enough to ensure the authenticity of the already stored data, said signal will be a digital zero. Capacitive element C9 monitors and stores the power supply voltage VDD to the random access memory.

For example, during a power outage, even if the voltage VDD is removed by cutting off the power supply to the entire system, the capacitive element C9 maintains the voltage VDD for a while, but eventually discharges. The voltage of capacitive element C9 is VDDS and is again supplied to linear integrated circuit U1 in the manner described above. Whether the output signal VOK is set to digital 1, which indicates that voltage VDD is too low, or digital O, which indicates that voltage VDD is a safe value, depends on voltage VDDS.

Microprocessor U2 also receives an input signal LVOLT at its input terminal AN3.

This voltage between 0 and volts is proportional to the voltage on the control line LINE. Microprocessor U2 uses this information in three ways. (1) Used to select the contact closure profile of contactor 10 as already described in connection with FIG. The appropriate closure profile depends on the line voltage.

Signal LVOLT provides voltage information to microprocessor U2, which adjusts the firing phase or delay angle α1 . of gate control device Q1, such as a triac, in response to changes in line voltage. Change α2 etc. (2
) The LVOLT signal is also used to determine whether the line voltage is high enough to close the contactor 10 (see Table 1). There is a lower limit of line voltage or control voltage for reliable closing operation to occur, often this lower limit is 65% of the nominal line voltage. In the preferred embodiment of the invention, this is chosen to be 78 VAC. (3) Finally,
A microprocessor utilizes the LVOLT signal to determine if there is a lower voltage limit that will logically open the contacts at the appropriate time. This voltage is often 40% of the maximum voltage. When the line voltage signal LVOLT indicates that the line voltage is less than 50% of the maximum value, the microprosser U2 automatically opens the contacts to perform a fail-safe operation. In the preferred embodiment of the invention, this is selected to be 48 VAC. Microprocessor U2 reads the LVOLT signal according to the "READ VOLTS" algorithm of FIG. 20. The LVOLT signal is utilized in the "READ VOLTS" algorithm of FIG. Decision block 162 asks, "Is this a positive voltage half cycle?" This question and its answer are performed in the same manner as in decision block 152 in FIG. If the answer to the question at decision block 162 is "no", the algorithm returns to the starting point. If the answer is "yes", instruction block 164 instructs the microprocessor to select the AN3 power of microprocessor U2 for analog-to-digital conversion of the signals present based on the determination of decision block 162. This information is stored in a memory location in microprocessor U2 under the instructions of instruction block 168 for use in the manner described above, and the algorithm returns to its starting point.

Again in Table 1, the next input to the microprocessor is designated C0LCUR. This is part of the closed loop coil current control system. Input ccr to linear circuit U1
is the coil 31 according to the voltage drop across the resistive element R7.
Measure the current flowing through the This information is scaled appropriately as described above and transmitted to microprocessor U2 by the Co I LCUR signal. Just as we must know the line voltage given by the LVOLT signal,
The coil current given by the C0LCUR signal must also be known.

The Co I LCUR signal is "CHOLD" shown in Figure 21.
First, the microprocessor is instructed to fetch a captive conduction delay, as indicated in instruction block 172. Angle α7 is a constant conduction delay angle, e.g., 5 mm. the sum of the second and this captured amount.Then the microprocessor U
2 waits until an appropriate point in time, ie, angle α7 has elapsed, and fires the triac or silicon controller Q1 according to the instructions of instruction block 174. The microprocessor accomplishes this firing by issuing a "TRIG" signal from terminal B52, which connects the TRIG input terminal of integrated circuit U1 via amplifier GA and its GATE output terminal in the manner described in connection with FIGS. 7A and 7B. Supplying a silicon controlled rectifier triac or similar gate control device Q
Activate gate 1. Then, according to the instructions of instruction block 176, the current flowing through resistive element R7 and measured at the CCI input of semi-custom integrated circuit U1 is transferred via amplifier CCA to the CCO output of microprocessor U2.
ignites the C0ILCUR silicon controller Q1 to terminal AN2. The microprocessor accomplishes this firing by issuing a "TRIG" signal from terminal B52, which connects the TRIG input of integrated circuit U1 via amplifier GA and its GATE output terminal in the manner described in connection with FIGS. 7A and 7B. terminal to operate the gate of a silicon controlled rectifier triac or similar gate controller Q1. Then, according to the instructions of instruction block 176, the current flows through resistive element R7 and CCI of semi-custom integrated circuit U1.
The current measured at the input is passed through amplifier CCA to C
The CO output is applied as the COILCUR signal to terminal AN2 of microprocessor U2. The microprocessor repeats this Co I LCUR signal and outputs the A/D
Find the maximum value by converting. This maximum current is then compared in microprocessor U2 to the regulation point provided to microprocessor U2 as determined by decision block 178 to determine whether the maximum current is greater than the current determined by the regulation point. In a preferred embodiment of the invention, the regulation point peak current is set to have a DC component of 200 milliamps. This excitation level is maintained by varying angle α7 as required. If the answer to the question at decision block 178 is "yes", the conduction delay is digitally incremented upward to the next highest value within the microprocessor. This is done by incrementing the counter by at least one significant bit at a time.

As a result, the delay angle α7 of FIG. 6, for example, is larger and therefore the current pulse 124 is smaller, resulting in a smaller average half-cycle current flowing through the gate control device Q1, such as a triac.

Conversely, if the answer to the question at decision block 178 is "
If no, the count in the microprocessor is decremented by at least one valid pit, thereby reducing the delay angle α7 and increasing the current pulse 124.Function Block 1
Regardless of the answer to the question at 78, once the increments required by instruction blocks 180 and 182 are completed, the algorithm returns to its starting point for subsequent periodic use.

By varying α7 every half cycle as needed, the coil current will be maintained at the regulated value throughout the HOLD phase regardless of changes in drive voltage or coil resistance.

The human power LVOLT and Co I LCUR are important values that determine when the trigger signal TRIG is supplied from the output B52 of the microprocessor U2 to the trigger human power TRIG of the linear circuit U1. The linear circuit U1 uses the trigger signal TRIG in the manner described above to supply the gate output signal GATE to the gate terminal of the thyristor Q1 in the manner described above.

Line current iL1. iL2. An apparatus and method for detecting and measuring iL3 will be described with reference to Figures 22.23.24.25 and Figures 7A and 7B. As for transmission gate U101, its ax, bx, and CX output terminals are collectively connected to one side of integrating capacitor C101. Microprocessor U2 controls the parameter selection in gate U101 by providing signals A, B, C to the relevant inputs of transmission gate U1 according to the digital arrangement shown in Table 2. This operation allows the secondary winding voltages of current transformers 62A, 62B, 62C to be sampled sequentially in 32 half-cycle increments. Integrating capacitor C101 is charged in a manner described below. As already mentioned, current transformer 6
2A, 62B.

The secondary winding output voltage of 62C is determined by the line current iL1.62C flowing through the main feed lines A, B, and C, respectively. iL2. Associated with iL3 mathematical differences. This voltage is converted into a charging current by applying it to each of the resistive elements R101, R102, or R103, so the voltage of the integrating capacitor C101 V Cl0
I also changes every line cycle. The capacitor is discharged only after 32 line cycles of integration have been performed in the manner described below.

Margin table below 2 UIOI logic input Sensing current BA 1 1 0 i LA1 0
1 i LBo 1 1
i LCo 0 0
The transmission gate U102 operating in conjunction with the i-GRDZ input signal changes the connection relationship of the integrating circuit, and the integrating capacitor C101 periodically activates the circuit operation. This occurs when Z;0. The output voltage VCl0I of the integrating capacitor C101 is fed to a buffer amplifier U105 including a gain to form a signal MCUR, which is fed to the ANI power of the microprocessor U2. The microprocessor U2 is connected to the RA shown in FIG.
digitizes the data given by the signal MCUR in the manner of the ``NGE'' algorithm.The voltage signal MCUR is an 8-bit, 5
It is fed as a single analog input to the Volt A/D converter 200. A/D converter 200 is shown in FIG. It is desirable to use the system of the present invention in order to be able to measure line currents that vary over a wide range depending on the application. For example, depending on the stage, it may be necessary to measure line currents as high as 1.200 amperes, and there may be cases where it is desired to measure line currents of 10 amperes or less. In order to widen the dynamic range of the system, the microprocessor U2 expands the predetermined 8-bit output of the built-in A/D converter 200 to 12 bits.

For convenience of explanation, the above-described operation will be explained in detail using an illustrated example related to the sensing current transformer 62A and the resistor R101. In addition,
Current transformer 62B and resistor R102° and current transformer 62C and resistor 103 can also be used in the same way. Furthermore, di(t) eo (t) ~□ dt holds true for all current functions. Air gap 11 in current transformer 62A
Assuming that the length 12 of 1 is constant for a particular application (or current transformer 62S of FIG. 18 is used), i
Assuming that (t) is a sine wave, i.e., I Ll sin ωt, the output voltage of the current transformer defined by equation (1) can be rewritten as shown in equation (5) below. .

K5 d (ILL sin ωt )eo (t
)=□ dt , , (5) The output voltage eo(t) is applied to the resistor R101 and the charging current i of the integrating capacitor C101 according to equation (6)
Converted to CH. This is expressed in unit amplitude (P, U,) in a graph as shown in FIG. 25B.

eo (t) K6 d (ILI sin
ωt)i CH= −= (6
) The charging current iCH of the RIOI d7 integral capacitor C101 is proportional to the derivative of the line current iL1 rather than the line current itself. Consequently, as is clear from equation (7), the voltage V Cl0I of the capacitive element C101 present as a result of the charging current 1CH(1) flowing during the negative half cycle can be expressed as:

Below margin 1 K6 d(I Ll sin ω
t) dtv clat = --(7) CIOI RIOI dtVG10
1=-7 ILL sin ωt
(8) Equation (8) represents equation (7) in a simpler form. FIG. 25A is a graph showing I Ll sin ωt expressed in per units (P, U,). The derivative of iLl sinωt after being integrated by capacitor Cl0I, i.e.
−7 ILL expressed in unit amplitude (P, U,)
FIG. 25C shows that sin ωt is incorporated. The charging current icH of the capacitive element C101 comes from the output terminal aX of the transmission gate U101. This current is supplied from the aOR input terminal to the transmission gate U101
(see Table 2). Similarly, the current from current transformer 62B is b
It can be used by selecting the oR-bX terminal, and the current from current transformer 62C can be used by selecting the cOR-cx terminal. The i4 children ax, bx, and cx are grouped together to form a lead and provide charging current to the integrating capacitor C101. The single lead connects to the ay and CX terminals of the transmission game 1-U 102. a of transmission gate U102
The x terminal is grounded, and the aoR common terminal is connected to capacitor C.
101 on one side. The cOR terminal is connected to the other side of capacitor C101. The bx terminal of the transmission gate UIO2 is connected to the negative input terminal of the operational amplifier U103, and the associated bOR common terminal is connected to the output of the operational amplifier U103. Under normal conditions, the diode circuit CRIOI-CR
103 is during the integration operation, the positive half cycle of the integration current ICH is connected to the diodes CR101, CR102 and the operational amplifier U.
Integrating capacitor C101 is bypassed through a bridge circuit including the output of 103, and the negative half cycle is configured to charge capacitive element C101 to the peak value of that half cycle. The capacitive element C101 is repeatedly charged to a gradually higher voltage value, and the charging voltage value each time corresponds to the peak value of the negative half cycle of the charging current.

It is not uncommon for a voltage as small as 0.25 millivolts to exist between the negative and positive input terminals of operational amplifier U103. The capacitive element ClO2 is periodically charged to the negative of said voltage value to form a zero net human offset voltage for the amplifier IU 103, ie a charging current iCH.

The "RANGE" algorithm performed in conjunction with the above integration circuit including the capacitive element C101 and the microprocessor U2 will be explained with reference to the examples shown in FIGS. 22, 23 and 25. It goes without saying that the dynamic range for detecting line current is important. However, as is clear from FIG.
D converter 200 has an input voltage upper limit that guarantees a reliable number of digital outputs. In the preferred embodiment of the present invention, A/D converter 200 operates at +5 volts to form an 8-bit signal that is applied to the first eight locations 204 of accumulator or storage device 202 located in the memory of microprocessor U2. It can tolerate input voltages up to In this case, the 5 volt upper limit input is represented by decimal number 256, which corresponds to the digital number in all eight locations of section 204 of accumulator 202.

FIG. 25B is a typical graph showing the amplitude change over time of the current i Ll sin ωt. The graph in Figure 25A is the charging current iCH which is the derivative of the line current in Figure 25B.
shows. Also, Figure 25A shows that only the negative half cycle of the current is integrated. In FIG. 25B, appropriate amplitude standards 220, 230, and 240 are taken as three examples, and the differences between the 1-vehicle amplitude, the station-level amplitude, and the 2-unit amplitude are illustrated. Amplitude 220A, 2 in the graph of Figure 25A
30A and 240A respectively correspond to the unit amplitude in the curve shown in FIG. 25B. Similarly, two curves 230B and 220B are illustrated as Examples 1 and 2. Second
246 in Figure 5C is the maximum input terminal of 5 volts. the second every half-cycle for 32 consecutive half-cycles.
The algorithm shown in Figure 2 is performed. Each half cycle during this time interval is identified by a number stored as HCYOLE. Half cycles 2, 4, 8.16 and 32
each represents an integration interval that is twice the previous half cycle. It is at the end of these defined intervals that the algorithm re-evaluates voltage VC101.

Assume that the input signal repeats every cycle in 32 intervals. In that case, HCYOLE=2.4,
The voltage VCIOI at the end of the interval denoted 8.16 or 32 will be twice the size at the end of the previous interval. Therefore, if the result of an A/D conversion in the previous interval is greater than or equal to 2.5 volts,
If it is 80H or more corresponding to the OI value, the VCIOI at the current interval will be 5 volts or more, and the A/D conversion result will be invalid. This is because the A/D converter cannot digitize values higher than 5 volts. Therefore, if the previous result is greater than or equal to 80H, the algorithm retains this result as the best possible A/D conversion.

Conversely, if the preceding A/D conversion is 80H or less, it may be considered that effective A/D conversion can be performed.

The current signal can be more than twice the previous value,
This is because it is less than 5 volts. The current A/D conversion is advantageous in that the size of the converted signal is twice as large as that of the previous A/D conversion, and a resolution with a larger number of bits can be obtained.

If the result of A/D conversion is found to be 80H or more, the interval at which A/D conversion was performed must be taken into consideration for adjustment. A left shift operation 188 performs this function. For example, the result obtained at the end of interval 4 is 80
H is the result of an input signal having twice the magnitude of the input signal producing result 80H at the end of interval 8. Therefore, shifting the result of interval 4 to the left doubles this result by the end of interval 8. The 12 bit answer contained in accumulator 202 of FIG. 23 at the end of 32 half cycles represents at least an approximation of the value of the current flowing through the line being measured.

It is this value that the microprocessor U2 utilizes in order to control the contactor 10 in the manner already described and to be described in more detail below. At HCYCLE 33, the entire process is then reinitialized for use with current transformer 628B and then 62C. It goes without saying that this initialization is repeated periodically by the microprocessor U2 in a known manner.

Straight line 220B in FIG. 25C shows that voltage V fl:101 increases with integration of current iCH in FIG. 25A. Integration is not performed in the positive half cycle of charging current iCH,
Integration is performed to draw a negative CO3 curve every negative half cycle. These integral values are accumulated to form the voltage V Cl0I. Therefore, for 33 half cycles the capacitive element C
101 increases with the value of the sampled line current over a time represented by 32 half cycles.

Next, the embodiment of the accumulator according to Example 1 will be explained along with FIGS. 22, 24, 25 and 26. Example: In order to charge C1ot and generate the capacitor voltage VCIOI in the capacitor 1, a charging current 1cH230a having a monthly amplitude is used. The 5th C is a summary of this voltage profile.
This is 230b in the figure. This voltage is sampled by the "RANGE" algorithm according to function block 184 of FIG. 22. "2", '4', 8s, '1
6” and 32”HCYCLE benchmarks, “
RANGE” algorithm is function block 1 in Figure 22.
86, the preceding A/D conversion result is 8.
Determine whether it is greater than or equal to 0 hex. 80 px is equal to 128 digital numbers. If the answer to this question is “no”, the input ANI of the A/D converter 200
The analog voltage VCIOI present at VCIOI is digitized and stored as shown in functional block 192 of FIG. 22 and graphically in FIG. HCYOLE
is incremented by one and the routine resumes. Advance A/D
As long as the conversion result is 80 hexes or less, there is no need to utilize the "left shift" technique of the present invention. Accordingly, Example 1 of FIG. 26 illustrates a sampling routine that does not require the use of left shift techniques. That is, in Example 1 of FIG. 26, when HCYCLE=2, the A/D converter 2
0.2 volts are obtained at the input terminal ANI of 00, which is digitized into a binary number corresponding to decimal 10. This binary number has digital ones in the "2" and "8" positions of memory portion 204 and digital zeros in all other bit positions. "HCYCLE 4" is analog voltage 0.4
The bolt is digitized and “16” of the memory part 204 is
and '4'' form a decimal number 20 with a digital 1 in the bit position and a digital 0 in all other positions.
Digitize 0.8 volts in ``CYCLE8'' to form a binary number equivalent to decimal 40 with digital 1s in locations ``32'' and ``8'' of memory portion 204. 1.6 volts in ``HCYCLE16''. is digitized to form a digital number representing the decimal number 81. This digital number is stored in memory portion 204 as "64" and "
It has a digital 1 at the 8” position.
At CLE=32'', 3.2 volts is digitized to form a digital number equivalent to 163 decimal.

The digital number is “128” in the accumulator 204, '3
At this point, the "RANGE" algorithm for Example 1 is complete.
The ANGE" algorithm does not proceed to function block 188 that requires a left shift. However, as discussed below in connection with Examples 2 and 3, a left shift may have to be utilized.

Next, with reference to FIGS. 22.24.25 and 27, a second example will be described in which a charging current 1 cH220a with an amplitude of 1 degree is used to generate the voltage vctot in the capacitive element C101. 220b in FIG. 25C depicts the generated voltage in comparison with HCYCLE. Again, the second
Example 1 where the 'RANGE' algorithm in Figure 2 is used
As in the case of 2", "4", -8", "16"
and memory location 2 in the 32”HCYCLE sample
The "RANGE" algorithm is used so that 02 is updated. 0 in the “2” HCYCLE sample.
The four volts are digitized to form a digital number corresponding to the decimal number 20 in the accumulator 2020 portion 204. This digital number has digital 1's in the "16" and "4" bit positions of portion 204 and digital 0's in all other bit positions. At HCYCLE=4, 0°8 volts is digitized to form a digital number corresponding to decimal number 40. This digital number is stored in the accumulator 202
has a digital 1 in the "32" and "8" bit positions of the portion 204. At HCYCLE=8, 1.6 volts is digitized to form a digital number in portion 204 of accumulator 202 corresponding to 81 decimal digits. This digital number consists of bit positions “64”, “16” and “1”.
“ has a digital or logical 1.HCYCLE=1
6, digitize 3.2 volts to decimal number 1
A digital number corresponding to 63 is formed in portion 204 of accumulator 202 . This digital number is bit position '1
28", "32", '2" and "1" have digital ones. At HCLYCLE=32, the RANGE” algorithm uses function block 186 to
It is determined that a digital number larger than 80 pixels has been formed as a result of the previous A/D conversion. Therefore, the function block 188 is utilized for the first time, and the "left shift" function is used for the first time.
will be held. As a result, even though there is 6.4 volts at the input of the A/D converter 200 to be digitized, the output of the A/D converter cannot be trusted with such a large analog number at the input. For this reason alone, digitization was not carried out, and the previous 3.
During digitization of a 2 volt analog signal, the digital number stored in portion 204 of accumulator 200 is shifted to the left by one place for each bit of the digital number.
Form a new digital number corresponding to the base number 326. This new digital number utilizes a portion of the spill over portion 206 of the accumulator 202, as shown in FIG. New digital number expanded accumulator 20
2's "256", "64", "4N" and "2"
It has a digital 1 in the bit position. 32”H in Figure 27
The digital number at the CYCLE location is the same as the digital number shown at HCYOLE location "16" but shifted to the left by one bit position. This example illustrates aspects of the left shift technique. The number stored in accumulator 202 at the end of the 32nd HCYCLE indicates whether the contactor 10 is overloaded, praised or not. 22. Line current measured in '+ system and G).
A third example of the left shift technique will be described with reference to FIGS. 24 and 25 and 28. In example 3, in order to obtain the voltage VC101, the second
The 24th amplitude charging current iCH shown at 2408 in Figure 5B is integrated by the capacitor C101. This voltage exhibits an output profile similar to that shown in FIG. 25C in connection with Examples 1 and 2, but exhibits a slope as outlined in FIG. 25C as Example 3. To avoid confusion, ignore the step relationship between voltages. However, much like in Examples 1 and 2, there is a step voltage in Example 3 as well. For example 3, the "RANGE" algorithm is HCYCLE
= "2", "4" and "8" and performs appropriate A/D conversion.
204 is updated. However, ) (accumulator 20 in CYCLE samples “16” and “32”)
Part 2 is updated not by A/D conversion, but by two successive sequential left shifts of the prior information stored in location 204.

It is clear that A/D conversion does not provide reliable results for sampling at "16" and "32". Specifically, in HCYCLE; “2”,
Digitize 0.8 volts to form a digital number equivalent to 40 decimal. This digital number has a digital number 1 at bits "32" and "8" of portion 204 of accumulator 202. In the “4” HCYCLE sample, 1.6 volts is digitized to 81 decimal.
form a digital number equivalent to . This digital number is “64” and “16” in the portion 204 of the accumulator 202.
and has a digital number 1 in the "1" bit position. HCY
At CLE=8, 3.2 volts is digitized to form a digital number equivalent to 163 decimal. This digital number is “1” in portion 204 of accumulator 202.
It has digital ones in the 28'', '32'', 2'' and 1'' bit positions. At HCYCLE=16, “RANG
E” algorithm is such that the previous A/D conversion result (corresponding to the digital number 163) is larger than 80 hexes, and therefore the accumulator 202 converts the voltage at the input of the inverter 200 into A/D. Instead, we recognize that the digital information already stored in accumulator 202 as a result of the completion of HCyclt, i = "8" samples is updated by shifting it to the left by one bit. As a result, "16" HCYCLE samples A digital number corresponding to the decimal number 326 is formed at . This is accomplished by shifting the information already stored in the accumulator to the left by one bit.
A spillover portion of 206 overflows to the 1-bit position.

This new digital number is “25” in accumulator 202.
6”, “64”, “4” and “2” bit positions have digital 1s. At the HCYCLE=“3” sample, by left shifting the number already stored in accumulator 202 once again in accumulator 202. , occupying two locations in spillover portion 206 and all eight locations in portion 2°4. This new digital number corresponds to the decimal number 652, and “
It has a digital 1 at the 512” position, the 12B position, the 8 bit position, and the 4 bit position.Using this number, the line current measured via the overload WE switchboard 60 can be expressed. In addition, the values stored in accumulator 202 are utilized in the manner described above to cause functions to be performed by contactor or controller 10.

Referring again to FIGS. 7A-7D, apparatus and methods associated with switch 5W101 and 8-bit static shift register U104 will now be described. Inputs HO through H4 of switch 5W101 represent a switch configuration that programs digital numbers that can be read by microprocessor U2 to make decisions and perform calculations regarding the ultimate value of the full load current sensed by the system. These switch values and the switch values associated with AM", "co", 01" are sequentially processed by the microprocessor U2 as part of the signal appearing on line SW in response to the input information provided by the A, B, C input signals. be read. Utilizing the heater switch configuration, four heater switches HO through H3 are programmed in a binary manner to
Six ultimate trip values can be selected.

These switches replace known mechanical heaters to adjust the motor overload range. Also provided are two inputs CO and C1 which are used to power the motor class. A class 1o motor can withstand 10 seconds of rotor locking without damage, a class 20 motor can withstand 20 seconds of rotor locking, and a class 30 motor can withstand 30 seconds of rotor locking. The current in the rotor locked state is assumed to be six times the normal current.

Referring again to FIGS. 7A and 7B, FIGS. 11 and 29,
An apparatus and method for discriminating true input signals from false input signals appearing at the "RUN", "5TART" and "RESET" inputs is described. FIG. 11 shows the distributed parasitic capacitance CLL between the power lines connecting the "E" and "P" terminals in the terminal block J1 of the relay board 28. This capacitance is connected to the pushbuttons “5TOP”, 5TART” and “
This is believed to be caused by the presence of an extremely long input line between RESET" and terminal block J1. Similar capacitance may also exist between the other lines shown in Figure 11. The parasitic capacitance is It has the undesirable effect of coupling signals between input lines, resulting in
Pushbuttons "5TOP", '5TART' and 'RESE'
A false signal is generated which the microprocessor U2 misinterprets as a true signal indicating that T" is in the closed state when it is actually open. Therefore, the purpose of the following device is to distinguish between the true signal appearing on the input line and The capacitive current i CLL flowing through the distributed parasitic capacitance CLL precedes the voltage in said capacitance, i.e. between terminals "E" and "P". a) shows the truncated VLINE received by microprocessor U2. FIG. 29(e) shows the pseudo current 1
This voltage VRUN is a false indication of the voltage that is indicative of the voltage that appears at e.g. terminal B41 of microprocessor U2 as a result of CLL flowing through resistive element R3, capacitive element c4 and the internal impedance at the RUN input terminal of circuit U1.
(F) precedes the voltage VLINE by a value γ. If capacitive elements CX and C4 are different from each other, specifically, if capacitive element CX is larger than capacitive element C4, then the true VR
The UN signal VRUN (T), ie, the signal generated by closing the 5TOP switch as shown in FIG. 11, is approximately in phase with the voltage VLINE.

The only difference between the two is due to the difference in capacitance between the capacitive elements CX and C4. If capacitive element CX is smaller than capacitive element C4, this difference will cause the true voltage VRU to
N (T) is VL by the amount Δ as shown in FIG. 29(b).
It lags behind INE. Therefore, the microprocessor U2 changes state when the voltage VLINE changes state, i.e., as shown in FIG. 29(a).
After passing through the change points "UP" and "DOWN", the voltage VLINE must be compared with the voltage at the input terminal B41 within a short time of Δ or less. The digital value of the voltage appearing at terminal B41 is the voltage VL at this point.
If the digital signal has the opposite polarity to the digital value associated with INE, this signal is a true signal as shown in FIG. 29(b). If the polarities are the same, it is a false signal as shown in FIG. 29(c). That is, for example, if the voltage VLINE is measured within a time Δ following the time ``UP'' and compared to the electrician appearing at terminal B41, the voltage at terminal B41 becomes digital O
Then, the voltage signal at terminal B41 is a true signal. but,
If the voltage signal is a digital 1, the voltage signal appearing at terminal B41 is a false signal. By appropriately setting the values of the capacitive elements CX and C4, it is possible to change the amount by which the true signal precedes the line voltage, that is, the amount of delay Δ. Since the value of Δ is smaller than the value γ, it is also impossible for the sign of the false signal to differ from the sign of the reference voltage during the sampling or comparison interval.

FIG. 30 shows an alternative embodiment of the printed circuit card also shown in FIGS. 8.9 and 10. In the embodiment of FIG. 30,
Elements that are the same as those of the apparatus shown in Figures 8.9 and 10 are given the same reference symbols with a dash (°). In the devices of Figures 8, 9 and 10, connect solder connector J2 to Jl.
A flat connector 64 is used to connect the OL and J102, but in the embodiment shown in FIG.
Instead, an electrically insulating base 300 is provided, on which a male plug connector 303 is placed. Connector 303 is shown above the overload m electrical board 60°. A female connector 302 corresponding to a male connector 300 of the relay board 60° is provided on the printed circuit board 28''. The female connector 302 has a recess or hole 304 that is complementary to the male plug 303 of the connector 300. As will be described later with reference to FIGS. 32 and 32, the bobbin 32' is soldered through a pin 318 inserted into an appropriate hole formed at the circuit board 28° in order to more securely support the circuit board 28''. Connect to circuit board 28゛. As in the case of the embodiment shown in Figures 8.9 and 10, after assembly the entire circuit board is folded to 100° and is illustrated in Figures 31 and 32 and described in connection with these Figures. The connector 302 is made to correspond to the connector 303 in this manner. Additionally, a separate internal communication circuit (IUCOM
) Terminal block JX is installed separately for connection.

31 and 32 show an embodiment of the invention similar to that shown in FIGS. 1 and 2. In this embodiment, elements that are identical or similar to elements of the apparatus shown in FIGS. 1 and 2 are given the same reference numerals with a prime (') added. Reference is made to the description associated with FIGS. 1 and 2 for a discussion of the cooperation, function and operation of the elements of FIGS. 31 and 32 which are identical or similar to those comprising the apparatus of FIGS. 1 and 2. Relay board 6
0° and printed circuit board 28° are shown in the fully assembled state with plug 303 connected to female connector 302 in the manner described above. That is, the male connector 303 is the female connector 3.
By being inserted into and making electrical contact with 02,
i4 is connected to the 60° element of the electrical board to the element of the printed circuit board 28'. Also, for example, the electric board 60' shown in FIGS. 31 and 32 is connected to the circuit board 28' leaving an offset section where a supplementary terminal block JX is disposed. In the embodiment shown in Figures 31 and 32, the contactor has terminal straps of 20° and 24° and terminal lugs of 14°.

16', and a one-piece thermoplastic insulating base 12' holding fixed contacts 22', 26'.
0 to secure the fixed contact and terminal strap to the base. The base 12° also includes the movable contacts 46° and 48°, the crossbar 44°, the spacer or carrier 42', and the armature 40°, which will be described in detail later! Acts as a guide/guidance system. The overload relay board 60' and the coil control board 28' are uniquely supported within the base 12'. Specifically, a permanent magnet or slug 36° that is identical or very similar to the armature 40° (as is particularly clear from FIG. 32) has a lip 329.
, which is pressed against a corresponding lip 330 on the base 12'' under the action of a retaining spring or member 316, which connects the slug or permanent magnet 36° to the base 12''. Slag or permanent Mi53
6'' has (as is particularly clear from FIG. 31) a second lip 314 which presses against a corresponding lip 315 on the bobbin 317 of the coil assembly 30'. 318 is provided, and the coil control panel 2
8' by soldering or the like, and fixedly supports a coil control board 28° including a flexible electrical insulating material at its center.

The 28° corner of the coil control board is supported directly on the base 12°, for example at 320. Overload relay board 60' is supported vertically above coil control board 28° by the interaction of bins and connectors 300, 302, 303 and 304. The other end of the coil assembly 30' is connected to the kick act spring 3.
4' and thus the bobbin 317 is supported by the spring 3
Due to the compressive force of 4°, the lip 31 of the magnet 36°
4 and the base 12'. 32, the top of the spring 34° is anchored to the bottom lip 340 of the carrier or spacer 42°, and the movable contacts 46', 48', the spacer 42" and the armature 4
During movement of the movable system including 0', it moves together with the carrier or spacer.

FIG. 32 shows magnetic members 36° and 4 that are approximately E-shaped.
0゛ configuration and interactions are illustrated. The movable armature 40' has a central leg 322 and two outer legs 330, 33.
In order to obtain the function of tightening the magnet 40°, including 1, it is sufficient that the legs 330 and 331 have cross-sectional areas that are slightly different from each other. This is because, during repeated use, the front surface of the magnetic disk outer legs 330, 331 repeatedly collides with the front surface of the magnetic slug or permanent magnet 36' which is in a complementary relationship with the magnetic slug, thereby creating a wear pattern. Therefore, the magnetic member 40' is used for maintenance purposes.

36" is removed periodically, it is desirable to reassemble it in its exact original orientation so that the existing wear pattern remains intact. Both members 40', 36" are preferably reassembled in their original orientation with respect to each other. If assembled in the opposite orientation, a new wear pattern will occur which is undesirable.If the sum of the cross-sectional areas of legs 330 and 331 is set to be approximately equal to the cross-sectional area of leg 332, effective magnetic flux conduction will be achieved. In a preferred embodiment of the invention, a large portion of the face of the central leg 332 is cut to form a protrusion or nipple 326 and two effective air gap regions 327, 328. When it comes into contact with °,
The complementary outer board legs 331 and 330 are in surface contact, and the center leg 3
The front portions of the nipples or protrusions 326 of 22 abut face-to-face in areas 327, 328 of both magnets, leaving a wide air gap. The presence of the air gap acts to reduce the residual magnetism of the magnetic circuit formed by the abutting armature 40° and the permanent magnet 336°. This is desirable because during the contact opening operation, the kickout spring 34° separates the magnetic member and opens the contact. It is known that magnetic noise can be introduced in magnetic structures subjected to the action of an OLD pulse (alternating or periodic). If the nipple 326 were not present, the HOL
Central leg 3 of movable armature 40° under the action of D-pulse
22 vibrates in the same way that the magnetic core of a radio speaker vibrates in the presence of drive 48. Furthermore, under the action of the periodic HOLD pulse, the rear projection 333 of the armature 40' is distorted towards the center, and the leg 330. Move as if rubbing. As a result, undesirable surface wear increases. A nipple or protrusion 326 is formed to prevent the distortion and wear and to ensure an air gap. This prevents the leg 322 from moving under the action of the HOLD pulse, yet reduces the residual magnetism to a level that does not interfere with the operation of the kickout spring 34°. Now, the operation of the current transformer 62S shown in FIG. 18 will be explained. This description applies to electrical devices such as contactors, meters, relay systems, etc.
23, the voltage output ■ proportional to the time derivative of the current i flowing through the primary winding can be expressed as
This is extremely useful in understanding the manner in which this occurs in the next winding. By utilizing this, it is possible to measure or detect a wide range of currents i, and the error in the output voltage (2) for a wide range of currents 1 is extremely small. For example, from 0.1 amp to 20
Output voltage V for current range varying up to 00 amps
is a relatively faithful representation of the magnitude of the derivative of current i with an error of no more than 1%. (R, M, Bozort
“Ferro-magnetism” 9.489 by h.
(as shown in Figure 11-11), if iron powder is used for the magnetic core 120 in Figure 34, the magnetic permeability μ depending on the magnetic field strength H will be in a region called the Raysaw region or Rayleigh range. takes an approximately constant value μO over a relatively wide range of known magnetic field strengths. The Rayleigh constant count equal to the slope (λ) of the magnetization curve is approximately equal to zero over the Rayleigh range of iron powder. Examples of gradient values for ferromagnetic materials other than iron powder are AA, BB, and CC, where the gradient λ is not equal to zero. Equation (9) shows the relationship between the magnetic permeability μ according to the initial magnetic permeability μ0, the Rayleigh constant count, and the strength H of the liB anchor in the Rayleigh range, regardless of the type of ferromagnetic material.

μ=μO+EH・・・・・・・・・(9) NI H=□・・・・・・(10) Equation (
10) shows the relationship between the magnetic field strength H and the current i.

μNli to Φ=β=μHmi□ (11) Sentence equation (11) shows the relationship among magnetic flux (φ), magnetic flux density (β), and magnetic field strength H).

dφ V=N 2□ ・・・・・・・・・(12)d
t Equation (12) shows the relationship between the output voltage according to the magnetic flux appearing in the secondary winding of the current/voltage transducer 62S and the number of turns N2 of the secondary winding.

NlN2A dμi V=□ □ ・・・・・・・・・(13)λ
dt Equation (13) represents equation (12) in a form in which its constant values are bundled.

Below Margin NlN2A d(μO+λH)i■ =□
□ λ dt・・・・・・
(14) Equation (14) shows the form obtained by substituting equation (9) into equation (13).

If NlN2A di λ=0, ■=□μO□...(15) Akira
dt Equation (15) shows equation (14) for the special case where μO is constant.

If di λ=:IO, ■=□ ・・・・・・(16)
dt Equation (16) is a simplified form of equation (15) and is within the excitation or non-saturation region of the magnetic core, e.g. 6i1 core 120.

[Brief explanation of the drawing]

Figure 1 is a perspective view of the electromagnetic contactor, Figure 2 is Figure 1 II-
11 line; FIG. 3 is a graph showing the force and armature velocity curves of a known contactor with a T6N armature acceleration coil, kickout spring and contact spring; FIG. FIG. 5 is a graph showing a family of curves similar to the curves of FIGS. 3 and 4, but for another embodiment of the invention; FIG. The figures are Figures 4 and 5.
Graphs 7A to 7D showing curve groups respectively corresponding to voltage and current waveforms in the device embodiments corresponding to the figures.
The figure is a circuit diagram showing a partial block diagram of the electrical control system in the contactor shown in Figures 1 and 2; Figure 8 shows the circuit elements of Figure 7, the contactor coil, current transformer, and A plan view of the printed circuit board containing the transformer; Figure 9 is an elevation view of the circuit board shown in Figure 8; Figure 10 shows the circuit board of Figures 8 and 9 attached to the contactor of Figure 2. FIG. FIG. 11 is a circuit/wiring diagram showing, in partial block diagram form, the contactors of FIGS. 2 and 7 used in conjunction with a motor controlled thereby; FIG.・Voltage transducer configuration diagram, 13th
The figure is a block diagram schematically illustrating the transducer of Fig. 12 together with an integrating circuit; Fig. 14 is a graph showing the relationship between air gap length and voltage/current ratio in the transducer of Figs. 12 and 13; Fig. 15 The figure shows the current using magnetic shims.
FIG. 16 is a block diagram showing an embodiment of a voltage transducer; FIG. 16 is a block diagram showing an embodiment of a current-voltage transducer using an adjustable protruding member; FIG. 17 is a current-voltage transducer using a movable magnetic core. A configuration diagram showing an embodiment of a transducer;
Fig. 18 is a block diagram showing an embodiment of a current-voltage transducer using a powdered gold R magnetic core; Fig. 19 shows a switch for the input circuit in the coil control panel shown in Fig. 7 and a capacitor. The algorithm used by the microprocessor for discharging is “READSWITCI (ES)”
FIG. 20 is a block diagram showing the algorithm "READVOLTS" for reading the line voltage in the coil control panel shown in FIG. 7; FIG. 21 is a block diagram showing the coil in the coil control panel shown in FIG. Algorithm for reading current “CHOL D”
FIG. 22 is a plotter diagram showing the algorithm ``RANGE'' for reading the line current determined by the overload N switchboard shown in FIG. 7. Fig. 23 is a simplified diagram showing the A/D converter and memory location that cooperate with line current reading by the microprocessor in the coil control panel of the present invention; Fig. 24 is a coil control try hook in the coil control panel shown in Fig. 7. The algorithm used by the microprocessor to start
IRE TRIA C'': Figure 25A is a graph showing the derivative of the line current shown in Figure 25B; A graph shown as a sine wave, FIG. 25C is a graph showing the relationship between A/D converter input voltage and half-cycle sampling interval (time) corresponding to the three line current amplitudes shown in FIG. 25A.
Figure 6 shows the line cycle for each station and the RANGE in Figure 22.
An array diagram of binary numbers corresponding to the first example of A/D conversion stored in the memory location of the microprocessor shown in FIG. 23 by six samplings performed according to the sampling routine; FIG. 27 shows one unit. 2nd in line cycle
FIG. 28 is an array diagram of binary numbers corresponding to the second example of A/D conversion stored in the memory location of the microprocessor shown in FIG. 23 through six samplings performed according to the RANGE sampling routine of FIG. 2; is a 2 unit line・
A stored in the microprocessor memory location shown in FIG. 23 by six samplings performed in accordance with the RANGE sampling routine of FIG. 22 in a cycle.
An array diagram of binary numbers corresponding to the third example of /D conversion: Figure 29 shows VLINE and VRI in the manual operation of a microprocessor.
FIG. 30 is a plan view of a printed circuit board similar to that shown in FIGS. 8 and 9 utilized in another embodiment of the invention; FIG. 31 is a vertical cross-sectional view of a contactor similar to that shown in FIGS. 1 and 2 in another embodiment of the invention; FIG. 32 is a vertical cross-sectional view of the contactor of FIG. A cross-sectional view taken along the line II; FIG. 33 is a partially sectional perspective view of a compressed powder iron current-voltage transducer having a magnetic core as shown in FIG. 18; FIG. 34 shows the 6n field strength of various magnetic materials; It is a graph showing the relationship of magnetic permeability.

Claims (1)

[Claims] 1. A magnetic core, and a primary winding that electromagnetically interacts with the magnetic core to generate a magnetic flux in the magnetic core that is approximately proportional to the amount of current I flowing through the primary winding. a current-voltage transducer comprising: the magnetic core having a discontinuously variable air gap; a secondary winding that outputs a voltage substantially proportional to the first derivative of the magnetic flux in the non-saturated magnetic core from an output terminal; A current-voltage transducer characterized by: 2. The transducer according to claim 1, wherein the primary winding is a single turn. 3. The transducer according to claim 1, wherein the magnetic reluctance of the air gap is changed by arranging a magnetic shim. 4. The transducer of claim 1, wherein the reluctance of the air gap is varied by adjusting a magnetic member protruding into the air gap. 5. The magnetic resistance of the air gap is changed by adjusting a portion of the magnetic core relative to another portion to change the air gap. transducer. 6. The transducer according to claim 1, wherein the magnetic core is formed of an annular body.