JPS62239887A - Control unit for ac motor - Google Patents

Abstract

PURPOSE:To prevent the fluctuation of terminal voltage and power factor caused by the load fluctuation, by providing a vector computing element to output the field current command and phase command from the armature current, power factor angle command and no-load terminal voltage command. CONSTITUTION:A phase differential angle operating table 201 finds a phase differential angle theta from the armature current Ia and the no-load terminal voltage VO, while a terminal voltage computing element 202 finds the terminal voltage V. Then, the field current command Ifp is found from a no-load saturation curve table 203, a d-axis component magnetizing current computing element 204, a q-axis armature reaction voltage computing element 205 and a field current computing element 206. The phase command gamma of a power converter is found by adding a commutation overlap angle signal U/2 from a commutation overlap angle computing element 208 and a power factor angle phi.

Description

DETAILED DESCRIPTION OF THE INVENTION [Field of Industrial Application] The present invention relates to a control device for an AC motor driven by a thyristor power converter.

[Conventional technology]

FIG. 7 is a configuration diagram showing an example of a thyristor motor for driving a conventional synchronous motor described in Japanese Patent Publication No. 59-1077.

In FIG. 7, 1 is a first converter that converts AC from a commercial AC power source into DC, 2 is a second converter that converts the DC into variable frequency AC, 3 is a synchronization machine a, F is the field winding; 4 is a position detector that outputs a position signal with a phase corresponding to the rotational angular position of the rotating shaft of the synchronous motor 3; Shift the phase according to the size of .

a γ control circuit that controls the control advance angle γ of the second converter 2;
Reference numeral 6 indicates a gate output circuit which outputs a Goo1 signal of the second converter 2 based on the output signal of the γ control circuit 5, 7 a speed generator, 8 a speed command circuit, and 9 a speed command signal of the speed command circuit 8. and a speed deviation amplifier that matches and amplifies the speed feedback signal which is the output signal of the speed generator 7;
11 is a current deviation amplifier that matches and amplifies the output signal of the speed deviation amplifier 9 and the current feedback signal of the current detector 10; 12 is the output of the current section difference amplifier 11; A gate pulse phaser that controls the firing phase of the first converter 1 based on the signal; 13 a field command circuit that outputs a command/IfP that commands the magnitude of the field current If; 14 a thyristor circuit 17 Current detection for detecting the magnitude of the AC input current; (:), 15 is the field command signal I
Current deviation amplification that matches and amplifies fp and the output signal of the current detector 14) t, 16 is a pulse phaser that controls the firing phase of the thyristor circuit 17, and 17 is a field current in the field winding F. This is a thyristor circuit that supplies If.

Next, to explain its operation, part numbers 7 to 12 change the hexagonal current of the first conversion 2 diagnosis 1 according to the speed deviation, that is, the magnitude of the armature '4 current of the motor 3, which is in a proportional relationship with this. Speed control circuit 1 to control Part numbers 4 to 6 are current detector 1
Circuit for controlling the control angle γ of the second converter 2 according to the output signal of 0, that is, the electric machine T-lightning current, part numbers 13 to 17
constitutes a field control circuit that causes the field current If to flow in proportion to the field command signal Ifp. Since these operations are similar to those of the already well-known so-called thyristor motor device, detailed explanation will be omitted.

FIG. 8 is a vector diagram showing the relationship between voltage and current of the motor in FIG. 7. The figure a shows the no-load condition, the figure shows the load condition when the field current If is kept constant and the control angle γ is controlled so that the power factor is constant, and the figure C shows the field current If separately. This is a vector diagram when the motor is controlled to be proportional to the armature current Ia and γ is kept constant.

As is clear from Fig. 8, even if the power factor can be maintained at a predetermined value, the terminal voltage V will increase as the armature current Ia increases (I
decrease (V. from al to Ia2).

From V2). This voltage drop causes the second converter 2
The maximum commutable current value decreases.

As a result, sufficient output cannot be obtained from the electric motor 3.

In addition, in the case of C in the same figure, the increase in armature current Ia (I a
, to I az), the terminal voltage V increases (from V, to V,), so there is no problem as shown in the figure.

However, at the time of overload, the terminal voltage (2) becomes higher than at the rated time, so the thyristor of the second converter 2 needs to have a high withstand voltage. Furthermore, because the electric motor itself undergoes magnetic saturation, it may not be possible to obtain as large an output as expected. Furthermore, as a result of the terminal voltage ■ decreasing at light loads,
This has the disadvantage that the power factor of the first converter 1 (1!! source power factor) decreases accordingly.

In addition, as a means of solving the above problems, the
Publication No. 77 describes how to control the magnitude of the no-load induced voltage E obtained by vectorial addition of the terminal voltage and the synchronous reactance drop, and the phase difference between this no-load induced voltage E and the armature current Ia. Accordingly, a method for controlling the terminal voltage to be constant regardless of the armature current is described in detail.

Although FIG. 9 is a vector diagram showing the principle of this operation, this operation will be briefly explained here. In order to keep the terminal voltage ■H constant, the no-load induced voltage E is adjusted according to the magnitude of the armature current Ia. The phase difference between the magnitude of E and the terminal voltage is 0 (
At the same time, the phase of the second converter (γ10) is controlled while maintaining the relationship of γ10 so that the phase difference γ between the armature current Ia and the terminal voltage is constant. There is.

However, in this method, since the terminal voltage is controlled to be constant, the commutation overlap angle U of the second converter changes depending on the magnitude of the armature current, and the arm element of the second converter changes. The application period (γ-U) of the reverse voltage to the thyristor changes.

At this time, when the second converter is made multi-phase (for example, 12 phases) to reduce torque pulsation and drive a large-capacity thyristor motor, commutation is performed every 30@, so that the other phase Due to the influence of commutation, as shown in Figure 10, the reverse voltage period of the thyristor, which is an arm element, is 30°-30° even when γ>30°.
In order to ensure stable commutation of the second converter, it is necessary to set the terminal voltage so that this conduction type angle does not become extremely large with respect to an increase in armature current.

In addition, in order to accurately control this voltage, it is necessary to consider the magnetic saturation characteristics of the AC motor 3.
The method disclosed in Japanese Patent No. 9-1077 has a problem in accuracy.

[Problem that the invention seeks to solve]

Conventional AC motor control devices are configured as described above, so the terminal voltage and power factor fluctuate significantly due to load fluctuations, causing the commutation of the second converter to become unstable or insufficient output. There were problems such as not being able to obtain

This invention was made to solve the above-mentioned problems, and it is an alternating current that can prevent fluctuations in terminal voltage and power factor due to load fluctuations, perform stable commutation, and obtain sufficient output. The purpose is to obtain a control device for an electric motor.

[Means for solving problems]

The AC motor control device according to the present invention has a phase difference of 0 (phase difference angle) between the terminal voltage and the no-load induced voltage according to the armature current.
It is equipped with a vector calculator that controls the field current and the magnitude of the terminal voltage so as to secure a predetermined commutation margin angle, and uses the terminal voltage signal of the AC motor as the phase signal of the AC motor. This is what I did.

[Effect]

The AC motor control device in this invention controls the locus of the terminal voltage by a vector calculator so that it is parallel to the axis (d-axis) of the field current, and the field current is a magnetizing current for generating the terminal voltage. The terminal voltage of the AC motor is controlled by the sum of the d-axis component of Let this be the phase signal of the AC motor.

〔Example〕

An embodiment of the present invention will be described below with reference to the drawings. 1st
In the figure, 18 is a power factor angle command circuit that commands the advance angle φ (power factor angle) of the armature current with respect to the terminal voltage of the motor 3;
Reference numeral 19 indicates a no-load terminal voltage command circuit for commanding the terminal voltage of the motor 3 during no-load, and 20 indicates a vector calculator, which is connected to the power factor angle command circuit 18 and the no-load terminal voltage command circuit 1.
9 and the armature current detection signal Ia are input, and the field current command Ifp and the phase command β of the second converter 2 are outputted. Reference numeral 21 denotes a phase control circuit, which controls the conduction phase angle of the second power converter 2 based on the output signal of the PT 22, which serves as a terminal voltage detector for detecting the terminal voltage of the motor 3, and the command of the vector calculator 20. do.

FIG. 2 shows a detailed configuration diagram of the vector calculator 20. As shown in FIG. Second
In the figure, 201 is defined by vo, Ia, and φ (No. 6 0 (
A θ function table 202 outputs the θ function table 201 (phase difference angle) and V.

203 is a no-load saturation curve table for the motor 3 that calculates the magnetizing current iμ from the output signal of this V-arithmetic circuit 202, and 204 is an output signal of this no-load saturation curve table 203 and iμ from O
The iμd arithmetic circuit 205 outputs q from Ia and φ.
An Eaq calculation circuit 206 calculates the shaft armature reaction voltage component Eaq, and 206 calculates a compensating field current component ifa of the armature reaction from the output signal of the Eaq calculation circuit 205.
a calculation circuit, 207 is an adder as a field current command generation circuit that adds the output signals of this ifa calculation circuit 206 and the 1 μd calculation circuit 204, and 208 is a U calculation that calculates the commutation overlap angle U using ■ and φ. A circuit 209 is an adder serving as a phase command generation circuit that adds the output signal - of the U calculation circuit 208 and φ.

Next, the principle of the operation of the above embodiment will be explained with reference to the vector diagram shown in FIG. As a reference axis, if the direction of the field current is the d-axis and the axis perpendicular to this is the q-axis, then q
A no-load induced voltage of the motor 3 is generated in the axial direction.

The basic control means in this invention is such that the vector locus of the terminal voltage V changes in parallel with the d-axis direction in response to the armature current Ia with respect to the no-load terminal voltage v0 on the q-axis. It's about controlling. If the phase difference (phase difference angle) between the terminal voltage V and the q-axis is θ, and the phase difference (power factor angle) between the armature current Ia and the terminal voltage V is φ, then the terminal voltage V is the no-load terminal voltage V.

and the armature reaction voltage component Ead=Xa generated in the d-axis direction
It is determined as a vector sum of q I acos (φ+0), and the following relationship holds true.

Votanθ: Xaq I a cos (φ+θ)
・=−(1) Transform equation (1) to obtain equation (2).

Here, the left side of equation (2) is the no-load terminal voltage V.

per unit (p
erunit) value. Using the θ function table, θ for a predetermined φ can be determined.

FIG. 4 is a graph showing an example of this O-function table.

The terminal voltage ■ is determined as a function of 0 using the following equation.

cos O (2) The calculation circuit 202 calculates the terminal voltage V according to equation (3). Next, the magnetizing current iμ generated in the direction orthogonal to this terminal voltage signal V is determined using the no-load saturation curve table 203. This no-load saturation curve table shows an example of curve 1 for the electric motor 3 as shown in a graph in FIG.
This shows the relationship between the induced voltage and field current at a predetermined speed, taking into account the magnetic saturation of 1aX, and this magnetizing current iμ corresponds to the composite magnetomotive force of the motor 3.

The d-axis component iμd of this magnetizing current iμ is calculated according to the following relational expression, and the iμd calculation circuit 204 executes the calculation of equation (4).

iμd=iμcO8O (4) On the other hand, the armature reaction voltage component Eaq in the q-axis direction is given by the following relational expression, and is calculated in the Eaq calculation circuit 205.

Eaq=XadIasin(φ+0)−(5) This q-axis armature reaction voltage component Eaq is controlled to be compensated by the field current component ifa in the d-axis direction. In this case, conversion from Eaq to ifa is performed by the ifa calculation circuit 206, using the tangential characteristic Kfa of the no-load saturation curve shown in FIG. 5 as a coefficient, as shown in the following equation.

1fa=Kfa−Eaq −・= (6) The d-axis field current components iμd and ifa obtained according to the above equations (4) and (6) are added by the adder 207, and the field is calculated as shown in the following equation. A magnetic current command IfP is obtained.

If p=i μd+if a −(7)
The ignition phase command γ of the second converter 2 is given by the sum of the power factor angle ψ and the commutation overlap angle U using the following relational expression with respect to the phase of the terminal voltage ■.

γ=φ10− (8) At this time, the firing phase angle β of the second converter 2 with respect to the q-axis direction is as follows.

β=0+φ1−・・・・・・(9) Here, the commutation overlap angle U is expressed by the following equation.

Note that equation (10) can be obtained by eliminating γ from cos(yu)-cosy:% (1) and equation (9). Furthermore, since the equation (10) is a function of the DC current Id of the second converter 2, it is necessary to convert this Id into the fundamental wave effective value Ia of the armature current.

If the commutation overlap angle U is considered, the armature current has a trapezoidal waveform as shown in FIG. 6, and the effective fundamental wave value Ia of the armature current at this time becomes a function of U as follows.

However, in large-capacity thyristor motors with 12 phases or more, the commutation overlap angle U is generally u <20''~25°
If the voltage is not limited, it will not be possible to secure the reverse voltage period for turning off the thyristor. At this point, in practical terms, I a-f-dan-
It can also be used as I d.

Therefore, if equation (1o) is transformed, it becomes as follows, and the U calculation circuit 208 executes the calculation according to equation (12).

As described above, since the device of the present invention is controlled according to the vector relational expressions (1) to (3), in order to ensure the thyristor commutation margin angle (reverse voltage application period) 30' -u, The above power factor angle φ and no-load terminal voltage V. It is only necessary to select an appropriate value.

The phase control circuit 21 only needs to perform a phase operation to advance by the phase command γ based on the phase signal of the terminal voltage of the motor 3 detected by the PT 22, and various types of this phase control method have been put into practical use. Since this is a well-known technique, its explanation will be omitted here.

In the above example, the constants Ka d, Ka q, and Kc mean dilQll armature reaction reactance, q# armature reaction reactance, and commutation reactance, respectively, and these constants are based on the frequency of the motor 3. Although this is omitted for convenience of explanation, it may be changed in proportion to the output signal of the speed generator 7. Similarly, no-load saturation curve table 2
03, when calculating the magnetizing current iμ, the input signal, the terminal voltage signal ■, may be converted into a signal inversely proportional to the speed of the motor 3 and provided.

Further, in the above embodiment, the detection signal of the armature current Ia is used as the input signal of the vector calculator 20, but the output signal of the speed deviation amplifier 9 may also be used. If the response characteristics of the current deviation amplifier are improved so that the deviation between the detection signal of the armature current 1a and the reference signal of the armature current which is the output signal of the speed deviation amplifier 9 becomes small, the same effect as in the above embodiment can be obtained. play.

Further, in the above embodiment, the calculations of the vector calculator 20 may be digitally processed by a microcomputer or the like, and in this case, the calculation accuracy is improved compared to analog processing. Furthermore, in the above embodiment, a six-phase rectifier circuit is shown as the second converter 2 in FIG.
Even if a plurality of second converters are arranged in parallel or in series to form a rectifier circuit with 12 or more phases, the same effects as in the above embodiment can be obtained.

〔Effect of the invention〕

As described above, according to the present invention, the vector locus of the terminal voltage changes in parallel with the d-axis direction with respect to the no-load terminal voltage by using a daple with a phase difference angle O. Since the vector calculation is performed and the no-load saturation curve is used to calculate the magnetizing current, the accuracy of the device can be improved and the device can perform stable commutation operation.

[Brief explanation of drawings]

Fig. 1 is a block diagram showing a control device for an AC motor according to an embodiment of the present invention, Fig. 2 is a detailed block diagram of the vector calculator in Fig. 1, and Fig. 3 is a diagram for explaining the operating principle of the present invention. Fig. 4 is a characteristic diagram of the θ calculation circuit, Fig. 5 is a characteristic diagram showing the no-load saturation curve, Fig. 6 is a waveform diagram of the armature current, Fig. 7 is a configuration diagram of the conventional device, Figure 8 is a vector diagram showing the relationship between motor voltage and current, and Figure 9 is a vector diagram showing the relationship between motor voltage and current.
A vector diagram for explaining the operation of the device shown in the figure, and FIG. 10 is a voltage waveform diagram of a thyristor. 1 is a first power converter, 2 is a second power converter, 3 is an AC motor (synchronous motor), 4 is a position detector, 18 is a power factor angle command circuit, 19 is a no-load terminal voltage command circuit, 20 is a vector calculator, 22 is a terminal voltage detector (PT), 201
is a phase difference angle calculation table, 202 is a terminal voltage calculation unit, 20
3 is a no-load saturation curve table, 204 is a d-axis component magnetizing current calculator, 205 is a qillll armature reaction voltage calculator, 206 is a field 6J1 current calculator, 207 is a field current command generation circuit (adder), 208 is the commutation overlap angle calculator,
209 is a phase command generation circuit (adder). In addition, in the figures, the same reference numerals indicate the same or equivalent parts. Figure 1 22] High 5th grade! 5 station area"; 4: Person 1L Okihashi 3 gloss "Finish - 2f1 =, (a) Figure 9 Figure 10

Claims (1)

[Claims] a power converter that performs frequency exchange of alternating current; an alternating current motor driven by the output of the power converter; a terminal voltage detector that detects the phase of a terminal voltage of the alternating current motor; and a no-load terminal voltage of the alternating current motor. a no-load terminal voltage command circuit that sets the magnitude of the power factor angle of the AC motor; a power factor angle command circuit that commands the power factor angle of the AC motor; a vector calculator that outputs a field current command of the AC motor and a phase command of the power converter according to the magnitude of the armature current; In order to perform a vector operation such that the vector locus of the terminal voltage of the AC motor changes in a direction perpendicular to the no-load terminal voltage, the phase difference angle is calculated by inputting the per unit value of the d-axis armature reaction voltage. a calculation table; a terminal voltage calculator for calculating the terminal voltage from the phase difference angle and the no-load terminal voltage signal; a no-load saturation curve table for the AC motor that calculates the magnetizing current from the terminal voltage signal; a d-axis component magnetizing current calculator that calculates the d-axis component of the magnetizing current; a q-axis armature reaction voltage calculator that calculates the q-axis armature reaction voltage from the phase difference angle, power factor angle, and armature current; A field current calculator for armature reaction compensation that calculates a field current component that compensates and cancels the armature reaction voltage component, and a field current calculator that calculates the field current by adding the armature reaction compensation field current signal and the d-axis component magnetizing current. a field current command generation circuit that generates a current command, a commutation overlap angle calculator that calculates a commutation overlap angle from the terminal voltage signal, armature current signal, and power factor angle, and a commutation overlap angle signal and a power factor angle. 1. A control device for an AC motor, comprising: a phase command generation circuit that generates a phase command for the power converter by adding .