JPS62136128A - Signal generator - Google Patents

Abstract

(57) [Summary] This bulletin contains application data before electronic filing, so abstract data is not recorded.

Description

DETAILED DESCRIPTION OF THE INVENTION Field of Application The present invention relates to signal generators, particularly those which can operate in the GHz range in either continuous wave (CW) mode or swept frequency mode, typically in transmission/reflection analysis. The present invention relates to stabilized signal generators used in equipment.

[Prior art]

Complementary source oscillators are available to generate very high frequency electrical signals, but all suffer from drift induced by ambient changes such as temperature, power supply fluctuations, and changes in the electrical components associated with the vibration source. receive more or less. Typical source oscillators utilized in very high frequency applications utilize YIG oscillators or varactor-tuned oscillators.

Techniques for renumbering frequency stabilization can be divided into the following two categories: (a) Techniques for non-fixed sources; (b) Techniques for fixed (usually phase-locked) sources. For non-fixed systems, an open-loop type of correction is usually combined with some type of linearization circuit. This type of method is
Varactor tuning circuits vary from simple diode shaping circuits common to varactor tuned circuits to more complex FROM data corrections incorporated into microprocessor control signal generators.

Since the circuit is open loop, no correction is possible for frequency shifts or misalignment errors with respect to time.

To help correct the source frequency for temperature drift, a temperature sensing thermistor may be used to generate a correction signal. However, in all other respects the device is essentially open loop.

For fixed systems, the signal is derived from a high frequency source using a reduced conversion principle to obtain a signal with a lower frequency than the source, and this reduced conversion signal is typically combined with a reference signal derived from a crystal controlled oscillator. be compared. The comparison is made with a phase detector, the output of which is a voltage proportional to the phase error of the two signals. A correction voltage is applied to the tuned portion of the high frequency source oscillator to correct for frequency shifts, and this nature of the system allows a significantly higher level of accuracy to be achieved.

An open-loop system typically uses 200
Although the accuracy lies within +/-5 MHz for signals close to 0 M)iz, a closed-loop phase-locked system can achieve accuracy better than 1 KHz at 2000 MH2.

As expected, closed loop phase identification systems are complex and more expensive than relatively simpler open loop non-fixed systems.

Both image-type sources may be configured to provide a CW ratio output or a swept output.

For open-loop sources, frequency sweeping is accomplished by adjusting the frequency-determining component to the starting frequency and then applying an increasing (or decreasing) tuning signal (usually a variable voltage) to the tuning port of the vibrating device. . This can be achieved by generating a voltage ramp, which can also be derived using analog circuitry (or derived from a digital signal using a digital-to-analog converter (DAC)). If the latter approach is used, conventionally a microprocessor is employed to control the generation of the ramp voltage. Whatever form of ramp signal is used (analog or digital) However, the oscillator has an inaccuracy that causes the frequency to vary, due in large part to nonlinearities in the tuned portion of the oscillator and temperature changes.

For phase-locked systems, two approaches are possible.

In one arrangement, the starting frequency at the beginning of each sweep is phase locked using a phase detector to relate the required output frequency to the reference frequency.

The source is tilted in an open loop manner and the sweep range is achieved as in a normal open loop.

However, the accuracy is greater because the frequency at the beginning of each sweep is phase locked.

Instead, the starting frequency is not only phase-locked, but the frequency is digitally raised (or lowered) in a series of steps, phase-locked to each frequency step, and then simply moved to the next frequency step after phase locking is achieved. A fully synthesized sweep is created by simply moving the

Of these two schemes, the first provides fast sweep update times but relatively poor frequency M degrees over the swept fluorescence range, while the second provides fairly high frequency M degrees during each sweep. level of frequency a accuracy, but the sweep update time is slow.

With any phase-locked version, frequency drift due to time, temperature or output mismatch will be corrected instantaneously, or at least before each sweep. Such a correction is not possible within an open loop swept frequency generator.

Because of the considerable expense of phase-locked systems, various attempts have been made to improve the accuracy of open-loop systems. One approach is the so-called [B I RD I
The purpose was to generate a marker.

It provides a signal generator and associated visible bottom water equipment,
This display device was designed to display electrically generated marker signals.

Systems of this kind operate by generating from a crystal control signal a so-called comb train (harmonic train) of frequencies, each of which is a multiple of the frequency of the original crystal control signal. The frequency generator output signal is sampled and the sample signal is
comb-row” signal. A beat, or rBIRDIJ, is generated between components of the rw-like sequence of frequencies that are closest to the frequency of the output signal. Using a low pass filter, the beat (or rBIRDIJ) signal can be separated from the high frequency signals and amplified and displayed on a CRT or similar display device. Frequency correction is accomplished by manually adjusting the mask oscillator to correct for any frequency drift. The frequency drift can be seen in the rBIRDIJ display on the screen.

Systems utilizing such red water devices have a closed-loose configuration (when the user is taken into account), but the accuracy achieved is much lower than phase-locked loop systems, and this type of system Apart from the inherent time delay, they have other disadvantages with respect to conventional open loop systems. (a) As the amplitude of the output signal decreases, the beat signal also decreases in amplitude and may ultimately be lost in the electrical noise within the system. Visual correction at this stage becomes very difficult.

(b) If a very frequency-selective component with a very steep response curve is tested, the beat signal may be lost in the steep component characteristics and frequency correction will also be very difficult.

On the other hand, generation of a beat signal from a crystal-controlled source represents a simple method of generating a feedback signal, and it is therefore an object of the present invention to provide a beat signal that is simpler and cheaper than conventional phase-locked loop systems, and that The objective is to provide a closed loop system that derives feedback information for control from the beat signal.

[Summary of the invention]

In accordance with the broadest aspect of the invention, it provides an output frequency of FMHz in a p-concave circle ranging from a few hundred to a few MHz, with coarse and fine frequency control elements, and with fine control, N In a method of configuring a signal generator capable of producing a direct expansion change in the output frequency for a corresponding change in the parameters of the electrical signal supplied to the element over only a small range of MHz, the output of the device 6: NM) f of the desired frequency FMHz
combining a multi-component reference signal comprising at least a component with a frequency of LMHz within FM I-1z;
adjusting the fine frequency control element until a beat signal is detected (at which point the beat signal is equal to or greater than the output signal and the reference signal (or one thereof); The device can be calibrated by storing the value of the frequency control signal at which the beat signal is detected, and using an algorithm or lookup table to determine the frequency of the device (F -L) determining the value of the correction signal required by the fine frequency control element to shift by MHz, and adding the correction signal to the stored signal and using the combined signal as a control signal for the device; A signal generator setting method is provided which is characterized in that a desired output signal of FMHz is thereby obtained.

Typically, calibration procedures, as well as frequency shifting, zero detection, signal storage and addition, are performed under microprocessor control.

According to another aspect of the invention, the output frequency is frequency F
a signal generator having an output configured to generate a radio frequency signal when operating at t, and having, in addition to a large frequency control element, a small frequency control element;
A main oscillator whose oscillation frequency can be varied over a small frequency band by linearly varying the parameters of the electrical signal supplied to it; a controller configured to generate an electrical signal; 5. means for deriving a sample signal from the output of the master oscillator; 4. a stabilized fixed tube wavenumber oscillation signal generator for generating a reference signal F (ref); , from the reference signal p (ref), the harmonic spectrum of the signal having at least one component within the small frequency band F (ref); 2 F (ref)
; 3 F (ref)...n F (ref)
6. a circuit for generating a harmonic spectral signal (referred to as a harmonic spectral signal); a mixing or sampling circuit that produces a signal having a frequency that is a difference; and a beat signal detection circuit that responds to the output of the Z mixing circuit and identifies when a beat signal pulse is occurring; For this purpose, the frequency-determining signal supplied to the main oscillator is changed until a beat signal is detected (caused by the interaction of the main oscillator output signal and one component N, F (ref) of the high frequency spectral signal). The control circuit is programmed at , and at this point the value of the frequency determining signal supplied to the master oscillator is (
N, F (ref) +
and the required frequency shift between the desired frequency F).

The associated beat signal can be selected from a number of sum and difference signals by frequency selective separation using a low pass filter.

Typically, the crystal controlled oscillator is set to operate at a frequency of approximately 25 MHz, with the master oscillator operating at a frequency in the range up to 1 MHz.

According to another aspect of the invention, a method for controlling the oscillation frequency of a main oscillator in a signal generator in which a linear frequency axis/embedded element has limited frequency sweeping capability comprises: 2. combining the samples of the main oscillator output signal into a base signal having one component within said limited sweep frequency of the main excavator; 5. the main oscillator output signal. Adjust the value of the frequency control signal until a beat signal between the and reference signal components is detected, and then change the value of the frequency control signal from the value corresponding to the calibration to A starting FS frequency control method comprises steps of adjusting the master oscillator's frequency response to a new value calculated using an algorithm or look-up table, thereby shifting the master oscillator output to a desired frequency. provided.

After the calibration procedure has been performed, the master oscillator frequency will become correct and remain so thereafter depending on the quality of the open-loop correction incorporated into the signal generator to compensate for thermal drift, etc. Let's find out.

When the master oscillator is required to sweep over a frequency range, the method of the invention preferably includes the following steps. (1) Using the corrected control signal as a starting point, vary the calculated control signal according to an algorithm that combines the oscillation frequency with the value of the control signal, and (2) reach the final target frequency of the sweep. (3) then repeating the correction process, including the method described above, until a beat signal is detected, after which the sweep process can be started again; including.

The basic signal may be a so-called high-frequency spectrum signal, typically interrogated from a crystal source by a step-off diode.

It will be appreciated that for this type of correction, the linearity of the swept frequency is determined, among other things, by the accuracy of the algorithm that couples the oscillator frequency to the control signal.

If the algorithm is an incomplete description of the frequency versus control signal relationship, or if the particular master oscillator is relatively linear and the algorithm is a theoretical description of the performance of a more linear version of this type of device, then The resulting sweep will not be particularly linear.

Unless sweep time is unacceptable, a preferred method according to the invention includes the following steps.

1. Tracing the range to be swept to identify a series of key frequencies, starting at the lowest or highest and increasing the frequency until a beat signal is generated by the interaction of the components of the master oscillator output signal and the reference signal. 2. correcting the control signal supplied to the oscillator; 2. correcting the corrected control signal according to a known algorithm for generating a control signal to obtain a frequency between the first frequency and a next target frequency; 3. Repeat the calibration process for the oscillator control signal as it approaches the next target frequency, then change the oscillator frequency to the next frequency? io and the steps of continuing to change to the next target frequency through the range.

In this way, the frequency of the master oscillator is corrected in each successive step throughout the entire range, and if high precision is required a correspondingly large number of steps are employed.

Typically, a digital signal is generated by a microprocessor and a digital-to-analog converter (DAC)
It functions to convert digital signals into voltage or current,
This can be used to control the frequency of the main oscillator.

It is convenient to select an appropriate beat signal from DC to
This is achieved using a low-pass filter set that transmits signals in the range f (where f is e.g. 5ooKH1 to 5
(may be in the MHz range).

Considering the attenuation effect of the low-pass filter, it is preferable to provide one signal amplification and pulse conditioning circuit after the low-pass filter and before the microprocessor.

Typically, the high precision fixed cylinder wavenumber low source is a crystal controlled oscillator.

A high level of accuracy is obtained by determining the precise center position of a selected beat signal pulse from the width of the beat signal pulse. This central location 14 thus corresponds to the zero frequency or DC content within the beat signal. Therefore, the microprocessor is preferably programmed to search for the center position of the beat signal pulse detected thereby and to use the determined center phase of the pulse as a target.

If the stored information is in digital form and the changes to this information are in the form of a digital error signal generated by a microprocessor, then both the stored information and the error signal can be converted to analog signals, and if not. Make it easier to add them than you would otherwise.

In sweep mode, i.e. when the generator generates a signal whose frequency varies with time from a first frequency (fl) to a second frequency (F2) defining a sweep band, two things of information are true. Conveniently, it is incorporated into a microprocessor and associated memory.

t sweep width (i.e. F2-Fl) 2, i sweep range (7) center (suf', cb chi, 1 (
p 2-4- p i )) The process of finding the starting point of the sweep range includes the following steps.

1. A digital-to-analog converter initialized by a microprocessor calculates control signals to produce one or more analog signals that should theoretically cause the master oscillator to generate a starting frequency.5.
Program the microprocessor from the sweep width and center information.

2. At the same time, set the error DAC (the output of which is combined with the first-mentioned DAC means) to the mid-range.

5. Until a beat or difference signal is detected, the error DAC
The error signal is corrected by the microprocessor so as to change the output from the microprocessor.

4. Storing the aggregate correction value for the value of the error signal required to produce the beat signal pulse in a buffer memory for later reference.

5. Using the value stored in the buffer as a starting point, according to the frequency/control signal characteristics of the main excavator,
An adjustment is made to the control signal provided to the master oscillator to obtain a new operating frequency of the master oscillator that corresponds to the start frequency of the sweep.

6 The microprocessor performs the sweep and produces a varying output signal. This adds fl to the main oscillator output frequency.
to the frequency control element of the main oscillator to produce the desired sweep from F2 to F2.

At the end of each sweep, the microprocessor should return to the search routine to recalibrate the oscillator (look for the beat signal so that the oscillator is corrected at the start of each sweep).

Preferably, the master oscillator comprises two frequency control elements, a first for changing the frequency by a large amount and a second for changing the frequency by a smaller amount, and a control such as a microprocessor driven controller. The apparatus is configured to generate a tenth control signal for the input of the first control element and a second control signal for the input of the second control element. If an error signal is generated by the control device, this signal will be added to or subtracted from the value of the second control signal.

For high frequency applications, the main oscillator is preferably a YIG oscillator with a main tuning coil for wide range tuning and a small FM coil for fine range tuning.

Alternatively, a varactor-tuned oscillator can also be employed.

[Explanation of Examples]

The present invention will be described below with reference to the drawings, with reference to preferred embodiments.

FIG. 1 is not intended to be illustrative of an embodiment of the invention (it shows how temperature correction may be applied to an oscillator; corrections of this nature may be included in embodiments of the invention).

In FIG. 1, a YIG oscillator 10 has a coarse tuning boat 12 associated with a main tuning coil and a fine tuning boat 14 associated with an FM coil. Signal amplifiers 16 and 18
act to provide control signals to boats 12 and 14, respectively. A coarse frequency adjustment, usually in the form of a potentiometer, is not shown, but a fine frequency drive input is shown as being derived from amplifier 20. The input of the amplifier thus includes a temperature compensation resistor or thermistor 22. By positioning a temperature compensation resistor or thermistor to sense changes in the temperature of oscillator 10, a compensated error signal can be provided to input 14 via amplifiers 20 and 18 to correct for temperature drift.

The output frequency on lfM24 is better stabilized than without temperature compensation.

FIG. 2 illustrates a more crude closed loop phase-locked excavator. Like reference numerals are used to designate parts common to FIG.

The main difference between the two circuits is that a closed loop is formed between the output 24 and the input of the 26 fine frequency drive amplifiers 18. This loop is closed by a programmable divider 28 that supplies the low frequency signal to a phase detector 30. Thus, the phase detector 50 is supplied with a reference signal 32 from the Listal controlled oscillator.

If there is a small change in the IN number between a frequency that is a submultiple of the output frequency on line 24 and the crystal control frequency on input 32,
This appears as a phase difference, and the error signal generated by the phase detector is passed through the amplifier 18 to the fine tuning port 14.
supplied to

Very high levels of accuracy can be obtained using this type of circuit.

Since this circuit is complex and expensive, especially at very high frequencies, there is an opportunity for an alternative approach to the present invention.

FIG. 3 is a block circuit diagram of a specific example of the present invention. As previously mentioned, common parts are designated by the reference numerals already used in FIGS. 1 and 2.

The present invention is primarily, but not exclusively, directed to ultra-high frequency signal generators.
It is particularly directed to signal generators generating signals in the range I MHz to 2000 MHz.

According to the invention, digital signals from a microprocessor (not shown) are provided to the amplifier 16 via a DAC 34 and to the input of the amplifier 18 via the DAC 36. DAC 38 provides an error signal that is combined with the output of DAC 36.

The sampled signal from line 24 is provided as an input to a mixer or sampler 40. The other input of the mixer is a crystal controlled oscillator 4 normally operating at 25MHz.
This is a signal derived from 2.

The output of this oscillator is then amplified by a power amplifier 44 and then fed to a step-off diode 46.
. . . produces a so-called "collapse train" signal or a harmonic spectrum signal. In this signal train, the lowest frequency is 25M
Hz, the maximum being determined by, among other things, the bandwidth of the circuit.

Converting a fixed tube wavenumber signal (25 MHz O) into a harmonic spectrum signal reliably generates a large number of so-called rBIRDIj markers, to which the sample signal can be compared.

The comparison is performed by a mixing technique that provides sum and difference signals at the output of mixer device 40. Low pass filter 48
is a low-pass filter that rejects all but the most useful beat signal pulses and thus has a cutoff frequency of IMHz. The attenuated output is amplified by amplifier 5° and provided to the microprocessor via pulse conditioning circuit 52.

4a and 4 represent the nature of the output of the pulses at the output of the regulating circuit 52. FIG.

In FIG. 4a, the output pulses from the low-pass filter are shown on a small scale. This means that the oscillation frequency is 2
each passing harmonic component of the 5 MHz reference signal is generated, and each pulse itself consists of a number of pulses (the 4th b
(see figure), the repetition frequency of which varies between a relatively high level near the edges of the pulse and a DC level (in theory) in the middle of the pulse. The edge regions are designated by reference numerals 54 and 56 in FIGS. 4a and 4b, and the central DC band is designated by reference numeral 58.

Pulses such as 60 shown in Figure 4a are relatively narrow;
To a first approximation, the leading 6+ (or trailing edge) of each said pulse is detectable and can be used to indicate the arrival of a beat signal.

In accordance with a preferred aspect of the invention, each pulse 60
An apparatus is included for determining the midpoint of the nollus, such that the true DC or zero frequency band is used to determine the actual presence of the norus.

Figure 5 shows the system shown in Figure 3? Is there a microprocessor control device that is not shown earlier that illustrates the developed system? Contains common forms.

Where parts common to previous figures are used, the same reference numerals have been adopted. Thus, YIG oscillator 10 provides a small RF power along line 24 and elements 40, 42 .
For convenience, the contents of 44.46.48.50 and 52 are as follows:
It is shown in FIG. 5 as a single element designated by reference numeral 62. One output from this element constitutes an RF output socket) 64, and the other output constitutes a so-called marker output. The beat signal pulse or marker-containing electrical pulse at this output is the output from the YIG oscillator and the local crystal controlled oscillator 4 of FIG.
is generated by interaction with the high=x wave derived from the output from 2.

The individual pulses are called markers, and in the inset in FIG. 5 such a marker of vI number and an enlarged view of the first marker are shown. From the enlarged view, the marker consists of a plurality of individual pulses of variable amplitude and frequency, reference number 68.
It has essentially four frequencies, namely the central region of DC, as indicated by .

Depending on the manner in which pulses such as 66 are generated, the presence of a DC level within pulse 66 dictates a particular relationship between the output of the YrG oscillator and the output of crystal controlled oscillator 42, and also the value of the control signal. can also be instructed. This control signal is initially supplied to the YIG oscillator.

Determination of the presence of a zero frequency band or DC level within a pulse such as 66 is most simply accomplished using a microprocessor controlled pulse analysis circuit, generally directed at 70, and included within block 62. A marker pulse interface device 72 (a portion of which is incorporated within the pulse conditioning circuit 52 of FIG. 3) is required between the output from the mixer and low-pass filter of the circuit to be processed and the input of the microprocessor controller 70. It is said that

Analysis circuitry determines the width and midpoint of the first marker pulse 66. This point corresponds to the 1=1 relationship between the YIG oscillator frequency and the frequency of the crystal controlled oscillator 42 in Figure @3. The microprocessor control unit is 7
Keyboards or other data input devices such as 4 are provided, which may serve to include dedicated control pads N such as start/stop and center 7 wide keys.

Although the signal generator of FIG.

The initial preset of the sweep width (scaling) and center is determined by the required start and stop frequencies at which the source sweeps. Thus, the YIG oscillator, as described with respect to FIG. It must be driven between the start and stop frequencies by controlling tl. However, to allow greater control over the sweep settings, three digital-to-analog converters are provided instead of the two elements described with respect to FIG. The first DAC 76 is
Center of sweep? and the second DAC is the scaling DA
C80′f generates a ramp voltage that must be supplied to one or the other of the drive amplifier 16 and the third
A DAC 82 converts the error signal to the input of a second drive amplifier for controlling drift and controlling source tuning.

The selector switch 84 allows the output from the scaling DAC 80 to be supplied as an input to the amplifier 16t1:18. enable.

A common source 86 connects three DACs 76, 78 and 8
A data bus from microprocessor drive controller 70 is provided as a main data input to each of the DACs.

Data highway 88, information about the tuning of the YIG oscillator 10 and sweep parameters such as start and stop frequencies, as well as DAC 80, sweep DAC 78
and provides sweep width information for the center DAC 76.

During each sweep, the sweep DAC is the only converter that needs to be dynamically changed. Information supplied via the data highway 8 and 92 minutes is provided by a sweep DAC 78'5
- generates a fixed slope signal, the actual shape and amplitude of which is determined by parameters within the microprocessor. The actual amplitude of the ramp signal is assigned by converter 80, also using information from the microprocessor, to cover the appropriate fr sweep width determined by information provided via data entry keyboard 74.

The function of the error DAC is the frequency that can be generated by the YIG oscillator 10? , maximum +/- under microprocessor control
The frequency is varied by 25 MHz. This is YIG
It is used to shift the kneading until the frequency of the signal generated by the oscillator circuit 10 is one integer multiple of the comb signal from the circuit element 46 + 7 harmonic components.

The response for the error DAC is that the frequency of the Y/G oscillator is
It must be calibrated, stable, and fast so that it can be easily changed accurately anywhere within the 50 MHz range dominated by the variation device. In this way, as long as the beat signal can be located at some point within the 50 MHz band, the YIG oscillator will be at the desired final frequency +/-2
Can be calibrated to frequencies within 5 MHz and subsequently shifted to the desired frequency by adding an additional error signal via the DAC 82 to the frequency control signal already provided to the oscillator during calibration Make it.

Thus, the variable action of 1 microprocessor and error DAC 82 serves two purposes. Namely: 1. During the calibration of the YIG oscillator at the beginning of each sweep, the variation device:
The frequency of the YIG oscillator is quickly and precisely shifted within a 50 MHz bandwidth, allowing at least one (preferably two) beat signals (commonly referred to as markers) to be found.

2. After calibration, the same two elements (i.e., microcassette 70 and error DAC 82) are required to shift the frequency of the YIG oscillator to the desired frequency required by the input information provided to the microprocessor.
It functions to generate a C shift signal.

The process for locating beat signals or marker pulses such as 66 is as follows.

From the selection of t4 wavenumber span, center, sweep width and sweep I
) Initialize all ACs and adjust error DAC 82 to midrange under microprocessor 700 control.

2. Invoke a search routine by the microprocessor when attempting to begin a sweep. In this case, teJ
The difference D A C 82 is first adjusted in one direction (usually downward) when the marker pulse 66 is detected by the marker inter 7 ace 72 .

Typically, this process is set to look for changes in amplitude that involve conversion from high frequency content to DC and back to high frequency content in related amplitude variations. The position of the zero point DC level within the detected pulse 66 is confirmed,
The error signal or variation value required to produce a particular beat signal is then stored in the microprocessor's memory.゛5 F of YIG oscillator to generate marker pulse
After setting up the required signals for the M-fill, the oscillator is enabled. Now, in order to obtain the desired frequency, it is up to the microprocessor to calculate the additional signal that should be added (or subtracted) to the known signal to obtain the same frequency component as the identified comb signal component. It's simple.

If the desired frequency happens to be the frequency of the marker (the a-column signal component), no additional addition or 't2 calculation is required.

In the many common cases where the desired star 1 frequency is not equal to one of the comb signal components, an algorithm or lookup is required to determine the signal correction needed to obtain the desired frequency from the calibrated frequency. table is adopted.

The algorithm or lookup table may be stored in microprocessor memory (or in ROM).
(or entered into machine memory via the keyboard) and describes the frequency response of the YIG oscillator 10 for the different signals provided to the main and side frequency determining coils.

If very high accuracy is required, the microprocessor search routine may take the steps required to identify the high frequency edge regions of marker pulse 66 identified at 94 and 95 in FIG. and the microprocessor is programmed to interpolate between the edges of the marker pulse to find the center position between the two edges 94 and 96. Since the pulse 66 occurs when the YIG oscillator frequency moves from one frequency above the comb signal frequency to the bottom (or vice versa), the center point 68 is the YIG oscillator frequency.
This corresponds to when the frequency of the G oscillator 10 is exactly equal to the comb signal frequency. The frequency of oscillator 10 is mixed with this frequency. By incorporating this additional calculation process into the microprocessor search program, unique and precise signal parameters can be determined by the processor 70 for the output from the error DAC 82, and from the YIG oscillator 10 to the crystal controlled oscillator 42. It is possible to generate a signal having exactly the same frequency as the frequency of the comb-like signal component derived from the comb-like signal component.

Current open loop signal generators provide frequency accuracy of approximately +/-5 MHz. The device embodying the present invention has 20
+/-100KHz accuracy at 00MHz? Achieved. Does YIG oscillator 10 have good linearity? and has a predictable frequency response to changes in the control signals from amplifiers 16 and 18? If so, the system shown in Figure 5? can be used to perform very precise sweeps.

If CW operation is required, the sweep width is set to zero and the operating frequency of YIG oscillator 10 is determined by center DAC 76. Set the error DAC82 to the center range,
By invoking a microprocessor search routine, YIG oscillator 10 finds the closest marker pulse, such as 66 (and its central location if that level of accuracy is required), and determines the actual frequency of YIG oscillator 10? What error or variation signal is needed to shift the source 10 to the desired dark wave number to be generated? Calibrated by calculation.

During CW operation, particularly thermally induced drifts can be controlled by the thermal compensation circuit described with respect to FIG. It may be programmed to perform a breech search and calibration routine during each interruption before turning the RFE back on.

Although the YIG oscillator 10 is preferably not turned on and off to reduce transient and switching problems, the RF circuit 62 is used in this mode to attenuate the output mid-RP signal during initial calibration and subsequent calibrations. It is also preferable to combine with.

FIG. 6 shows an example of hardware in which the microprocessors 2 are not combined. However, this embodiment is limited in accuracy to simply detecting the presence of marker pulses such as 66.

Here! - The Cainter 7 Ace is equipped with a pulse presence detector 114 which detects either the leading or trailing edge of the pulse, indicated at 66, by any convenient means.

The frequency corresponding to the required sweep is on the front panel 9.
8 and the frequency command circuit 100 calculates the appropriate drive signal and sweep signal for the drive DAC 76.
Set the analog voltage for the digital data or drive amplifier 16.18. The circuit also generates a voltage corresponding to the distance the required starting frequency is from the calibration point. This is called a fluctuation signal and can be turned on and off with switch 81. Thus, if the required starting frequency is 57 MHz and the calibration point is obtained at 50 MHz, a signal corresponding to -)-7 MHz is generated.

If 42MHz is the required starting point, -8M
) Is must be generated. A NULL signal (0 volts) is generated for a start frequency equal to the calibration point.

This also applies if the starting frequency is a multiple of 25 MHz.

The frequency command circuit 100 sets the drive circuit so that the start frequency is a multiple of the calibration point, inputs a fluctuation signal after calibration, and obtains the required start frequency by adding this to the signal set by the signal 9100. .

Sweep ramp generator 102 generates a fixed ramp signal that drives oscillator 10 through drive electronics 78,80 over the required range. The generator initiates a ramp signal in response to a signal at a trigger input. When the slope signal ends, generator 1
02 indicates this by providing a signal to the end of sweep output 10'4.

The other part of the hardware, namely the controller 108,
The ramp signal starts from its minimum point as required by a zero-out ramp start human power 110 signal that provides a second ramp signal over a small frequency range sufficient to cover the maximum expected error; Stops when commanded by lock ramp stop human power 112.

The RF output is provided to marker presence detector 114 (already mentioned) and its output is latched by latch circuit 116. Controller 108 is operative such that in the presence of marker pulse 66 (corresponding to a calibration point) a lock ramp stop signal is generated on input 112 and the ramp value from 108 is held in sample and hold circuit 118. do.

The order of events that occur during a typical sweep under the control of signals from timing circuit 120 is as follows.

Frequency parameters are entered via a control number on the front panel 98. The frequency command circuit 100 sets the frequency to a desired stop frequency.

The fluctuating signal must be rejected, while the YIG oscillator 10
It will be appreciated that the is calibrated at the beginning and end of each sweep. For this reason, switch S1 is provided.

If greater accuracy is required, detector 114 may interpolate from the width of pulse 66 as described above with respect to FIG.
Because it generates information corresponding to the zero point 68 of the detector, it can be a more complex device.

FIG. 7 shows a double reduction conversion system based on FIG. 3, which allows software fixing to reach 30 GHz and above by appropriate selection of the two drive frequencies.

The design and operation of the system follows similar lines to FIG. 3, and the same reference numerals are adopted where appropriate.

The RF output 1i-wave samples are passed to a first reduction conversion stage that includes a mixer or sampling device 126.

The mixer includes a high frequency reference source 128 (e.g. 500'
MHz SAW oscillator), a power amplifier 130 and a comb-column signal for driving through a step-off diode 132. The low frequency reduced conversion signal is 134.13
6 provides a broadband drive signal to a second reduction conversion stage consisting of an amplification, P-wave, mixer or sampling device 138 (corresponding to the mixer or sampling device of FIG. 3). This mixer 138 is driven with a drive signal that has a much lower frequency than the drive signal for mixer 126. The drive reference signal for mixer 138 is divided into a first reference signal (i.e., 500
MHz SAW source).

The rest of the circuit is the same as described for No. 311! Works the same way.

During the initial tuning of the output frequency, both low and high frequency pulses are processed to accommodate the increased non-linearity and pointing inaccuracy that would occur at such high frequencies.

For this purpose, a second process i4B' is applied to the output of 136
, 50' and 52' are provided.

In use, the YIG oscillator 10 is approximately tuned and the most closely related 500 MHz marker is searched. Once this is found, it can be compared to the associated 25MHz marker on the low frequency marker output. Then, for that output frequency range, only a comparison against the 25 MHz marker is required. 500 MH only if the output frequency is to be shifted significantly (e.g. 3 GHz)
A comparison to the z marker output would be required.

[Brief explanation of drawings]

1 is a schematic diagram of an open-loop oscillator with temperature compensation; FIG. 2 is a schematic diagram of a closed-loop oscillator or synthesizer; and FIG. A block diagram of a crystal control beat signal (marker), FIGS. 4a and 4b are diagrams showing typical marker pulse configurations, and FIG. 5 shows a drive circuit and a beat signal (marker) fixed loop embodying the present invention. FIG. 6 is a block diagram of an RF source with an R
4 is a block diagram illustrating a double reduction conversion system based on the circuit of FIG. 3, where the F range can reach as high as 30 GHz; FIG. 10: YIG oscillator 12: Coarse tuning 14: Fine tuning board 16. 18: (Signal) amplifier 20; Amplifier 228 Temperature compensation resistor or thermistor 28 Niprogrammable amplifier 30: Phase detector 34. 36. 58: DAC (Digital-analog humperter) 40: Mixer or sampling device (Stake sampler) 42: Crystal controlled oscillator 44: Power increase @A unit 46: Step-off 7 diode 48: Low-pass filter 50: Amplifier 52: Adjustment circuit

Claims (6)

[Claims] (1) FM within the range of up to several hundred to several thousand MHz
Provides an output frequency of FMHz in Hz and is equipped with coarse and fine frequency control elements, with fine frequency control, over only a small range of NMHz, the output frequency roughly changes over a corresponding change in the parameters of the electrical signal supplied to the element. In a method of configuring a signal generating device capable of producing a linear change in the output of the device, the output of the device is combined with a multi-component reference signal comprising at least a component having a frequency of LMHz within NMHz of a desired frequency of FMHz and supplied to the device. the frequency control signal to be detected is adjusted to produce an output signal having a frequency that is also within NMHz of FMHz, the fine frequency control element is adjusted until a beat signal is detected, and at this point the beat signal is detected. indicates a one-to-one relationship between the output signal and the reference signal (or one component thereof) so that the device can be calibrated, the value of the frequency control signal at which the beat signal is detected is memorized, and an algorithm or look-up table can be used to The value of the correction signal required by the fine frequency control element to shift the frequency of the device by (F-L) MHz is determined, and the desired output signal at FMHz is the signal in which the correction signal is stored. A method for configuring a signal generator, characterized in that it is obtained by, in addition to: using the combined signal as a control signal for the device. 2. The method of claim 1, wherein the calibration procedure and frequency shifting, zero detection, signal storage and addition are performed under microprocessor control. (3) A signal generator whose output frequency is set to frequency F, which has an output that generates a radio frequency signal when operated, and has a large band frequency control element and a small band control element, A main oscillator whose oscillation frequency can be varied over a small frequency band by linearly varying the parameters of the electrical signal supplied to the element, A. From the input information, the main oscillator B. means for deriving a sample signal from the output of the master oscillator; C. a stabilizing fixture for generating a reference signal F(ref); A harmonic spectrum of a signal having at least one component within the small frequency band, that is, a comb-like array F(ref); 2F(ref); 3F(ref)
)...nF(ref) (referred to as a harmonic spectrum signal); F. a mixing or sampling circuit that produces a signal having a frequency that is the arithmetic difference between the components of the wave spectral signal; The main oscillator tuning (caused by the interaction of the main oscillator output signal and one component N, F(ref) of the harmonic spectrum signal) until a beat signal is detected for the purpose of frequency calibration. The controller is programmed to vary the frequency control signal supplied to the controller, and at this point the value of the frequency determining signal supplied to the master oscillator sets the oscillation frequency equal to (N, F(ref) + X). (where X is N, F(ref)).
and the required frequency shift between the desired frequency F). (4) In a method for controlling the oscillation frequency of a main oscillator in a signal generator in which a linear frequency control element has a limited frequency sweep capability, A. a frequency control signal that logically generates a frequency F at the output of the main oscillator; B. combining a sample of the master oscillator output signal with a reference signal having one component within said limited sweep frequency of the master oscillator; C. a beat signal between the master oscillator output signal and the reference signal component; D, then adjust the value of the frequency control signal from the value corresponding to the calibration to an algorithm or lookup table that describes the frequency response of the master oscillator to the value of the frequency control signal. A method for controlling an oscillation frequency comprising steps of adjusting the main oscillator output to a new value calculated using the oscillator, thereby shifting the main oscillator output to a desired frequency. (5) When the master oscillator is required, A. Using the corrected control signal as a starting point,
Vary the calculated control signal according to an algorithm that combines the oscillation frequency with the value of the control signal; B. inhibit the output of the signal generator when the final target frequency of the sweep is reached; C. then the beat signal 5. A method as claimed in claim 4, including the steps of repeating the correction process comprising the method described above until detected again, after which the sweeping process can be started again. (6) If the reference signal is a harmonic spectral signal, and A, identify a series of key frequencies through the range to be swept, starting at the lowest or highest, and by the interaction of the components of the reference signal with the main oscillator output signal. B. making a correction to the control signal supplied to the main oscillator until a beat signal is generated; B. generating a control signal for obtaining a frequency between the first frequency and a next target frequency using the corrected control signal; C, repeating the calibration process for the oscillator control signal as the next target frequency is approached, and then continuing to change the oscillator frequency through the next frequency range to the next target frequency. An oscillation frequency control method according to claim 5, comprising steps.