JPS59156185A - Controlling method of induction motor - Google Patents

Abstract

PURPOSE:To control the voltage and torque of a motor by detecting a motor voltage, obtaining the parallel component and the perpendicular component to an exciting current, controlling the primary frequency so that the parallel component becomes zero and correcting the exciting current so that the perpendicular component becomes the prescribed value. CONSTITUTION:A voltage detecting transformer 14 detects a motor voltage, voltage component detectors 15, 16 respectively obtain the parallel component ed and the perpendicular component eq to the exciting current component decided by a motor voltage control system, an oscillator 22 is controlled so that the parallel component ed becomes zero to control the primary frequency, and the exciting current im is controlled by a voltage deviation amplifier 24 and a sample-holding circuit 25 so that the perpendicular component becomes the prescribed value.

Description

DETAILED DESCRIPTION OF THE INVENTION [Field of Application of the Invention] The present invention relates to a method for controlling an induction motor, and more particularly to a method for controlling an induction motor for highly controlling the torque and torque of the motor.

[Background of the invention]

So-called vector control is popular, in which the current of an induction motor is divided into an excitation component and a torque generation component, each of which is controlled independently to perform speed control with a high speed response and a high speed. In vector control, the motor current is controlled in proportion to a primary current command pattern signal that is calculated and synthesized based on each current component command. Whether each component inside the motor is controlled according to the command value is related to the control accuracy of the subbetical frequency (one missing frequency λ).The current is decomposed into the excitation component and the torque generation component using the subbechi frequency as a medium. It is thought that
Therefore, unless the slip frequency is at an appropriate value, each component cannot be controlled consistently to the command value.

The reason why all 9 frequencies are not appropriate values is the 2nd frequency of the electric motor.
Examples include that the secondary resistance changes due to temperature, and that the excitation inductance changes due to iron core saturation (I:F). This effect causes torque response delay and pulsation when the load changes, and also causes problems such as the inability to control the motor voltage (magnetic flux) according to the command value.

[Purpose of the invention]

The object of the present invention is to solve the above-mentioned problems and to control a control method for an induction generating machine that can always control the overall frequency and excitation current to appropriate values even if the secondary resistance and excitation inductance fluctuate. Our goal is to provide the following.

[Summary of the invention]

The feature of the present invention is to calculate the motor voltage, obtain the parallel component and orthogonal component to the excitation flow component determined by the motor voltage control system, and set the primary frequency so that the parallel voltage component becomes zero. At the same time, the excitation current is fully commanded so that the orthogonal pressure component becomes a predetermined value, and when the rotation speed becomes lower than the set value, the magnitude of the parallel voltage component and orthogonal d pressure component is maintained at the set value. This is because I decided to give money to Yuuri.

[Embodiments of the invention]

FIG. 1 shows an example of an actual flow of the present invention.

In Figure 1, 1 is a PW consisting of a switching element such as a gate turn-off thyristor, a diode, etc.
M inverter, 2 is an induction motor, 3 is a speed command circuit for commanding the total rotational speed, 4 is a speed detector, 5 is a speed deviation amplification device for amplifying the total deviation between the speed command signal and the detection signal;
7 is an excitation current command device that commands the entire excitation current of the #1 machine, 8
is an adder; 9 is a two-phase sinusoidal flow command pattern signal 1 based on the excitation current command signal 1 from the adder 8 and the torque flow command signal 5 ft from the amplifier 5;
10 is a three-phase sinusoidal current command pattern signal I/ based on the signal 1781β.

11 is a current detector that detects the instantaneous value of the motor heart flow; 12 is a current detector that compares the current command pattern signal with the magnetic current detection signal; This is a comparator with hysteresis characteristics that outputs a PWM signal for on/off control, a 13-tri comparator, and a gate circuit that supplies gates and signals to switching elements. Note that although there are circuits corresponding to the 11 to 13\'iU phase outputs, and similar circuits corresponding to the V and W phases, they are omitted from illustration. 14 is a 4-voltage detection transformer, 15.16 is a voltage component detector that detects all axial components of the orthogonal rotational coordinate system of the motor voltage, and 17 is a voltage component detector that detects when the rotational speed is "slow". 1-high speed”
A speed high/low discriminator outputs to 1N $ of HJ, 18 is a number code that outputs the output signal of detector 15 as it is when the rotation speed is high speed J, and stores and outputs the entire output signal when the rotation speed is 1 low speed. A hold circuit 19 is a slip frequency command circuit that outputs a slip angular frequency command signal ω8* proportional to the ratio of the signal 1t* and the signal 1ffi*; a multiplier that outputs a slip frequency correction signal Δω;

22 is an oscillator that outputs a two-phase sinusoidal inverse signal with a frequency proportional to the frequency command signal; It is a command device for awakening, inducing and inducing power. Note that when the frequency is constant, the output signal of the adder 21 becomes the voltage command signal, so the command device is not necessary. 24 is a voltage deviation amplifier that amplifies the entire deviation between the three acid command signals and one output word of the detector 16; 25 is a voltage deviation amplifier that outputs the output signal of the amplifier 24 as is when the rotational speed is "high speed"; It is also a sample-and-hold circuit that holds, stores, and outputs the output signal when the speed is "low."

Next, the operation of the above circuit will be explained. First, an overview of the present invention will be described. The control principle of vector control is as follows. That is, assuming that one axis 2d of the orthogonal rotating magnetic field coordinate system and 41I perpendicular thereto are the q-axis, the primary current ci
, q-axis component 11d r Fq can be controlled as shown in the following equation, ild can be controlled in accordance with the exciting current i, , and lI can be controlled in correspondence with the torque generating current 1t.

l'+l=N/"Confucian angle 7 ・・・・・・(
1) 1 1”°...
....) T2 1td Here, 11 : Primary current ω, : Slip angular frequency T2 : Secondary time constant θ : Phase of the primary current with respect to d@, that is, the magnitude of the primary current (1) control according to the formula,
If the slip angular frequency is controlled according to equation (2), and the phase of the primary current is controlled in relation to equation (3), then ild
Depending on the magnetic flux φ, also! 19, all torques T can be controlled independently. At this time, as shown by the following equation, the torque T is controlled by 1j without response delay with respect to '+qK.
4) Here, k: proportionality constant Next, according to the relationships of equations (1) to (3), l111. The circuit operation for controlling ω and θ will be explained.

First, the operation of the circuit in which the primary current is controlled according to equations (1) and (3) will be described.
Fully outputs a two-note string wave signal with a frequency proportional to the frequency command signal from. These signals have a phase difference of 90 degrees from each other and serve as a phase reference signal for a current command pattern signal, which will be described later. The coordinate converter 9 receives flow command signals 1ffl*, t,
*Based on the output number 18 of the oscillator 2'2, calculate the following formula to obtain the current index of the 2-note string wave.

Furthermore, in the phase number converter lO, a three-phase sinusoidal current command pattern signal 1u-1, , is obtained according to the following formula:
u-1, can be expressed as in the following equation.

i, *==Acos(ωlt+θni −== ACO3(ω11−”−yr〇)i,*
= ),, cos ((dl t ten-rc ten)
−・−・(7) Here, A−1m”+1t”
#θ-tan-'=1m* The current command pattern signal 1u and the current detection signal 18 are compared in the comparator 12, and if the deviation between both signals is greater than a predetermined value, the output signal polarity of the comparator 12 is reversed. An action is taken. As a result, the comparator is turned on (GT<),
A PWM signal for off control is extracted. In the V phase and W phase, P is similarly set by a comparator (not shown).
A WM signal is taken out. As a result, the output current of each phase of the inverter is a signal! . It is controlled to be proportional to ~l. In this way, 1゜((11d is controlled in proportion to it, and ilq is controlled in proportion to it9, and the above-mentioned (1) and (3)
The relationship in Eq.

' Next, the operation that satisfies equation (2) will be explained.

In the adder 21, the total frequency command signal (signal i-
(proportional to), the speed detection signal is increased, and a frequency command signal is taken out (the operations of the detector 15, old circuit 18, and multiplier 20 will be described later). That is, the motor primary angular frequency ω1 is determined according to the quintic equation.

ω1-ω, +ω1 ・・・・・・(
8) Here, ω-The detected value ω- of the two electrical rotational angular frequencies:
The command value of the horizontal angular frequency and the actual value ω of the horizontal angular frequency will be shown in the next section.

ωwork=ω1+ω, ・・・・・・(9
) Here, ω2: Actual value of electrical rotational angular frequency Here, if ω1 described above coincides with ω1, then ω, *-ω, from the relationship of equations (8) and (9).

Furthermore, since ω and * are determined according to equation (10), the slip d angular frequency is ultimately controlled to follow equation (2).

'Here, the relationships of equations (1) to (3) are satisfied as the T2.secondary time constant is greater than or equal to the reference value.

However, since the secondary time constant changes (due to changes in the secondary resistance value due to temperature changes or changes in excitation inductance due to the influence of iron core saturation), generally T 2 = T
Therefore, the slip frequency controlled according to the equation (10) cannot satisfy equation (2).

The influence of such control errors causes an increase in motor voltage, low torque, and torque pulsation during torque control, which adversely affects the motor, the inverter, and the load driven by the motor.

As described later, the influence of this control error can be detected as a fluctuation in the generator voltage (magnetic flux), so the control error is compensated for by controlling the primary frequency and excitation current of the Kuradoiso using the fluctuation from the reference value. can do. In addition, in very low frequency operation, voltage detection becomes uncertain, so this compensation method is ineffective, but in that case, just before entering very low frequency operation, the function signal of the quadratic time constant extracted up to that point is and the excitation current compensation signal are retained and memorized,
By performing the above-mentioned control using this, the compensation operation can be performed continuously. The compensation principle of the present invention will be explained in more detail below.

FIG. 2 shows the variation of the motor flux with respect to the slip angular frequency ω when the signals l− and i,* are constant. φd and φ are the magnetic flux components of the d and q axes, and φ is the vector composite magnetic flux of φd and φ. Figure (a) shows the rated value where 5- is positive, (
b) shows the case where i- is zero, and (C) shows the case where it* is a negative rated value. In Figure (a), the mark X is the normal operating point where φd=φ2 (reference value) and φq='O1'' are satisfied.

The overall angular frequency ω at this operating point is expressed by equation (2), but it is due to the fluctuation of the second-order time constant mentioned above! When the overall angular frequency ω, which is controlled according to equation (10), no longer matches the previous ω (i.e., when ω changes), φ becomes
The polarity changes with the normal operating point as φ,=¥=0. Therefore, when φ,〉0, the missing frequency f1 is incremented by 1, and φ
When , , is 0, the operating point returns to the normal state by lowering it and correcting and controlling ω, * according to φ. That is, (
2) Operation that satisfies the equation can be performed. Figures (b) and (
In C), similar operation can be performed by the above-mentioned control.

'On the other hand, the motor magnetic flux φ2 at the normal operating point is expressed by the following equation.

φ2;φd ””Mjta Bright...(11) Therefore, if the excitation inductance M fluctuates due to the influence of iron core saturation, etc., the motor magnetic flux (pressure at ') cannot be maintained at the command value, but in this case, the excitation Current command signal io*, φd
It is possible to compensate for the influence of fluctuations in M by making corrections according to fluctuations in M from the reference value.

The above is the compensation principle for satisfying equation (2) and maintaining the magnetic flux (') at the command value.Next, the operation of the above relationship in the circuit shown in FIG. 1 will be explained.

The voltage component detectors 15 and 16 detect two axes of the prime mover induced motive force according to equation (12), that is, a component ed with the same phase and a component e with a 90 degree phase difference with respect to the excitation current phase reference signal. are detected respectively.

Here, ed: output signal e of the detector 15,: output signal V C=va Vu-V of the detector 16,: motor phase voltage rt: primary resistance tl, L2': primary and secondary leakage inductance iB: d @ sleep current proportional to t, q-axis current proportional to 1 The above calculation is based on the motor voltage detection signal, the output signal of the oscillator 22, and the frequency command obtained from the adder 21. signal ω
1 and current command signal i:, i-, it is clear that this can be done using multipliers and adders. Na2
The second term on the right side of the equation is for compensating for the influence of the leakage impedance drop of the motor.

The signals e and e and the aforementioned φd and φ have the following relationship.

Q4==-ω1φ.

e, = ω1φ, ...
...(13) That is, the signal corresponding to ED is
Also, a signal corresponding to e and φd is detected. 1d No.e
d is added to an adder 21 via a hold circuit 18 and a multiplier 20. At this time, in the case of ed negative (φ, >0), the output signals of the adder 21 are added with a polarity that increases the inverter output frequency. In this manner, ω1 is controlled so that ed=o(φ, 20) at all times according to the above-described principle, and the slip angular frequency ω6 is controlled to the value of the normal operating point. At this time, the compensation signal Δω of the slip angular frequency outputted from the multiplier 20 corresponds to the difference between ω shown in equation (2) V and ω shown in equation (lO). Therefore, it is expressed by the following formula.

The multiplier 20 produces a signal proportional to ω and a hold signal. The secondary time constant T2 is a function of the secondary resistance and the secondary inductance, but the variation in the secondary resistance is a long change in the time constant related to the thermal time constant of the secondary winding, and the excitation inductance can be regarded as constant in the medium and low speed operating range where magnetic flux weakening control is not performed, so the response of the circuit from the output of the detector 15 to the input of the multiplier 2o may be slow. During extremely low speed operation, the voltage detection becomes uncertain, so the above-mentioned compensation operation cannot be performed.
For the reasons mentioned above, the compensation operation can be performed continuously as follows. In other words, the duration of extremely low speed operation is usually sufficiently short compared to the thermal time constant of the winding, and during this period, the output signal of the hold circuit 18 is changed to the value immediately before entering extremely low speed operation. To store and store the value 9
, the compensation operation can be performed continuously. The discriminator 17 is for causing the hold circuit 18 to perform the above-mentioned operation, and generates a signal instructing the hold operation when the vehicle is in the extremely low speed operating range.

On the other hand, the signal e is the frequency command signal ω in the amplifier 24.
1, the deviation is amplified, and it is added to the adder 8 via the hold circuit 25. As a result, ωl and eq are controlled to be proportional, and at this time, eq/ωl = φd (constant)...
The relationship (15) holds true, and the magnetic flux φd becomes constant.

On the other hand, in the magnetic flux weakening operation region [&, the induced electromotive force command signal from the command unit 23 and the signal e are matched, and the same operation as described above is performed. At this time, the relationship eq=ωlφd (constant) (16) is established, the magnetic flux φd is inversely proportional to ω1, and magnetic flux weakening control is performed.

By the way, the excitation current compensation signal Δ10 outputted from the hold circuit 25 by ρ corresponds to the excitation current necessary to generate the above-mentioned magnetic flux φd when the excitation inductance is the standard value M, and the actual value M of the excitation inductance. Since it corresponds to the difference between the two required excitation currents, it is expressed by the following equation.

By the way, during very low speed operation, voltage detection becomes uncertain, so the above-mentioned compensation operation cannot be performed, but in the medium and low speed operation range where magnetic flux weakening control m is not performed, the magnetic flux and excitation inductance M remain constant. As is clear from equation (17), Δi is also one pair. Therefore, during extremely low speed operation, the compensation operation can be continued by holding and storing the output signal of the hold circuit 25 at the value immediately before starting the operation.

This operation of the hold circuit 25 is performed in accordance with the signal from the discriminator 17, as in the case of the bold circuit 18 described above. The output signal of the hold circuit 25 can be directly applied to the coordinate converter 9 by omitting the command unit 7 and the adder 8. The output signal of the hold circuit in this case is the compensated excitation current command i. -A, but it is clear that the same control as described above can be performed. By controlling the voltage (magnetic flux) as described above, the following effects can be obtained.

1) Due to manufacturing errors in the motor, even if the excitation inductance M does not match the design value, the exciter voltage (magnetic flux)
can always be controlled to a predetermined value. Therefore, the initial adjustment of 1, .

22 Due to the influence of iron core saturation, the excitation inductance M
Even if the motor voltage (magnetic flux) may fluctuate, the motor voltage (magnetic flux) can always be kept at a predetermined value i'1ilIf111. Therefore, highly accurate magnetic flux (weakening) electric current lJ1 wholesale can be performed.

3) Even if the difference occurs during extremely low speed operation, the compensation operations of 1) and 2) can be continued.

As described above, according to the present invention, even if the secondary resistance and excitation inductance are fm, the overall frequency and excitation current can always be controlled to appropriate values.

In the embodiment described above, e, full detection' is corrected to excitation current command signal 1□ according to the variation from the command value, and eq(
φd) was controlled to a predetermined value, but since φ and φd have a similar relationship as shown in Figure 2, eq(α
Instead of φd), induced electromotive force e(-v'e d2+ e
A similar effect can be obtained by controlling q"αφ) to a predetermined value. In this case, the control circuit uses an amplifier to replace the detection signal of the magnitude of the motor voltage with the output signal of all the detectors 16. 2
4 is different from that in FIG. 1, but the rest is the same.

In the embodiment described in @, ed and e were detected, φd was controlled to a predetermined value, and φ was controlled to be zero, but φd and φ were calculated and detected, and φd, φ
, similar control can be performed by controlling . FIG. 3 is a circuit diagram of this embodiment. The magnetic flux detector 26 converts the motor voltage into two-phase AC signals va and vβ according to the following equation, and detects magnetic fluxes φα and φβ (two-phase AC No. 100) by integrating them.

Here, Zl indicates that the influence of the leakage impedance drop of the motor is compensated using the primary current i (actual value or command value n) in order to improve the detection accuracy of the magnetic flux.

The magnetic flux component detectors 15' and 16' detect the two axes of magnetic flux, that is, the same phase and 90 degree phase difference components with respect to the excitation current phase reference signal, respectively φQ: # 15
' The signal φ is applied to the hold circuit 18. ratio signal φ
d is matched with the magnetic flux command signal from the magnetic flux command circuit 23' and applied to the magnetic flux deviation amplifier 24'. Increased width-24
' outputs a signal according to the fluctuation of φd, which is sent to the adder 8
added to. The other parts and their operations are the same as those in FIG. 1, and the compensation operation described above is performed in this embodiment as well, so that the same effects as in FIG. 1 can be obtained. Note that since φ and φd have a similar relationship as described above, the detector 1
Similar control can be achieved by using a magnetic flux detector that detects φ (=V50]7) in place of p of 6'.

〔Effect of the invention〕

As explained above, according to the present invention, even if the secondary resistance and excitation inductance vary, all frequencies and excitation current can always be kept at appropriate values, and this can be done stably even when the motor rotates at low speed.

The above two or more embodiments are examples of application to PWM < pulse width modulation) inverters, but this invention applies to other types of inverters as long as their output frequency and output voltage (current) can be controlled. The invention can be applied. In addition, although the above embodiments have been described based on application examples to devices that perform speed control #,
The present invention can be similarly applied to a system in which the signal i- is adjusted by something other than a speed regulator, such as a torque control system.

[Brief explanation of drawings]

FIG. 1 is a block diagram showing an example of the present invention, FIG. 2 is a characteristic diagram for explaining the present invention in detail, and FIG. 3 is a block diagram showing another embodiment of the present invention. 1... Inverter, 2... Induction motor, 15.16
...Voltage component detector, 18.25... Sample and hold circuit, 24... Voltage deviation amplifier, 22... Oscillator. '020 (Ko) Stone n 2nd mouth (C) 6

Claims (1)

[Claims] ■. In an induction motor control method that detects the rotational speed of the induction motor, controls the magnitude of the primary current and the frequency plate, and independently controls the primary torque flow and the excitation current, the motor voltage of the induction motor is controlled. The motor voltage control system detects the parallel component and modified component for the determined excitation first current component, and controls the primary frequency of the induction motor so that the parallel voltage component becomes zero, and also controls the orthogonal voltage component. When the rotational speed of the induction heart motor becomes lower than the set value, the magnitudes of the parallel acid three components and the orthogonal voltage component are adjusted to be the same as when the rotational speed is at the set value. A method of controlling an inductive heart motor in which the % characteristic is that the control is maintained at a certain value.