JPH07118959B2 - Induction motor control method - Google Patents

Description

The present invention relates to a method for controlling an induction motor, and more particularly to a control method for controlling the voltage and torque of the motor with high accuracy.

[Conventional technology]

There is known a so-called vector control in which the current of an induction motor is divided into an excitation component and a torque component and each is independently controlled to perform high-speed response and highly accurate speed control. In this configuration, the electric motor current is controlled based on each current component command signal. Whether or not the magnetic flux is controlled according to the command value inside the motor depends on the iron core magnetic saturation characteristics of the electric motor. Due to the change in the magnetic flux of the electric motor, a response delay and pulsation of the torque occur when the load changes, and a problem occurs such that the electric motor voltage cannot be controlled according to the command value.

Therefore, conventionally, as disclosed in Japanese Patent Laid-Open No. 59-156184, a method has been proposed in which the output voltage of the inverter is detected and the exciting current command is corrected based on this voltage detection signal.

[Problems to be Solved by the Invention]

However, the above-mentioned conventional technique requires a detector for detecting the output voltage of the inverter, which causes a problem that the hardware configuration becomes complicated.

An object of the present invention is to provide a control method which eliminates the detector of the output voltage and compensates the magnetic flux change due to the iron core magnetic saturation of the electric motor to always control the magnetic flux to an appropriate value.

[Means for Solving the Problems]

The purpose is to calculate the amount of fluctuation of the voltage command from the reference value based on the deviation between the torque current command signal of the electric motor and the detected value of the torque current in the rotating magnetic field coordinate system, and to generate the exciting current command in accordance with this voltage fluctuation amount. This is achieved by modifying the signal.

[Action]

In the one in which the motor voltage is controlled so that the deviation between the torque current command signal of the motor and the detected value of the torque current in the rotating magnetic field coordinate system becomes 0, the fluctuation amount of the voltage is 1
Although the voltage drop component due to the change of the secondary resistance and the voltage fluctuation component due to the magnetic flux change due to the magnetic saturation of the iron core are included, the influence of the change of the primary resistance is small in the high primary angular frequency range. Further, since magnetic flux weakening control is required in a high primary angular frequency range, in the present invention, the fluctuation component of the motor voltage is determined based on the fluctuation amount of the voltage command signal, and the fluctuation component due to iron core magnetic saturation is determined from this fluctuation component. Since the signal related to the change in the magnetic flux is taken out and the reference value of the exciting current command is corrected so that it becomes 0, the detector of the output voltage is unnecessary.

〔Example〕

The first embodiment of the present invention will be described below with reference to FIG.
The induction motor 2 is supplied with power from the power conversion circuit 1,
The three-phase AC current flowing in the is detected by the current detectors 3U, 3V, 3W. This three-phase alternating current is converted into a two-phase alternating current by the three-phase / two-phase converter 4, and its output is the coordinate converter 5.
Are vector-divided into exciting current and torque current components in the rotating magnetic field coordinate system, and the detected values are matched with respective command signals by adders 6 and 7. The rotation speed of the electric motor 2 is detected by the speed detector 8, the output of which is matched with the command signal ω _r * in the adder 9 and the adder 9 is added.
Entered in 10. The output of the adder 9 is input to the speed controller 11 to output the torque current command signal I _q * of the electric motor 2, and the output thereof is input to the slip calculator 12 and the adder 7. The output signal ω _s * of the slip calculator 12 is added to the rotation speed ω _r in the adder 10, and the output signal ω ₁ * is input to the coordinate reference generator 13 and the voltage command calculation circuit 14. The output of the adder 6 is the exciting current command signal I _d ** and the detected value I _{d of the} coordinate converter 5.
And the output signal is input to the current controller 15, and the voltage component command signal Δv _d * of the rotating magnetic field coordinate system is output to the adder 21 so that the deviation becomes zero. The output signal of the adder 7 outputs a deviation signal between the torque current command signal I _q * and the detected value I _q of the coordinate converter 5, and the output signal is input to the current controller 16 and the deviation becomes zero. Thus, the voltage component command Δv _d * of the rotating magnetic field coordinate system is output to the adder 22. The voltage command calculation circuit 14 calculates the voltage component command signals v _d * and v _{q of the} rotating magnetic field coordinate system based on the excitation current command signal I _d **, the torque current command signal I _q *, and the primary angular frequency command signal ω ₁ *. * And are calculated, and the output signals are input to the adders 21 and 22. The output signals of these adders 21 and 22 are input to the coordinate converter 17, and the voltage command signals v _u *, v _v *, v _w * of the stator coordinate system are output based on the output signals of the coordinate reference generator 13. To do. These voltage command signals v _u *, v _v *, v _w * are input to the pulse width modulation (PWM) control circuit 19 and output signals v _u , v _v , v _w PWM-controlled based on the carrier frequency. Is output to the power converter 1. The speed command signal ω _r * is input to the magnetic flux command generator 25, its output signal φ * is input to the excitation current command calculator 24, and its output signal I _d * is output to the adder 23. Also the current controller
The output signal Δv _q * of 16 is input to the arithmetic circuit 18, and the motor 2
An exciting current command signal ΔI _d * for compensating the magnetic flux change due to the iron core magnetic saturation is output to the adder 23. This exciting current command signal ΔI _d * is added to the exciting current command signal I _d * in the adder 23, and the corrected exciting current command signal I _d ** is output to the adder 6.

First, the operation of the above configuration will be briefly described. In this embodiment, the current command signals v _d * and v _q are based on the exciting current command signal I _d ** and the torque current command signal I _q * in the rotating magnetic field coordinate system.
* Is calculated, and a feedback control system is constructed for the exciting current and the torque current, and the voltage command signal is output by the output signal of each current controller so that the deviation between the command value and the actual value becomes zero. v _d *, v _q * is corrected, and the corrected voltage command signal v _d **, v _q ** is converted into the three-phase AC voltage command signal of the stator coordinate system by the coordinate converter 17,
The voltage supplied to the electric motor 2 is controlled by the power converter 1 based on the converted three-phase AC voltage command signal.

Next, details and operation of the arithmetic circuit 18 relating to the present invention will be described with reference to FIG. The motor 2 is normally a voltage command calculation circuit 14
Is carried by a voltage determined by the output signals v _d *, v _q * of the. By the way, in the exciting current command calculating circuit 24, the exciting current I _d * is calculated from the magnetic flux command signal φ * using the mutual inductance M and the secondary time constant T ₂ with respect to the exciting current at the rated voltage of the motor 2 as shown in the following equation. Although the calculation is performed, since the electric motor 2 has the magnetic saturation characteristics of the iron core as shown in FIG.
During flux weakening, the mutual inductance M, the actual value of the secondary time constant T ₂ is changed, no longer matches the mutual inductance M, a secondary time set value of the constant T _2.

Here, S is a differential operator.

Further, the speed controller 11 corrects the torque current command signal I _q * so that a required torque is generated in the electric motor 2, and compensates the torque change due to the difference between the magnetic flux inside the electric motor and the magnetic flux command signal φ *. As a result of the above operation, in the high primary angular frequency range where the magnetic flux weakening control is required, the output signal Δv _q * of the torque current of the current controller 16 is different from the magnetic flux inside the motor and the magnetic flux command signal φ *. The output voltage component by is output. Its size is expressed by the following equation.

Δv _q * = (φ * −φ _d ) · ω ₁ * (2) Therefore, the correction amount ΔI _d * of the exciting current command for the mutual inductance M is expressed by the following equation.

Here, k is a proportional gain. That is, when the primary angular frequency ω ₁ * is positive and the internal magnetic flux φ _{d of the} motor is smaller than the magnetic flux command signal φ *, ΔI _d * becomes positive and acts to increase the exciting current command, and conversely φ _{When d} is larger than φ *, ΔI _d * becomes negative and the exciting current command is decreased. These operations can automatically consider the sign of ω ₁ *.

Here, S _gn (ω ₁ *) outputs the polarities +1 and −1 of ω ₁ *.

The expression (2a) or the expression (2b) is the operation content of the operation circuit 18.

According to this embodiment, the mutual inductance M (= Δφ / ΔI _d ) used in the equation (2a) or (2b) also changes due to the magnetic saturation of the iron core, but since the closed loop control is performed, the mutual inductance change is also included. The exciting current can be corrected by automatically compensating.

FIG. 3 shows a second embodiment of the present invention. The same parts as those in the first embodiment shown in FIG. 1 are designated by the same reference numerals, and the description thereof will be omitted. The difference from the first embodiment is that the switch 26
Is set to the a side, the adjustment operation is performed prior to the actual operation, the characteristics of the induced electromotive force e _q * of the induction motor 2 with respect to the output I _d * of the exciting current command generator 27 is measured, and the relationship between them is written in the memory. Then, the mutual inductance M * and the second-order time constant T ₂ * are calculated by the equations (6) and (7) described below, and the fourth value is calculated.
As shown in the figure, these are written in the memory corresponding to the magnetic flux φ *. In actual operation, switch 26 is on the b side, M *, T ₂ * is read from the memory based on the magnetic flux command φ *,
Based on this, in the exciting current command calculation circuit 24, I _d **
It is the point that is calculated. The induced electromotive force e _q * can be calculated from the output signal v _q ** of the adder 22 by the following equation.

e _q * = v _q **-r ₁ * I _q * -ω ₁ * (l ₁ * + l ₂ ′ *) ・ I _d
* (4) Here, r ₁ * is the primary resistance, and l ₁ *, l ₂ ′ * is the set value of the primary and secondary leakage inductances.

Also, magnetic flux φ *, mutual inductance M *, second-order time constant T ₂
* Is expressed by the following equation.

φ * = e _q * / ω ₁ *… (5) M * = e _q * / (ω ₁ * · I _d *)… (6) T ₂ * = (M * + l ₂ ′ *) / r ₂ * (7) where r ₂ * is the secondary resistance setting. Excitation current command signal
I _d ** is given by the following equation.

I _d ** = φ * · (1 + T ₂ * · S) / M * (8) where S is a differential operator.

According to the present embodiment, the feedforward compensation can be performed based on the magnetic flux command signal φ *, and the influence of the disturbance due to the operation of the current control system is not affected as compared with the first embodiment.

FIG. 5 shows a third embodiment of the present invention. The same parts as those in the first embodiment shown in FIG. 1 are designated by the same reference numerals, and a description thereof will be omitted. The difference from the first embodiment is that the magnitude of the mutual inductance M * used in the exciting current command calculator 24 is modified based on the q-axis voltage command signal Δv _q * to obtain the exciting current command signal I _d **. This is the point I asked for. According to this embodiment, the same effect as that of the first embodiment can be obtained.

In the above, each of the embodiments has been described in the form of a block by an analog circuit in order to make the explanation easy to understand, but it goes without saying that it can be implemented by digital control using a microprocessor.

〔The invention's effect〕

According to the present invention, a magnetic flux weakening control can be performed with high accuracy by compensating for a magnetic flux change due to iron core magnetic saturation of an electric motor without using an output voltage detector of an inverter.

[Brief description of drawings]

FIG. 1 is a control block diagram showing the first embodiment of the present invention, and FIG.
FIG. 3 is a characteristic diagram for explaining the relationship between magnetic flux and exciting current when the induction motor has iron core magnetic saturation, FIG. 3 is a control configuration diagram showing a second embodiment of the present invention, and FIG. 4 is a second embodiment. FIG. 5 is a characteristic diagram for explaining the operation principle of the example, and FIG. 5 is a control configuration diagram showing the third embodiment of the present invention. 1 ... Power converter, 2 ... Induction motor, 13 ... Coordinate reference generator, 14 ... Voltage command calculation circuit, 15, 16 ... Current controller, 18 ... Calculation circuit.

Front page continued (72) Inventor Junichi Takahashi 52-1 Omika-cho, Hitachi City, Ibaraki Hitachi Ltd. Omika Plant, Ltd. (72) Inventor Shigeru Sugiyama 5-2-1 Omika-cho, Hitachi City, Ibaraki Incorporated company Hitachi Ltd. Omika factory (56) References JP-A-63-95879 (JP, A)

Claims (1)

[Claims] 1. An electric current of an induction motor is d- in a rotating magnetic field coordinate system.
The excitation current component (d-axis) and the torque current component (q-axis) on the q-axis are divided, and each is independently controlled based on each current component command signal, and at least the current controller uses the torque current command signal and the torque current detection signal. In the control method of the induction motor for controlling the voltage on the q-axis in the rotating magnetic field coordinate system so as to eliminate the deviation between the magnetic field weakening control and the induction current command signal from the magnitude of the magnetic flux command. In a region where the primary angular frequency is high, the amount of fluctuation of the q-axis voltage command signal obtained from the current controller from the reference value is calculated, and the exciting current command signal is corrected by the amount of fluctuation, so that the induction motor A method of controlling an induction motor, which is characterized in that a change in magnetic flux due to magnetic saturation of an iron core is compensated.