JPS58184842A - Receiver - Google Patents

Abstract

(57) [Summary] This bulletin contains application data before electronic filing, so abstract data is not recorded.

Description

[Detailed Description of the Invention] The present invention is completely compatible with communication systems, particularly △M radio transmission systems that transmit two types of signals using carrier waves, for example, broadcast bands A to 1 of monaural and stereo receivers. Get AM
This invention relates to a system for transmitting and receiving stereo signals with almost no distortion.

There are various methods for transmitting and receiving the N1 array signal. The simplest method is an unmodified orthogonal signal method in which two carrier waves having the same frequency and orthogonal phase are used to transmit two interpolated signals Δ and B, such as a left (L) signal and a right (R) signal.

This system is similar to the NTSC system defined in the US Color Prevision Transmission, which is used to transmit two types of color signals using one type of carrier wave. However, in current monaural receivers that use signal current rectifiers to extract C audio signals, there is a frequency distortion proportional to twice the frequency of the signal. jshiAshin'y4 /J<
It is derived from an old fact that M-hontsuri is expressed by the following formula.

The term in the radical sign here represents the amplitude, and is assumed to be φ-tan 1(L--R)/(1+L+R).

However, in a monaural receiver, the amplitude of the received signal is approximately the sum of the carrier wave amplitude and the audio signal amplitude, which is
+L 10R). Because of this (1--R
) term represents distortion, so this term is a squared term! The frequency distortion is doubled. The φ term also represents phase modulation and will not produce an output from the conventional envelope detector of a mono receiver unless there is significant amplitude or phase distortion in the signal throughout the scheme.

Other conventional systems employ techniques for transmitting a single carrier that is amplitude modulated with (L+R) information and frequency modulated with (LR) information. In this case, the complex spectrum of the transmitted signal will cause undesirable distortion in monaural and stereo receivers, assuming that the received signal has 1' wavenumber or phase distortion. (+-R) If the signal contains a low-frequency Therefore, the FM component is subjected to spurious conversion for amplitude modulation.

Furthermore, in the other 6 equations, the sum and difference signals are transmitted in an orthogonal relationship, but the amplitude of the envelope is compensated to make them compatible (I F
The R> component is distorted. For this purpose, the in-phase component is maintained from (1+l+R) so that the magnitude of the orthogonal component does not change. This distorts the phase of the stereo information and increases the number of sidebands, resulting in a significant increase in distortion in monophonic and stereo receivers. An object of the present invention is to provide an AM stereo broadcasting system that is compatible with current AM monaural receivers by only slightly modifying current transmitters and slightly complicating the stereo decoding circuit of the receiver. It's in the middle of the day.

In order to obtain an exceptional stereo signal with the communication system of the present invention (
The L+R) component, or monaural information, and the phase, or stereo information, are included in the transmitted signal, and the (1-R) component, or difference information, is not included in its envelope. Therefore, the signal for the monaural circuit is the same as for normal AM monaural transmission. The desired changes to the transmitter are small and therefore the circuitry for the ΔM SDE receiver is less complex. The present invention multiplies the orthogonal signal by a factor related to the phase of the stereo information at the transmitter, and at the stereo receiver, the received signal is multiplied by the same factor as described above. This is based on the fact that the complete original orthogonal signal can be reproduced in 1q.

The invention will be explained with reference to the drawings.

The conventional AM orthogonal communication system shown in FIG. 1 and the compatible communication system according to the present invention shown in FIG. As will be explained, the present invention is not limited to this, and can of course be applied to a method of transmitting and receiving two arbitrary types of signals using a single carrier wave.

The communication system according to the present invention shown in FIG. 3 and the communication method shown in FIG. 1(
As is clear from the unmodified and compatible orthogonal system of 11, the 1-alternate transmitter 10 has a block that supplies the signal component (1+L+R') from the first input terminal 11 to the first modulator 12. [ ] A Durham signal path and a signal path for supplying the signal component (LR) from the second input terminal 133 to the second modulator 14 are provided. Further, the carrier wave signal generated from the RF exciter 1b is supplied to the first modulator 12 and is also supplied to the 90° phase shifter 1.
6 and then supplied to a second modulator 14. The outputs of the rain tone modulators 12 and 14 are adjusted by the signal adder 17, and the signals sent as usual are adjusted to 5! make it possible to survive. This signal can be expressed mathematically using the following equation.

Coconi φ-tan 1 (1--R) / (1+L+R
). When this signal is received by the stereo receiver i18 and demodulated by the product detector 1 and the multipliers 20 and 21, special signals (14L+R) and (t-R) are obtained. However, the signal output demodulated by the envelope detector 22 of the monaural receiver 23 can be expressed by the following equation.

This output level = R, that is, it can be used only for monophonic signals.

The phase vector of FIG. 2 shows the trajectory 24 of the modulated and transmitted signal for the communication system of FIG.

The phase vector 25 indicates the unmodulated carrier wave 1cosoot, the phase vector 26 indicates the in-phase modulation signal (L+R),
Phase vector 27 indicates a quadrature signal (+-R). Further, φ represents the instantaneous phase angle of the composite phase vector 28, and as is clear from the locus 24, this angle cannot be greater than ±45°.

FIG. 3 shows an AM stereo broadcast system compatible with the present invention. In the present invention, there are also two input terminals 11' (1+
L+R) and 13' (+----R) from the transmitter 30
are connected to two transformers 12' and 14', respectively. RF exciter 15' and 90° phase shifter 16' are also connected in the same manner as described with respect to FIG. The output horns of the modulators 12' and 14' are tightened with the signal IJII Korean device 17',
Amplitude changes are removed by a limiter 31 so that only phase information remains. The phase modulated carrier wave thus obtained is converted into a signal component (
1+L+R). sent
The number is expressed as (1+L+R)cos(ω°C+Φ).

This signal is the original stereo (C) from adder 17'.
Multiplied by OSφ, ie (1+L+R), this restored signal is completely compatible. That is, when this signal is received by the monophonic receiver 23' and demodulated by the envelope detector 22', the output is the signal component (L - +,
R). When the transmitted signal is received by the stereo receiver 33, the amplitude of this signal is limited by the limiter 34.

The stereo information obtained in this way is applied to V C0 by the multiplier 35.
Comparison with the phase of COSωt from No. 36 is reversed. This V C0
36 is synchronized with the phase of the RF exciter 15' of the transmitter 30, as will be described later. Therefore, the phase difference in this case becomes COSφ, and the output of the multiplier 35 is also proportional to COSφ.

A collector circuit '37 shown in detail later in FIG. 7 divides the signal by the output of the multiplier 35, thereby reproducing the original stereo output of the adder 17'. VCO3
The signal COSω from 6 is input to phase shifters 3B and 39
45[deg.] and supplies the power of the collector circuit 37 to multipliers 40 and 41, respectively. Therefore, the multipliers 40 and 41 generate outputs of the sum of 1- and R and the DC term, respectively.

FIG. 4 shows a deformation locus 45 of the phase vector of the transmitted signal at a5 in the method of the present invention shown in FIG. Each point in the trajectory 45 is C
This corresponds to each point within the locus 24 multiplied by OSφ. By multiplying COSφ in this manner, it is possible to generate a minimum number of high-order sideband waves corresponding to the transmission of a compatible monophonic signal with minimum distortion.

The transmitter of the invention is shown in more detail in FIG. In a monaural transmitter, an RF signal consisting of a crystal oscillator
A carrier wave from exciter 15' is supplied to modulator 32.

The desired processing circuit 49, which in this case transforms the output of the oscillator according to the invention, is shown within the dotted line. The frequency of the carrier wave from the excavator 1b' is divided and one of the frequencies is shifted by 90° by a phase shifter 16'. Then 5>? The +11 and 2-) autologous carrier waves are supplied to modulators 12' and 14', and the outputs of these modulators are supplied to a 7) and 11-equipped circuit 17'. A portion of the phase-shifted and unmodulated WI signal is also supplied to adder 17' via a carrier level controller 50 which determines the level of the unmodulated carrier. The output of adder 17' is sent to limiter 3.
1 to remove the C amplitude modulation component, thereby allowing an unmodulated carrier having only phase or stereo information to be provided to the high level modulator 32. Program channel input terminal 52 (1,) a3 and 53 (R
) are connected to program level limiters 54 and 55 and monitoring instruments 56J3 and 57, respectively. Also, the signal component (L-
+ R) is formed.

Furthermore, the 1 (signal) is inverted by an inverter 60 and supplied to an adder 61 connected to the multiplier 14', where it is combined with the 1 signal to form a signal component (LR). (1+ R) feeding the second output of adder 58 to high level modulator 32 via time delay circuit 62;

A delay time equal to the delay time of the signal processing circuit 49 is obtained by the d delay circuit 62. Because of this, the modulator 32
The output is a signal that is amplitude modulated with (L + R) information and phase modulated with stereo information.

FIG. 6 shows the stereo reception section 33 of FIG. 3 in more detail. The received signal is supplied to an RF mixing-IF amplification stage 65. Since this RF mixing-IF amplification stage 65 is a conventional one, a description of its operation will be omitted.

The amplitude modulation component of the signal at the output terminal 66b of the M combining amplifier stage 65 is removed by the limiter 34. The output of this limiter 34 is expressed as COS (ω°C 1φ), and this output is supplied to one input side of a multiplier 35, which is an in-phase detector, and one input side of a multiplier 70, which is a quadrature detector. Also feed on the side. This multiplier 70 is a phase-locked loop 7
It constitutes one integrating stage. The low pass filter 72 prevents sudden phase changes from reaching VC03G, but allows phase drift to pass through. Thus, the output of the VCO 36 is very constantly controlled and can be fed to the π 2 90° phase shifter 73 since it is orthogonal to the output of the transmitter oscillator 15'. Output COS of phase shifter 73
ωt is supplied to the second input terminal of the multiplier 35. Multiplier 35
The output IQcO3φ appearing at the output side 74 of is supplied to the rectifier circuit 37. Collector circuit 3, which will be described later with reference to FIG.
7, the im signal at the output terminal 66a of the mixing-amplification stage 65 is divided by the output of the multiplier 35 so that an orthogonal signal can be reproduced. The portion of the circuit of FIG. 6 is similar to that described with respect to FIG.

FIG. 7 shows the multiplier 35 and Tl of the receiver 33 in FIG.
3 shows the rectifier circuit 37 in more detail. The output of the limiter 34 is supplied to the input terminal 80 of the multiplier 35, which is a phase detector. The output of limiter 34 alternately switches the active pair of transistors 81 and 82 into a conductive state 101 times to the incoming carrier. Further, a reference input signal at a terminal 84 taken out from the phase locked loop 71 is supplied to a current source 83 constituted by a transistor C via a phase shifter 73. This phase shifter 73 also acts as a low-pass filter and supplies a substantially sinusoidal reference current to the transistor current +1iii83. The DC reference voltage 1t at the base 85 of the I-funister 82 is supplied from an emitter follower 88.

This J-mitter follower 88 is connected to a differential amplifier pair 81.82. In addition, a current mirror 87 applies the output current to the output side 74 of the differential amplifier pair to balance any static current from the transistor current source and 33.
and 84 are proportional to the cosine of the angular difference between the input signals. The current pulse from multiplier 35 is smoothed by integrating capacitor 86.

In order to bring the output of the output side 74 of the multiplier 35 sufficiently close to a cosine function, it is necessary to substantially eliminate high-order harmonics at one of the input terminals 80 or 84.

Therefore, the phase shift network 73 is used as a low pass filter to remove odd harmonics from the square wave of the C oscillator.

” The collector circuit 37 includes a pair of transistors 100 and 1
01 with a stationary amplifier is more than θj'. The current in the emitters of transistors 100 and 101 is supplied from current NQ 102. Also, since two transistors 103 and 104 form a C current mirror, the current of transistor 104 is equal to that of transistor 1.
00 current. transistors 100 and 1
When the currents of transistor 01 are equal, the current of transistor 104 is equal to the current of transistor 101, and current 10 becomes zero.

The signal voltage taken out from the signal input section 66a is connected to two resistors 1.
08 and 109.2 diodes 110 and 11
1 and the transistor 100 via the reference voltage source 112
and , are supplied between the bases of =1, respectively. This reference voltage source 112 consists of three resistors 114.11! b and 1
Emitter follower 113 coupled to a voltage divider consisting of 16
Consists of. The base of the non-transistor 113 is connected to the connection point of the resistors 114 and 115 to obtain a reference voltage. The emitter of the [mitter] wire 113 supplies a low impedance reference voltage to the transistor pair 100 and ]01 forming the C differential amplifier.

Current Ir from multiplier 35 flows through diodes 110 and 1
11, resistors 108 and 109, voltage source 112 and human power source 66 to forward bias diodes 110 and 111.

The forward impedance of diodes 110 and 111 and the resistors 108 and 109 form the base of transistor 100 and transistor 1.
The voltage 11- supplied between the bases of 01 and 11 is connected to the diode 11.
The ratio of the forward resistances of 0 and 111 to the resistances 108 and 109 is reduced by the ratio.

Next, the collector circuit 37 is connected to its current and the output I of the multiplier 35.
This will be explained using r-ImaXCO3φ. output current l
o=l+I'S, denoted by '117, and let 11 be the current supplied by the current source 102. Is is the human-powered current at the terminal 66a, expressed as e8/'2r. Here, 2r is made equal to the sum of two extremely large resistances 91. Also, e8 is e. (1+l'+R)cos(
ωC[+φ), and let eo be the width of the unmodulated carrier wave. Furthermore, 1 is the peak QQ current of the transistor 83. Therefore, the following formula is formed.

1 = (leo(1+l-+R) cos (ωc
t +φ))2rJj and 1o=(1+eo(1+-L,+R)COs (ωC
[+φ), 12rl cosφO ax In this case, COSφ-(1-+-L, + R) / (
1 Do] - 10R) 2+(L-R)
-'(1+ ecz2r Imax) (10L+
R)2+ (L R)2co's (ωct-+φ
) This is the desired quadrature signal C.

FIG. 8 shows another example of a receiver compatible with the desired operation according to the present invention. In this example, the receiver circuit 37 is connected to the audio section of the receiver, and two identical receiver circuits 37a and 17b are connected. ) The output 66 of the MAF mixer-IF amplifier stage 65 is a single output and is connected to the multipliers 40 and 41. When the output of the multiplier 40 is L cosφ, [this is the collector circuit 37
a, and then divides it by COSφ to obtain an L output. The output of the multiplier 41 is set as RCO3φ, and this is supplied to the rectifier circuit 37b, where COSφ
to obtain the UR ratio output. For this reason, the output current on the output side 74 of the multiplier 35 is divided and the two collector circuits 3
7a and 37b.

8I! FIG. 0 shows yet another example of the receiver of FIGS. 7 and 8. In this example, two input terminals 831J-3 and 74 of collector circuit 37C are connected to phase shifter 73 and multiplier 35, respectively. The output side 95 of the collector circuit 37G is connected to the input side of the phase shifters 38 and 39, and the COS
It is assumed that the reference voltage J is divided into φC. This is why multiplier 4
The outputs of 0 and 41 become [ and R signals, respectively.

FIG. 10 shows an example of a transmitter similar to the transmitter of FIG. 5 in the -6SSB communication system, that is, an auto-communication system that changes with COSφ. In this example, the 1 and H human signals are added by the adder 58, and the adder 61'C' is reduced.The output of the sieve 61 is phase-shifted by the phase shifter 9! In the same way, it is supplied to the transmitter.In addition, in the stereo receiver of Soso, the decoding angle is changed (
L+R) output 96 and (1,-R><π/2 output 97
Make it possible to extract. This output 97 is phase-shifted by 1π7'2 by a phase shifter 98, and the output is supplied to a matrix circuit 99 of the receiver in the same manner as the output 96.

Therefore, the outputs of the nine matrix circuits are 1- and R
It becomes a signal.

FIG. 11 shows the receiver of FIG. 10 in more detail. In other words, the input side of the collector circuit 37 is connected to the RF mixer of the receiver.
IF amplifier stage 'e 5 is connected to the output 66 of the collector circuit 37 and the output of the collector circuit 37 is connected to the multipliers 40 and 41, the phase-locked loop and phase-shifting network being similar to that described with respect to FIG. In this example, one output 97 of the multiplier is phase-shifted by 4, and the outputs of both multipliers are supplied to the matrix circuit 99, and the R output is connected to allow it to occur.

FIG. 12 shows a signal spectrum when the L signal of the transmitted signal is included in one set of sideband waves and the R signal is included in another set of sideband waves. Also, this transmitted signal has 2
Of course, a higher-order complementary II sideband that is transmitted with twice the sideband is included.

FIG. 13 shows an example of another single-sideband communication system similar to the communication system of FIG. In this example, one of the program input signals, for example, the R signal, is phase-shifted by 90 degrees by a phase shifter 95. Next, the phase-shifted signal is supplied to the adder 58, and the phase-shifted signal is supplied to the adder 58.
0 and is then supplied to the adder 61. The second program signal 8, for example, is fed directly to adders 58 and 61. The outputs of these adder devices 58 and 661 are (L+R<
π/2) signal and (L, R<π/2) signal. These signals are in the case of a transmitter that performs cosine complement 1 [and fi'il
+7i modulated by carrier wave. When performing cosine correction on the 11th transmission and reception, the complementary IF signal l- and 1 sentence (π・'22 times, in this case, the R signal is changed to -90' phase by the phase shifter 98. There will be a delay.

Figure 14 shows the spectrum of transmission No. 1.1 when the sum and difference signals are transmitted on the middle sideband (In this case, the information is transmitted on the 2-8 sideband.

Therefore, by sifting the monophonic signal by the cosine of the angle π before transmitting it and dividing it by the same cosine at the receiver, monophonic reception aC can be completely balanced by the communication IJ formula, and a stereonotonic receiver can be used. It can generate signals that are easily recognized. In this case, φ is the angle formed by the vector sum of the first orthogonal carrier waves and the bisector of the angle between the two orthogonal carrier waves. All the transmitted signals are within the envelope detector (there is a gain that allows quadrature modulation without distortion. Therefore, the monophonic interchromatic components lost by the U, t medium waves can be minimized and the R Therefore, the communication system of the present invention can be made compatible with a seven-noft nick receiver by using both envelope detection a5 and l+JI Ill detection.1. Synchronous detection In order to optimize the characteristics of the vessel] Rector (
Although it requires a correction circuit, satisfactory characteristics can be obtained even with an unmodified synchronous receiver.

[Brief explanation of the drawing]

Figure 1 is a block diagram showing a conventional communication system that transmits and receives two types of signals that are orthogonally amplitude modulated using carrier waves. Figure 2 is a phase diagram showing the carrier wave and sideband waves transmitted using the communication system shown in Figure 1. 814 is a phase vector diagram showing the signal transmitted by the communication method of FIG. 3, and FIG. 3 is a diagram showing the desired operation of the present invention. FIG. 6 is a 10-block diagram showing an example of a receiver that is compatible with the desired operation of the present invention, and FIG. Circuit diagram showing part of the machine in detail, 8V! Figure E3 is a Bucock diagram showing another example of a receiver that is compatible with the Honko Akira 1# system; Figure t) also shows another example; - Figure 11179 shows the IjssB communication system, Figure 11 is a block diagram supporting the receiver of the communication system in Figure 10, Figure 12 is a spectrum diagram of the signal transmitted using the communication system in Figure 10, FIG. 13 shows another example of the SS13 communication method. FIG. 14 is a spectrum diagram of a signal transmitted using the communication method shown in FIG. 10... Orthogonal transmitter it, 11'... First input terminal 12, 12'... First modulator 13.13'... Second manual input terminal f, 11.1114
．． 14'...Second curve adjuster 15, 1! l'...Exciter 16, 16'...90° phase shifter 17.17'...Signal decorator 18...Stereo receiver 20.21...Multiplicative detection Ware (Johanware) 22.22'
... Envelope detector 23.23' ... Monaural receiver 24.45 ... Locus of phase vector 25.26.27 ... Phase vector 28 ... Combined phase vector 30 ... Transmitter 31. 34...Limiter 32...High level modulator (multilayer) 33...Switch A reception II 35...Multiplier (common mode detector) 36...VCO37... Collector circuit 3
8.39...45'' Phase shifter 40.41... Multiplier 49... Signal processing circuit 50... Carrier wave level controller! 12.53... Program channel input terminal 54
．． 5ri...prog humure,, bell limiter 56.57
... Monitoring instrument 58 ... Adder 60 ... Inverter 61 ... Tightening device 62 ... Time delay circuit
65...RF mixing-IF amplification stages 66a, 66b--
・Output terminal (65) lO...East sieve (quadrature detector) 71...Phase synchronization (1) loop 72...Low pass filter 73...π,'2 (90°) Phase shifter 95... Phase shifter 98... Phase shifter special crystal')
Applicant Mo]-]” - F. Inco Vol. 9 Figure 101ffl

Claims (1)

[Claims] 1, amplitude modulated by signal information proportional to the sum of the first and second information signals (A) and (F3) [one φ=ta
n −' (C+ (AR)/ (C2+A+8)
) (where 01 and C2 are constants) in a receiver for receiving a broadcast carrier signal phase modulated by a signal relative to an angle φ of into an intermediate frequency carrier signal and a mixer that converts the amplitude and phase modulation information into an intermediate frequency carrier signal.
an intermediate frequency amplifying device for amplifying an intermediate frequency carrier signal with a bandwidth sufficient to A correction and demodulation device that processes the output signal and generates a signal substantially equal to the first and second information. 2. A receiver according to claim 1, characterized in that the correction and demodulation device includes a device for dividing the output of the amplifier device by a signal proportional to the angle φ. 3. The receiver according to claim 2, wherein the signal proportional to the angle φ is made proportional to the cosine of the angle φ. 4. The receiver includes an oscillation device, a limiter device that limits a signal proportional to the output of the amplification device, and a first power receiving the outputs of the oscillation device and the limiter device and supplying the output to the correction and demodulation device. The receiver according to scope 2 of the patent, further comprising an IM device. 5. The receiver includes a first phase shifter that shifts the output of the oscillation device by 45 degrees, a second power sieve device that receives and multiplies the outputs of the first phase shifter and the correction device, and an oscillation device. The present invention is characterized in that it includes a second phase shifter that shifts the phase of the output of the second phase shifter and the demodulator by 45 degrees, and a third multiplier that receives and multiplies the outputs of the second phase shifter and the demodulator. A receiver according to claim 4.