JPH1198891A - Torque controller for induction motor - Google Patents

Abstract

PROBLEM TO BE SOLVED: To perform highly accurate high speed control by estimating a revolu tion frequency using the difference of the torque current component obtained by subjecting a current error vector, represented by the difference between each command and actual value of the exciting current component or the torque current component of a primary current in phase with or orthogonal to the primary flux vector of an induction motor, to coordinate conversion. SOLUTION: A current component command operating means 4 receives a torque command to an induction motor 1 and an estimated value ωm * of the revolution frequency from a first integrator 8 and outputs an exciting current component command Io * and a torque current component Iqs *. A speed error estimating means 13 receives the d-axis component Ids and the q-axis component Iqs * (torque current component) of the primary current and a phase angle Ψ obtained through coordinate conversion of a current error vector from a current component operating means 5. A first integrator 8 receives the speed error Δω between the actual value ω of the revolution frequency of the primary flux vector and an estimated value ω* obtained through control operation and outputs a revolution frequency ^ωm . The torque τm of the induction motor 1 is controlled to match the torque command m*.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a torque control device for an induction motor, and more particularly to a speed detector and a voltage (magnetic flux).
The present invention relates to a torque control device for an induction motor suitable for performing high-accuracy and high-speed response torque control over a wide range of operation without using a detector, without using high-speed and low-speed rotation speeds and powering and regenerative operations. It is.

[0002]

2. Description of the Related Art As a conventional torque control device for an induction motor that does not use a speed detector and a voltage (magnetic flux) detector, there is one having the configuration shown in FIG. 13 disclosed in Japanese Patent Application Laid-Open No. 9-9700.

A torque control device for an induction motor shown in FIG.
Induction motor 1, current detector 2, PWM inverter 3, current component command calculation means 4, current component calculation means 5, slip frequency calculation circuit 6, subtractor 7, integrators 8, 10, adder 9, current component control circuit 11a, a voltage command calculation circuit 12, and the current control means includes a current component control circuit 11a, a voltage command calculation circuit 12, and a PWM inverter 3.
It is composed of This control device sets the primary current command to be controlled based on the primary magnetic flux of the induction motor so that the primary current command at the time of high-speed rotation or overload can be obtained.
This is to suppress the fluctuation range of the next voltage.

FIG. 14 is a block diagram showing a detailed configuration of the current component command calculation means 4. 14, 20 is an absolute value circuit, 21 is a function generator, 22 and 24 are coefficient units,
23 is a divider.

FIG. 15 is a block diagram showing a detailed configuration of the slip frequency calculating circuit 6. As shown in FIG. In FIG. 15, 30,
32, 34 and 36 are coefficient units, 31 is a differentiator, 33 is an adder, 35 is a subtractor, and 37 is a divider.

FIG. 16 is a block diagram showing a detailed configuration of the current component control circuit 11a. In FIG. 16, 40,
43, 46, 50 and 52 are coefficient units;
And 53 are subtractors, 42 and 51 are multipliers, 44 is a divider, 48 and 54 are amplifiers, and 49 and 55 are adders.

Next, prior to describing the operation of the torque control device for an induction motor, the control principle of the control device will be described.

First, a rotating coordinate axis rotating at a primary frequency
The voltage-current equation of the induction motor (on the d-q axis) is expressed by the following equations (1a) to (1d) as is known.

[0009]

(Equation 1)

However, in the above equation (1), V _ds and V _qs
Are the d-axis and q-axis components of the primary voltage, I _ds and I _{qs, respectively.}
Are the d-axis and q-axis components of the primary current, I _dr and I _{qr, respectively.}
Are the d-axis and q-axis components of the secondary current, respectively. Also,
R _s is the primary resistance, R _r is the secondary resistance, L _s is the primary self inductance, L _r is the secondary self inductance, M is the primary secondary
Secondary mutual inductance, p is the differential operator (= d / dt)
It is. Further, ω is the rotation frequency of the primary magnetic flux vector,
That is, it is the primary frequency and ω _s is the slip frequency.

Next, the d-axis component and the q-axis component of the primary magnetic flux are given by the following equations (2a) and (2b), respectively.

[0012]

(Equation 2)

Now, the direction of the primary magnetic flux vector is d
When they are made to coincide with the axis, the generated torque τ _m becomes as in the following equation (3).

[0014]

(Equation 3)

Here, _Pm is the number of pole pairs of the induction motor.

The generated torque τ _m is the d-axis component Φ _{ds of the} primary magnetic flux
And the product is proportional to the product of the q-axis component of the primary current I _qs and the control becomes easy. Therefore, the d-axis component Φ _ds of the primary magnetic flux is matched with the command value Φ _ds ^* given by the equation (4), and the direction of the primary magnetic flux vector is matched with the d-axis (the q-axis component of the primary magnetic flux). Φ
_qs = 0), the slip frequency ω _s
As shown in equation (5), the d-axis component I _ds of the primary current is given as in equation (6).

[0017]

(Equation 4)

Here, I ₀ ^* is an exciting current component command, and σ is a leakage coefficient {= 1−M ² / (L _s L _r )}.

[0019] From (6), the exciting current component I ₀ corresponding to the exciting current component command I ₀ ^* is expressed by the following equation.

[0020]

(Equation 5)

Therefore, if the excitation current component I ₀ is controlled so as to coincide with the command value I ₀ ^* , the d-axis component Φ _ds of the primary magnetic flux can be made to coincide with the command value Φ _ds ^*. The direction of the next magnetic flux vector can be made to coincide with the d-axis.

Here, the primary frequency ω and the slip frequency ω _s
As is well known, there is a relationship such as the following equation (8).

[0023]

(Equation 6)

Here, ω _m is the rotation frequency (rotation speed) of the induction motor.

Thus, the direction of the primary magnetic flux vector is d
When the axes are matched, the equations (1a) and (1
The equation (b) is as shown in the following equations (9a) and (9b).

[0026]

(Equation 7)

Therefore, the command values V _ds ^* and V _qs ^* of the d-axis and q-axis components of the primary voltage are given as in the following equations (10a) and (10b).

[0028]

(Equation 8)

Here, ω ^* is a set value (estimated value) of the primary magnetic flux vector used for the control calculation.

From equation (6), the d-axis component I of the primary current
It gives the d-axis component command I _ds ^* corresponding to _ds at the following equation (11).

[0031]

(Equation 9)

[0032] Here, equation (8), in order to set the setpoint of the primary flux vector omega ^*, the value of the rotational frequency omega _m of the induction motor is required. However, this control method is a control method that does not use a speed detector, and it is necessary to calculate the rotation frequency of the induction motor as an estimated value ω ＾ _m of the rotation frequency of the induction motor by some alternative method. Since the primary magnetic flux vector is obtained using ω ＾ _m , the equation (8) is obtained by the following equation (12).
Is replaced by

[0033]

(Equation 10)

The estimated value ω of the rotation frequency of the induction motor
The operation principle of ＾ _m is as follows.

If the control is performed using a PWM inverter so that the primary voltage of the induction motor matches the command value, ω
^{If *} = ω, then I _qs ^* = I _qs and I _ds ^* = I _ds hold. However, in the case of ω ^* ≠ ω, I _qs ^* ≠ I _qs and I _ds ^* ≠
_Ids . Therefore, I _qs ^* = I _qs and I _ds ^* = I _ds
If ω ^* that satisfies is satisfied, ω ^* = ω, and the estimated value of the rotation frequency of the induction motor can be made to match the actual value.

Therefore, the primary current q-axis component command I _qs ^*
And the q-axis component I _qs is integrated, and the integration result is an estimated value ω ＾ _m of the rotation frequency of the induction motor, I _qs ^* = I _qs in a steady state, and ω ^* = ω holds. At that time the estimated value ω ^ _m is consistent with the actual value ω _m.

Here, the d-axis component I _ds and the q-axis component I _qs of the primary current can be obtained using the following equations (13a) and (13b).

[0038]

[Equation 11]

Here, θ is the phase of the primary magnetic flux vector, and is obtained by integrating the above primary frequency ω, that is, the rotation frequency of the primary magnetic flux vector.

Also, the primary voltage commands V _us ^* , V _vs ^* and V _ws
^* Can be obtained using the following equation (14).

[0041]

(Equation 12)

Next, the specific operation of the torque control device for an induction motor shown in FIG. 13 will be described. First, when the torque command τ ^* _m of the induction motor 1 and the estimated value ω ＾ _m of the rotational frequency of the induction motor 1 output from the first integrator 8 are input to the current component command calculation means 4, the excitation current component command I ₀ ^* and the torque current component command I _qs ^* are output.

More specifically, as shown in FIG. 14, an estimated value ω of the rotational frequency of the induction motor 1 is calculated by the absolute value circuit 20.
^ Absolute value of _m | _ω ^ _m | is output. Further, when this absolute value | ω ＾ _m | is input to the function generator 21, the following equation (1) is obtained.
The calculation of 5) is performed, and the excitation current component command I ₀ ^* is output.

[0044]

(Equation 13)

Where I _0-100% is the rated value of the exciting current component,
ω _b is the rated frequency of the induction motor 1.

Next, when this exciting current component command I ₀ ^* is input to the coefficient unit 22, the operation of equation (4) is performed, and the d-axis component command Φ _ds ^{* of the} primary magnetic flux is obtained. Further, the divider 23
When the torque command τ ^* _m of the induction motor 1 input from the outside is divided by the d-axis component command _Φds ^* of the primary magnetic flux and then input to the coefficient unit 24, the torque is obtained from the relationship of the equation (3). The current component command I _qs ^* is output.

Next, 1 is detected by the current detector 2.
When the secondary currents I _us (U phase), I _vs (V phase) and the phase θ of the primary magnetic flux vector output from the second integrator 10 are input to the current component calculating means 5, the calculation of the equation (13) is performed. Then, the d-axis component I _ds and the q-axis component I _{qs of the} primary current are output.

Subsequently, the deviation between the torque current component command I _qs ^* output from the current component command calculating means 4 and the q-axis component (torque current component) I _qs of the primary current output from the current component calculating means 5 is calculated. After having been obtained by the subtractor 7, this deviation
, An estimated value ω ＾ _m of the rotational frequency of the induction motor 1 is output.

Next, the exciting current component command I ₀ ^* output from the current component command calculating means 4 and the d-axis component I _ds and the q-axis component I _qs of the primary current output from the current component calculating means 5 are shown. When input to the slip frequency calculation circuit 6, the slip frequency ω _s
Is output.

More specifically, as shown in FIG. 15, the coefficient units 30 and 32, the differentiator 31, and the adder 33 provide (5)
The operation of the numerator on the right side of the equation is performed. On the other hand, the coefficient unit 34,
The calculation of the denominator on the right side of equation (5) is performed by 36 and the subtractor 35. Therefore, the adder 3 is added by the divider 37.
When the output of 3 is divided by the output of the coefficient unit 36, the operation of Expression (5) is performed, and the slip frequency ω _s is obtained.

Subsequently, when the output of the first integrator 8 and the output of the slip frequency calculation circuit 6 are added by the adder 9, the calculation of the equation (12) is performed, and the set value of the rotation frequency of the primary magnetic flux vector is obtained. ω ^* is required. Further, the set value ω ^* of the rotation frequency of the primary magnetic flux vector is
When input to 0, the phase θ of the primary magnetic flux vector is output.

Next, the exciting current component command I ₀ ^* and the torque current component command I _qs output from the current component command calculating means 4
^* , The set value ω ^* of the rotation frequency of the primary magnetic flux vector output from the adder 9, the d-axis component I _ds and the q-axis component I _qs of the primary current output from the current component calculating means 5, respectively.
When input to the current component control circuit 11a, a primary voltage d-axis component command _Vds ^* and a q-axis component command _Vqs ^* are output.

More specifically, as shown in FIG. 16, a signal obtained by inputting the d-axis component I _ds of the primary current to the coefficient unit 40 is subtracted by the subtractor 41 from the excitation current component command I ₀ ^*. When it is reduced, (I ₀ ^* −σI _ds ) is output. Then, the multiplier 4
After squaring the q-axis current component (torque current component) I _qs by ² and inputting it to the coefficient unit 43, σI _qs ² is output.
Further, the divider 44 converts this σI _qs ² into a subtractor 4
After dividing by (I ₀ ^* −σI _ds ) output from 1 and subtracting from the d-axis component I _{ds of the} primary current by the subtractor 45,
(7) calculation is performed of the exciting current component I ₀ is output.

Subsequently, when the d-axis component I _ds of the primary current is input to the coefficient unit 46, a d-axis voltage drop due to the primary resistance is output. Further, in order to make the d-axis component of the primary current coincide with the command value, the exciting current component command I ₀ ^* and the exciting current component I ₀ output from the subtractor 45 are output by the subtractor 47.
, The difference is amplified by an amplifier 48, the output of the amplifier 48 and the output of the coefficient unit 46 are added by an adder 49, and the obtained signal is converted into a primary voltage represented by the following equation (16). _Is output as the d-axis component command V _ds ^* .

[0055]

[Equation 14]

On the other hand, the exciting current component command I ₀ ^* is
, The operation of equation (4) is performed, and the d-axis component command Φ _ds ^{* of the} primary magnetic flux vector is output. Furthermore, this Φ
When the _ds ^* and the set value of the rotational frequency of the primary flux vector omega ^* is multiplied by the multiplier 51, ω ^* · Φ _ds ^* is output.
When the torque current component I _qs is input to the coefficient unit 52,
The q-axis voltage drop due to the primary resistance is output.

Further, in order to make the torque current component I _qs coincide with the command value, a difference between the torque current component command I _qs ^* and the torque current component I _qs is obtained by a subtractor 53, and this difference is amplified by an amplifier 54. At the same time, the output of the amplifier 54 and the outputs of the multiplier 51 and the coefficient unit 52 are added to an adder 55.
And the obtained signal is represented by the following equation (17).
Output as the q-axis component command V _qs ^* of the next voltage.

[0058]

(Equation 15)

Next, the d-axis component command V _ds ^* and the q-axis component command V _qs ^{* of the} primary voltage output from the current component control circuit 11a ^.
And the phase θ of the primary magnetic flux vector output from the second integrator 10 is input to the voltage command calculation circuit 12, and (14)
Equations are calculated and the primary voltage commands V _us ^* , V _vs ^* and V _ws
^* Is output. Further, the PWM inverter 3 controls the primary voltage of the induction motor 1 so as to match the above-mentioned command.

By the above operation, I ₀ ^* = I ₀ and I _qs ^* =
I _ds and I _qs are controlled so that I _qs holds. Accordingly, the d-axis component Φ _ds of the primary magnetic flux matches the command value, and the set value ω ^* of the primary magnetic flux vector output from the adder 9 becomes ω ^* = ω, which matches the actual value. Furthermore, the second
Output from the integrator 10 coincides with the phase of the primary magnetic flux vector. As a result, the generated torque τ _m of the induction motor 1 is controlled so as to match the torque command τ ^* _m .

[0061]

In the torque control device for an induction motor described in the prior art, torque control can be performed without using a speed detector and a voltage (magnetic flux) detector. However, the control device described above has a problem in that if the regenerative operation is performed while the induction motor is rotating at a low speed, the control system becomes unstable and control becomes impossible.

[0062] That is, the sign of the slip frequency omega _s when represented by (12) performing a regeneration operation, the code and the opposite sign of the estimated value omega ^ _m of the rotational frequency of the induction motor. Especially when trying to perform regenerative operation at low speed rotation ｛ω _S nearly equal
As the state approaches ω { _m }, the absolute value of the set value ω ^* of the rotation frequency of the primary magnetic flux vector becomes {| ω ^* | nearly equal 0} as shown in Expression (12). At this time, the sign of the deviation between the q-axis component command I _qs ^{* of the} primary current and the q-axis component I _qs has the opposite sign to the sign of the deviation when control is normally performed in the power running operation. That is, obtains an estimated value of the rotation frequency of the induction motor to match the I _qs to I _qs ^*, (1
Even if the control is performed by _obtaining the q-axis component command V _qs ^* of the primary voltage by the calculation of the expression 0b), it does not converge in the direction of I _qs = I _qs ^*, and conversely, I _qs ≠ I _qs ^*. It diverges in the direction.
Therefore, control cannot be performed so that ω ^* = ω.
The calculation of the primary frequency ω cannot be performed normally and the control becomes impossible. From the above, it has been necessary to limit the operation region of the induction motor to the region excluding the low-speed regeneration operation so that the conventional control device does not lose control as described above.

The present invention has been made in order to solve the above-mentioned problems, and has a wide operating area including a low-speed rotation and a regenerative operation without using a speed detector and a voltage (magnetic flux) detector. Accordingly, it is an object of the present invention to provide a torque control device for an induction motor that realizes always stable torque control performance with high accuracy and high-speed response.

[0064]

According to a first aspect of the present invention, there is provided a torque control device for an induction motor, wherein a primary current of the induction motor is converted into a component having the same phase as a primary magnetic flux vector by a current component calculating means.
(Excitation current component) and an orthogonal component (torque current component). The primary current is calculated by the current component command calculation means based on the command signal of the torque to be output by the induction motor and the estimated value of the rotation frequency of the induction motor. The excitation current component command and the torque current component command are calculated.
The primary current of the induction motor is controlled so that the current components match the respective current component commands, and the torque error component command, the torque current component and the excitation current component command are controlled by a speed error estimating means. Based on the exciting current component, an error (speed error) between the actual value of the rotation frequency of the primary magnetic flux vector and the set value of the rotation frequency of the primary magnetic flux vector used in the control calculation is calculated. Is integrated by a first integrator to obtain an estimated value of the rotation frequency of the induction motor, thereby obtaining the rotation frequency of the induction motor. The slip frequency operation circuit calculates the excitation current component command or the excitation current component. And the slip frequency of the induction motor is determined from the torque current component command or the torque current component, and an adder is used to estimate the rotation frequency of the induction motor. Values and the induction motor rotational frequency and without the addition to the primary flux vector slip frequency,
Further, the second integrator integrates the rotation frequency of the primary magnetic flux vector to obtain the phase of the primary magnetic flux vector and calculates the phase of the primary magnetic flux vector, thereby achieving high-speed response and Thus, a highly stable torque control characteristic can be obtained.

According to a second aspect of the present invention, in the torque control device for an induction motor, the speed error estimating means according to the first aspect is configured such that the deviation of the torque current component command and the torque current component, the excitation current component command and the excitation current component command. The current error vector on the orthogonal coordinates represented by the deviation of the current component is subjected to coordinate conversion, and the actual value of the rotation frequency of the primary magnetic flux vector and the set value of the rotation frequency of the primary magnetic flux vector used in the control calculation are calculated. And the deviation of the excitation current component command after the coordinate conversion and the excitation current are output. Further, the current control means controls the primary current using the deviation as an input, thereby achieving low-speed regeneration. The control system is always kept stable over a wide range of operation, including during operation, and high-speed response torque control characteristics can be obtained.

According to a third aspect of the present invention, there is provided a torque control device for an induction motor, wherein the actual value of the rotational frequency of the primary magnetic flux vector and the rotational frequency of the primary magnetic flux vector used in the control calculation are calculated by the speed error estimating means. Estimation calculation of the error (speed error) from the set value of the above is performed by calculating the current error on the orthogonal coordinates expressed by the deviation between the torque current component command and the torque current component and the deviation between the excitation current component command and the excitation current component. When the vector is coordinate-transformed, the conversion phase angle of the coordinate transformation is switched by the powering / regeneration operation of the induction motor, so that the operation is always stable, high-precision, and high-speed over a wide range of operation, including low-speed regeneration operation. The torque control characteristic of the response is obtained.

According to a fourth aspect of the present invention, in the torque control device for an induction motor according to the fourth aspect, the switching of the conversion phase angle of the coordinate conversion performed by the speed error estimating means is performed by
By performing the operation through the buffer circuit, the degree of the change in the conversion phase angle at the time of switching is suppressed, and vibration and shock transiently generated at the time of switching are reduced.

According to a fifth aspect of the present invention, in the torque control device for an induction motor according to the first aspect, the speed error output from the speed error estimating means is determined according to an absolute value of a rotation frequency of the primary magnetic flux vector. By adjusting, an equal control response is obtained over a wide range of operation.

[0069]

BEST MODE FOR CARRYING OUT THE INVENTION

Embodiment 1 Before describing embodiments related to the first and second aspects of the present invention, a method for estimating a speed error which is a basis of the present invention will be described.

As described in the control method in the prior art, the q-axis component command I _qs ^* and q-axis component I _qs of the primary current, the d-axis component command I _ds ^{* of the} primary current, and the d-axis component I _ds
With respect to the above, if a set value ω ^* of the rotation frequency of the primary magnetic flux vector that satisfies I _qs ^* = I _qs and I _ds ^* = I _ds is obtained, ω ^* = ω, and the actual value of the rotation frequency of the primary magnetic flux vector is obtained. ω can be calculated. However, the deviation of the I _qs ^* and I _qs deviation and I _ds ^* and I _ds can be speed error is constant, 1
It varies depending on the condition of the actual value ω and torque tau _m of the rotational frequency of the next magnetic flux vector. This is illustrated in FIGS. 4 and 5.

FIG. 4 shows that the induction motor performs a power running operation (rated output torque τ _m100% ), and a certain speed error Δω (= ω ^*).
−ω)> 0, an error generated in the primary current due to the speed error is represented as a current error vector on the dq coordinate axis while changing the rotation frequency ω of the primary magnetic flux vector. Things. That is, the primary current of the q-axis component command I _qs ^* and q-axis component I _qs deviation ΔI _qs (= I _qs ^* -
I _qs ) is the q-axis component of the current error vector, and the deviation ΔI _ds (= d between the d-axis component command I _ds ^{* of the} primary current and the d-axis component I _ds
I _ds ^* -I _ds) is d-axis component of the current error vector.
Similarly, FIG. 5 shows a case where the induction motor performs a regenerative operation.

As shown in FIG. 4, when the induction motor is in power running operation, the current error vector fluctuates according to the rotation frequency ω of the primary magnetic flux vector, but the fluctuation of the d-axis component ΔI _ds is small and the q-axis component ΔI _qs Changes are the main cause. Also,
1 for both the d-axis component ΔI _ds and the q-axis component ΔI _qs
The sign does not reverse according to the change of the rotation frequency ω of the next magnetic flux vector, and the sign is opposite to the sign of the speed error Δω. This indicates that the signal related to the speed error estimation calculation in the control system is always negative feedback, and acts to stabilize the control system.

On the other hand, as shown in FIG. 5, when the induction motor is in the regenerative operation, the current error vector fluctuates according to the rotation frequency ω of the primary magnetic flux vector, and its d-axis component ΔI _ds and q
The fluctuations of the axis component ΔI _qs are both larger than those in FIG. In addition, there is a place where the sign of the d-axis component ΔI _ds · q-axis component ΔI _qs is inverted according to the change of the rotation frequency ω of the primary magnetic flux vector, and the sign is the same as the sign of the velocity error Δω. The opposite case exists. This indicates that in the control system, the signal relating to the speed error estimating calculation is positive feedback and negative feedback. Positive feedback occurs when the rotation frequency ω of the primary magnetic flux vector is low. The locus of the current error vector passes through the origin, but at this time, no current error occurs due to the speed error.

Here, in the regenerative operation, a signal relating to the speed error estimating operation becomes positive feedback, and the speed error estimating operation diverges, resulting in a problem that the control system becomes unstable. To solve this problem, when estimating the speed error based on the current error vector, d
It is necessary to take some measures regarding the inversion of the signs of the axis component ΔI _ds and the q-axis component ΔI _qs so as not to provide a positive feedback.

Therefore, in the present invention, the current error vector represented on the dq coordinate axis is subjected to coordinate conversion, and the speed error is estimated based on the current error vector after the coordinate conversion. Eliminate points.

Now, from FIGS. 4 and 5, when a certain speed error .DELTA..omega. (= .Omega. ^* -. Omega ^. )> 0 exists, it is desired that the current error vector exists so that the control system is always stable. The area (hereinafter, referred to as “desired area”) is represented as shown in FIG. However, as described above, in the regenerative operation, the current error vector may exist outside this “desired region”. Therefore, as shown in FIG. 7, consider enlarging the “desired region” by coordinate transformation.

In FIG. 7, d-axis and q-axis are coordinate axes before coordinate conversion, and d'-axis and q'-axis are coordinate axes after coordinate conversion. Ψ is a conversion phase angle of coordinate conversion. The current error vector changes according to each operating point such as the rotation frequency and the generated torque of the induction motor. Therefore, the conversion phase angle ψ is determined so that the current error vector is within the “desired region” so that the control system is always kept stable at any operating point within the desired operating range of the induction motor. You. The coordinate transformation of the primary d-axis component command I _ds ^* and q-axis component command of the d-axis component I _ds of the deviation [Delta] I _ds, the primary current I _qs ^* and the deviation [Delta] I _qs q-axis component I _qs currents , The following equation (18
As shown in a) and (18b), they are converted into ΔI ′ _ds and ΔI ′ _qs .

[0078]

(Equation 16)

By performing a speed error estimating operation based on the current error vector after the coordinate conversion, the signal used for the estimating operation is positive feedback at any operating point regardless of powering or regenerative operation. Therefore, the control system can be kept stable.

The first embodiment related to the first and second aspects of the present invention will be described below with reference to the drawings. FIG.
FIG. 1 is a block diagram showing a configuration of a torque control device for an induction motor according to a first embodiment.

In FIG. 1, the current component control circuit 11b,
The speed error estimating means 13 is a novel configuration according to the first embodiment. 1 is an induction motor, 2 is a current detector, 3 is a PWM inverter, 4 is a current component command calculating means, 5 is a current component calculating means, 6 is a slip frequency calculating circuit, and 8 and 10
Is an integrator, 9 is an adder, and 12 is a voltage command operation circuit, which is the same as the configuration in FIG. The current control means includes a current component control circuit 11b, a voltage command calculation circuit 12, and a PWM inverter 3.

FIG. 2 is a block diagram showing a detailed configuration of the current component control circuit 11b. In FIG. 2, reference numeral 56 denotes an adder. In FIG. 2, the same reference numerals as those in FIG.
Indicates the same or corresponding parts.

FIG. 3 is a block diagram showing a detailed configuration of the speed error estimating means 13. In FIG. 3, 60 and 63 are coefficient units, 61, 66, 67, 72 and 75 are subtractors, 6
5 is an adder, 62, 70, 71, 73 and 74 are multipliers, 64 is a divider, 68 is a cosine function unit, and 69 is a sine function unit.

Next, the operation of the torque control device for an induction motor according to the first embodiment will be described. First, the induction motor 1
When the torque command τ _m and the estimated value ω ＾ _m of the rotational frequency of the induction motor 1 output from the first integrator 8 are input to the current component command calculating means 4, the excitation current component command I ₀ ^* and the torque current component The command I _qs ^* is output.

Further, 1 is detected by the current detector 2.
When the secondary currents I _us (U phase), I _vs (V phase) and the phase θ of the primary magnetic flux vector output from the second integrator 10 are input to the current component calculating means 5, the d-axis component of the primary current _Ids and the q-axis component _Iqs are output.

Subsequently, the exciting current component command I ₀ ^* output from the current component command calculating means 4 and the torque current component command I
_qs ^* , d of the primary current output from the current component calculating means 5
When the axis component I _ds , the q-axis component (torque current component) I _qs, and the conversion phase angle に お_{け る in} the coordinate conversion of a preset current error vector are input to the speed error estimating means 13, An error (speed error) Δω between the value ω and the set value (estimated value) ω ^* used in the control calculation and a deviation ΔI ′ _ds of the d′-axis component of the primary current after the coordinate conversion are output.

More specifically, as shown in FIG. 3, the excitation current component command I ₀ ^* and the torque current component command I _qs ^* , the d-axis component I _{ds of the} primary current and the q-axis component (torque current component) I _qs and the converted phase angle ψ are input to the speed error estimating means 13. Then, first, the signal obtained by inputting the d-axis component I _ds of the primary current to the coefficient unit 60 is subtracted from the excitation current component command I ₀ ^* by the subtractor 61, and I ₀ ^* −σ I _ds is output. You. Subsequently, after the q-axis current component (torque current component) I _qs is squared by the multiplier 62 and input to the coefficient unit 63, σI _qs ²
Is output. Further, the σI
After dividing _qs ² by I ₀ ^* −σI _ds output from the subtractor 61 and adding the exciting current component command I ₀ ^* by the adder 65, the primary current d-axis component command I _ds ^* is output. You.
Subsequently, when the d-axis component I _ds is subtracted from the d-axis component command I _ds ^{* of the} primary current by the subtracter 66, the deviation ΔI _ds of the d-axis component of the primary current is output. When the q-axis component I _qs is subtracted from the q-axis component command I _qs ^{* of the} primary current by the subtracter 67, a deviation ΔI _qs of the q-axis component of the primary current is output.

Next, when the converted phase angle ψ is input to the cosine function unit 68, the cosine function value cosψ is output. [Delta] I _ds · cos is outputted further receives the output of the output subtracter 66 of the cosine function 68 to a multiplier 73. When the output of the cosine function unit 68 and the output of the subtractor 67 are input to the multiplier 71, ΔI _qs · cos 出力 is output. Then, ΔI _ds · sinψ is output when an output of the output subtracter 66 of the sine function 69 in multiplier 70. When the output of the sine function unit 69 and the output of the subtracter 67 are input to the multiplier 74, ΔI _qs · sinψ is output.

Subsequently, the multiplier 73 is operated by the subtractor 75.
Is subtracted from the output of the multiplier 74, the deviation ΔI ′ of the d′-axis component of the primary current after the coordinate conversion based on the equation (18a)
_ds is found. When the output of the multiplier 70 is subtracted from the output of the multiplier 71 by the subtractor 72, the deviation ΔI ′ _{qs of the q′-} axis component of the primary current after the coordinate conversion based on the equation (18b) is obtained. Here, the deviation ΔI ′ _qs of the _q′- axis component is defined as an error (speed error) Δω between the actual value ω of the rotation frequency of the primary magnetic flux vector and the set value (estimated value) ω ^* used for the control calculation.

Next, the speed error Δω is calculated by the first integrator 8
, An estimated value of the rotational frequency of the induction motor 1 ω ＾
_m is output. Here, the gain K of the first integrator 8
Is set such that the speed error Δω is sufficiently small even during the acceleration / deceleration operation.

Subsequently, the torque current component command I _qs ^* output from the current component command calculating means 4 and the d-axis component I _ds and the q-axis component I _qs of the primary current output from the current component calculating means 5 are calculated. When input to the slip frequency calculation circuit 6, the slip frequency ω _s is output.

[0092] Subsequently, when the output omega _s of the frequency calculation circuit 6 and slip output omega ^ _m of the first integrator 8 added by the adder 9, (12) operation is performed, the primary flux vector The set value ω ^{* of the} rotation frequency is obtained. Furthermore, when the set value ω ^* of the rotation frequency of the primary magnetic flux vector is input to the second integrator 10, the phase θ of the primary magnetic flux vector is output.

Next, the exciting current component command I ₀ ^* and the torque current component command I _qs output from the current component command calculating means 4
^* , The set value ω ^* of the rotation frequency of the primary magnetic flux vector output from the adder 9, the deviation ΔI ′ _ds of the d′-axis component of the primary current after coordinate conversion output from the speed error estimating means 13,
The d-axis component I _ds and the q-axis component I _qs of the primary current output from the current component calculation means 5 are respectively supplied to the current component control circuit 1.
When input to 1b, a primary voltage d-axis component command V _ds ^* and a q-axis component command V _qs ^* are output.

More specifically, as shown in FIG. 2, a signal obtained by inputting the d-axis component I _ds of the primary current to the coefficient unit 40 is subtracted by the subtractor 41 from the excitation current component command I ₀ ^*. When reduced, I ₀ ^* -σI _ds is output. Then, the multiplier 4
After squaring the q-axis current component (torque current component) I _qs by ² and inputting it to the coefficient unit 43, σI _qs ² is output.
Further, the divider 44 converts this σI _qs ² into a subtractor 4
After dividing by I ₀ ^* −σI _ds output from 1, the adder 5
6, the excitation current component command is added, and a primary current d-axis component command I _ds ^* is output.

Then, the primary current d-axis component command I _ds ^*
Is input to the coefficient unit 46, a d-axis voltage drop due to the primary resistance is output. Further, in order to make the d-axis component of the primary current coincide with the command value, d ′ of the primary current after coordinate conversion is used.
The deviation ΔI ′ _ds of the axis component is amplified by the amplifier 48, and the signal obtained by adding the output of the amplifier 48 and the output of the coefficient unit 46 by the adder 49 is expressed by the following equation (19)
_Is output as the primary voltage d-axis component command V _ds ^* .

[0096]

[Equation 17]

On the other hand, when the excitation current component command is input to the coefficient unit 50, the operation of equation (4) is performed, and the primary magnetic flux d-axis component command Φ _ds ^* is output. Further, when the multiplier 51 multiplies this Φ _ds ^{* by} the set value ω ^* of the rotation frequency of the primary magnetic flux vector, ω ^* · Φ _ds ^* is output. When the torque current component I _qs is input to the coefficient unit 52, a q-axis voltage drop due to the primary resistance is output.

Further, in order to make the torque current component coincide with the command value I _qs , the difference between the torque current component command I _qs ^* and the torque current component I _qs is obtained by the subtractor 53, and this difference is amplified by the amplifier 54. At the same time, the output of the amplifier 54 and the outputs of the multiplier 51 and the coefficient unit 52 are added to an adder 55.
_Is output as a q-axis component command V _qs ^* of the primary voltage represented by the following equation (20).

[0099]

(Equation 18)

Next, the d-axis component command V _ds ^{* of the} primary voltage output from the current component control circuit 11b and the q-axis component command V _{ds
When qs} ^* and the phase θ of the primary magnetic flux vector output from the second integrator 10 are input to the voltage command calculation circuit 12, the primary voltage commands _Vus ^* , V _vs ^* and V _ws ^* are obtained. Is output. Further, the PWM inverter 3 controls the primary voltage of the induction motor 1 so as to match the above-mentioned command.

According to the above operation, I _ds and I _qs are controlled so that I ₀ ^* = I ₀ and I _qs ^* = I _qs are satisfied. Accordingly, the d-axis component Φ _ds of the primary magnetic flux matches the command value, and the set value ω ^* of the rotation frequency of the primary magnetic flux vector output from the adder 9 matches the actual value. Further, the phase θ output from the second integrator 10 matches the phase of the primary magnetic flux vector. As a result, the generated torque τ _m of the induction motor 1 is controlled so as to match the torque command τ ^* _m .

Second Embodiment Next, a second embodiment related to the third and fourth aspects of the present invention will be described with reference to the drawings.

FIG. 8 is a block diagram showing a configuration of a torque control device for an induction motor according to the second embodiment. still,
In the figure, the same reference numerals as those of the related art and FIG. 1 denote the same or corresponding parts, and a description thereof will be omitted.

In FIG. 8, reference numeral 14 denotes a phase angle selection circuit,
5 is a phase angle buffer circuit. Here, the current control means
Current component control circuit 11b, voltage command operation circuit 12 and P
It is composed of a WM inverter 3.

The configuration of the torque control device for the induction motor shown in FIG. 8 is different from the configuration of the torque control device for the induction motor in the first embodiment shown in FIG. 1 in that a phase angle selection circuit 14 and a phase angle buffering circuit 15 are provided. Is added.

FIG. 9 is a block diagram showing a detailed configuration of the phase angle selection circuit 14. As shown in FIG. In FIG. 9, reference numeral 80 denotes a sign determination unit, and 81 denotes a selection switch. FIG. 10 is a block diagram showing a configuration of the phase angle buffering circuit 15. In FIG. 10, reference numeral 82 denotes a first-order delay circuit.

Next, the operation of the torque control device for an induction motor according to the second embodiment will be described. First, the induction motor 1
Torque command tau _m ^* of the induction motor 1 set in advance to input the converted phase angle [psi _R in the case of converting the phase angle [psi _P and the regenerative operation when the force action operation to the phase angle selection circuit 14, [psi _P, [psi
_{One of R} is selected and output as ψ.

More specifically, referring to FIG.
When the _m ^* is input to the code decision unit 80, a separate power running operation and the regenerative operation by tau _m ^* of codes is determined, when the power running operation 1
However, in the case of the regenerative operation, -1 is output.

Subsequently, when the output of the sign determination unit 80 is input to the selection switch 81, the phase angle ψ is output as 位相_P when the input is 1, and ψ _R when the input is −1. next,
When the phase angle buffer circuit 15 inputs the phase angle [psi, was inhibited rate of change of [psi when switching the above [psi _P and [psi _R occurs [psi 'output by the first-order delay circuit 82. This ψ ′ is input to the speed error estimating means 13 and the operation described in the first embodiment is performed.

Embodiment 3. As shown in FIGS. 1 and 2, when a constant speed error Δω (= ω ^* −ω) exists, the absolute amount of the current error caused by the speed error Varies with the rotation frequency of the primary magnetic flux vector. Therefore, the input of the first integrator 8 in FIG. 1 is adjusted according to the rotation frequency of the primary magnetic flux vector.

The third embodiment related to the fifth aspect of the present invention will be described below with reference to the drawings. FIG. 11 is a block diagram showing a configuration of a torque control device for an induction motor according to the present embodiment. In the drawings, the same reference numerals as those in the conventional art and FIG. 1 indicate the same or corresponding parts, and thus the description thereof will be omitted.

In FIG. 11, reference numeral 16 denotes a rotation frequency estimation calculation circuit. Here, the current control means includes a current component control circuit 11b, a voltage command calculation circuit 12, and a PWM inverter 3.

The structure of the torque control device for an induction motor shown in FIG. 11 is different from that of the torque control device for an induction motor of the first embodiment shown in FIG. This is the one using the arithmetic circuit 16.

FIG. 12 is a block diagram showing a detailed configuration of the rotation frequency estimation calculation circuit 16. As shown in FIG. In FIG. 12, 9
0 is an absolute value circuit, 91 is a function generator, 92 is a multiplier, 9
3 is an integrator.

Next, the operation of the torque control device for an induction motor according to the third embodiment will be described. First, when the set value ω ^* of the rotation frequency of the primary magnetic flux vector output from the adder 9 is input to the absolute value circuit 90, the absolute value | ω ^* | is output. Further, when this absolute value | ω ^* | is input to the function generator 91, an adjustment coefficient (gain) for adjusting the speed error Δω according to | ω ^* | is output. Further, when the adjustment coefficient (gain) and the speed error Δω are multiplied by the multiplier 92 and further input to the integrator 93, the induction motor 1
Is output as the estimated value ω ＾ _m of the rotation frequency of Here, the gain K of the integrator 93 is set such that the speed error Δω is sufficiently small even during the acceleration / deceleration operation. As described above, the same control response can be obtained over a wide range of operation.

Embodiment 4 In the first to third embodiments, the slip frequency ω _{s is} calculated using the q-axis component (torque current component) I _qs of the primary current output from the current component calculation means 5.
Were subjected to calculation of, I _qs by the first integrator 8 is controlled so as to match the current component command calculation unit 4 is output from the torque current component command I _qs ^*. Thus may be performed operations to Beri frequency omega _s with I _qs ^* instead of I _qs.

Embodiment 5 FIG. In the first to fourth embodiments, the d-axis component command I _ds ^{* of the} primary current is used by the current component control circuit 11b for the d-axis voltage drop due to the primary resistance ^.
Was calculated using the q-axis component I _qs of the primary current, but the d-axis component I _ds of the primary current was converted to I _ds ^* and I _qs was calculated by the current control means. Control is performed so as to match the q-axis component command I _qs ^* of the primary current. Thus, the I _ds instead of I _ds ^*, by using the I _qs ^* instead of I _qs, may be calculated each voltage drop.

Embodiment 6 FIG. Note that the configuration described in the third embodiment can be used in combination with the configuration described in the second embodiment. Further, the hardware configured in the above embodiment may be realized by software processing using a microcomputer.

[0119]

As described above, according to the first to fourth aspects of the present invention, the deviation between the command of the exciting current component of the primary current having the same phase as the primary magnetic flux vector of the induction motor and the actual value, and A coordinate conversion is performed on a current error vector expressed from a command of a torque current component of a primary current orthogonal to the primary magnetic flux vector and a deviation of an actual value, and the deviation of the torque current component after the coordinate conversion is used. By configuring to estimate the rotational frequency of the induction motor used for control, a wide range of operation including the low-speed rotation-regenerative operation range, which was conventionally unstable and could not be controlled, was performed without using a current detector. There is an effect that a stable, high-accuracy, high-speed response torque control characteristic can be obtained over the entire range.

According to the fifth aspect of the present invention, in the configuration for performing the estimation calculation of the rotation frequency of the induction motor used for the control, the speed error (the rotation frequency of the primary magnetic flux vector of the induction motor, By adjusting the difference between the actual value and the set value used for the control calculation with the set value of the rotation frequency of the primary magnetic flux vector, there is an effect that uniform control response can be obtained regardless of the rotation frequency.

[Brief description of the drawings]

FIG. 1 is a block diagram showing an overall configuration of a torque control device for an induction motor according to Embodiment 1 of the present invention.

FIG. 2 is a block diagram showing a configuration of a current component control circuit according to the first embodiment of the present invention.

FIG. 3 is a block diagram illustrating a configuration of a speed error estimating unit according to the first embodiment of the present invention.

FIG. 4 is an explanatory diagram of an operation principle according to the first embodiment of the present invention.

FIG. 5 is an explanatory diagram of an operation principle according to the first embodiment of the present invention.

FIG. 6 is an explanatory diagram of an operation principle according to the first embodiment of the present invention.

FIG. 7 is an explanatory diagram of an operation principle according to the first embodiment of the present invention.

FIG. 8 is a block diagram showing an overall configuration of a torque control device for an induction motor according to a second embodiment of the present invention.

FIG. 9 is a block diagram showing a configuration of a phase angle selection circuit according to a second embodiment of the present invention.

FIG. 10 is a block diagram showing a configuration of a phase angle buffer circuit according to Embodiment 2 of the present invention.

FIG. 11 is a block diagram showing an overall configuration of a torque control device for an induction motor according to Embodiment 3 of the present invention.

FIG. 12 is a block diagram illustrating a configuration of a rotation frequency estimation calculation circuit according to Embodiment 3 of the present invention;

FIG. 13 is a block diagram showing an overall configuration of a conventional torque control device for an induction motor.

FIG. 14 is a block diagram showing a configuration of a current component command calculating means of a conventional torque control device for an induction motor.

FIG. 15 is a block diagram showing a configuration of a slip frequency calculation circuit of a conventional torque control device for an induction motor.

FIG. 16 is a block diagram showing a configuration of a current component control circuit of a conventional torque control device for an induction motor.

[Explanation of symbols]

REFERENCE SIGNS LIST 1 induction motor, 2 current detector, 3 PWM inverter, 4 current component command calculation means, 5 current component calculation means, 6 slip frequency calculation circuit, 7, 35, 41, 4
5, 47, 53, 61, 66, 67, 72, 75 subtractor, 8 first integrator, 9, 33, 49, 55, 56,
65 adder, 10 second integrator, 11b current component control circuit, 12 voltage command operation circuit, 13 speed error estimating means, 14 phase angle selection circuit, 15 phase angle buffer circuit, 2
0, 90 absolute value circuit, 21, 91 function generator, 2
2, 24, 30, 32, 34, 36, 40, 43, 4
6, 50, 52, 60, 63 coefficient units, 23, 37, 4
4, 64 divider, 31 differentiator, 42, 51, 62,
70, 71, 73, 74, 92 multiplier, 48, 54 amplifier, 68 cosine function unit, 69 sine function unit, 80 sign decision unit, 81 selection switch, 82 primary delay circuit,
93 Integrator.

Claims (5)

[Claims] An induction motor, a command signal for torque to be output from the induction motor, an excitation current component command having the same phase as a primary magnetic flux vector of the induction motor, and a torque orthogonal to the primary magnetic flux vector. A current component command calculating means for calculating a current component command; a current detector for detecting a primary current of the induction motor; an excitation current component of the induction motor based on the phases of the primary current and the primary magnetic flux vector. Current component calculating means for calculating the torque current component and the torque current component command, the torque current component, the excitation current component command, and the excitation current component,
Error between the actual value of the rotation frequency of the secondary magnetic flux vector and the set value of the rotation frequency of the primary magnetic flux vector used in the control calculation
Speed error estimating means for calculating (speed error); a first integrator for integrating the speed error output from the speed error estimating means to obtain a rotation frequency of the induction motor; and a current component command calculating means. A slip frequency calculation circuit for calculating the slip frequency of the induction motor based on the output of the induction motor and the output of the current component calculation means; and a signal obtained by adding the rotation frequency of the induction motor and the slip frequency. An adder for setting the rotation frequency of the primary magnetic flux vector to a set value of the rotation frequency of the primary magnetic flux vector; and a second signal for forming a signal obtained by integrating the set value of the rotation frequency of the primary magnetic flux vector to the phase of the primary magnetic flux vector.
And an electric current control means for controlling a primary current of the induction motor so that the exciting current component and the torque current component follow the exciting current component command and the torque current component command, respectively. A torque control device for an induction motor, characterized in that: 2. The method according to claim 1, wherein the speed error estimating means coordinates a current error vector on a rectangular coordinate system represented by a deviation between the torque current component command and the torque current component and a deviation between the excitation current component command and the excitation current component. And output the deviation between the excitation current component command and the excitation current component after the speed error and coordinate conversion, and the current control means outputs the excitation current component after the coordinate conversion output from the speed error estimation means. 2. The torque control device for an induction motor according to claim 1, wherein a deviation between the command and the exciting current component is input. 3. The speed error estimating means includes a phase angle selection circuit that switches a conversion phase angle when performing coordinate conversion on the current error vector by a powering operation and a regenerative operation of the induction motor. The torque control device for an induction motor according to claim 2, wherein: 4. A phase angle buffer circuit, which receives the converted phase angle output from the phase angle selection circuit and suppresses the degree of change of the phase angle when the phase angle is switched and outputs the phase angle buffer circuit. The torque control device for an induction motor according to claim 3. 5. The torque control device for an induction motor according to claim 1, wherein the speed error outputted by the speed error estimating means is different from the speed error.
A rotation frequency estimation calculation circuit for inputting a set value of a rotation frequency of the next magnetic flux vector, adjusting the speed error according to the rotation frequency, integrating the adjusted speed error and forming a rotation frequency of the induction motor; A torque control device for an induction motor.