JP3530721B2 - Induction motor torque control device - Google Patents

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a torque control device for an induction motor, and particularly to a speed detector and a voltage (magnetic flux).
The present invention relates to a torque control device for an induction motor, which is suitable for performing high-accuracy and high-speed response torque control over a wide range of operation regions without using a detector, regardless of high-speed / low-speed rotation speed and power running / regenerative operation. Is.

[0002]

2. Description of the Related Art As a conventional torque control device for an induction motor which does not use a speed detector and a voltage (magnetic flux) detector, there is a structure shown in FIG. 13 disclosed in Japanese Patent Laid-Open No. 9-9700.

The torque control device for an induction motor shown in FIG.
Induction motor 1, current detector 2, PWM inverter 3, current component command calculation means 4, current component calculation means 5, slip frequency calculation circuit 6, subtractor 7, integrators 8, 10, adder 9, current component control circuit 11a and a voltage command calculation circuit 12, and the current control means includes a current component control circuit 11a, a voltage command calculation circuit 12 and a PWM inverter 3.
It consists of This control device uses a primary current command to be controlled as a reference based on the primary magnetic flux of the induction motor, so that the primary current command at the time of high-speed rotation or overload is
The fluctuation range of the next voltage is suppressed.

FIG. 14 is a block diagram showing a detailed structure of the current component command calculating means 4. In FIG. 14, 20 is an absolute value circuit, 21 is a function generator, 22 and 24 are coefficient multipliers,
23 is a divider.

FIG. 15 is a block diagram showing a detailed structure of the slip frequency calculation circuit 6. In FIG. 15, 30,
32, 34 and 36 are coefficient units, 31 is a differentiator, 33 is an adder, 35 is a subtractor, and 37 is a divider.

FIG. 16 is a block diagram showing a detailed structure of the current component control circuit 11a. In FIG. 16, 40,
43, 46, 50 and 52 are coefficient units, 41, 45, 47
And 53 are subtractors, 42 and 51 are multipliers, 44 is a divider, 48 and 54 are amplifiers, and 49 and 55 are adders.

Next, before explaining the operation of the torque control device for the induction motor, the control principle of this control device will be described.

First, a rotating coordinate axis that rotates at a primary frequency
The voltage-current equation of the induction motor on the (d-q axes) is represented by the following equations (1a) to (1d) as is well known.

[0009]

[Equation 1]

However, in the above equation (1), V _ds and V _qs
Are the d-axis and q-axis components of the primary voltage, I _ds and I _{qs, respectively.}
Are the d-axis and q-axis components of the primary current, I _dr and I _{qr, respectively.}
Are the d-axis and q-axis components of the secondary current, respectively. Also,
R _s is primary resistance, R _r is secondary resistance, L _s is primary self-inductance, L _r is secondary self-inductance, M is primary 2
Next mutual inductance, p is differential operator (= d / dt)
Is. Furthermore, ω is the rotation frequency of the primary magnetic flux vector,
That is, it is the primary frequency and ω _s is the slip frequency.

Next, the d-axis component and the q-axis component of the primary magnetic flux are given by the following equations (2a) and (2b), respectively.

[0012]

[Equation 2]

Now, let the direction of the primary magnetic flux vector be d
When the axes are matched with each other, the generated torque τ _m is given by the following expression (3).

[0014]

[Equation 3]

However, P _m is the number of pole pairs of the induction motor.

The generated torque τ _m is the d-axis component Φ _{ds of the} primary magnetic flux.
And is proportional to the product of the q-axis component of the primary current I _qs , which facilitates control. Therefore, the d-axis component Φ _ds of the primary magnetic flux is made to coincide with the command value Φ _ds ^* given by equation (4), and the direction of the primary magnetic flux vector is made to coincide with the d-axis (q-axis component of the primary magnetic flux). Φ
_qs = 0), the slip frequency ω _s
As in the equation (5), the d-axis component I _ds of the primary current is given as in the equation (6).

[0017]

[Equation 4]

However, I ₀ ^* is an exciting current component command, and σ is a leakage coefficient {= 1-M ² / (L _s L _r )}.

[0019] From (6), the exciting current component I ₀ corresponding to the exciting current component command I ₀ ^* is expressed by the following equation.

[0020]

[Equation 5]

Therefore, if the exciting current component I ₀ is controlled so as to match the command value I ₀ ^* , the d-axis component Φ _ds of the primary magnetic flux can be matched with the command value Φ _ds ^*. The direction of the secondary magnetic flux vector can be matched with the d axis.

Here, the primary frequency ω and the slip frequency ω _s
As is well known, there is a relation as shown in the following equation (8).

[0023]

[Equation 6]

However, ω _m is the rotation frequency (rotation speed) of the induction motor.

Thus, the direction of the primary magnetic flux vector is changed to d
If they are aligned with the axis, in steady state equation (1a) and (1
The equation (b) becomes the following equations (9a) and (9b).

[0026]

[Equation 7]

Therefore, command values V _ds ^* and V _qs ^* of the d-axis and q-axis components of the primary voltage are given by the following equations (10a) and (10b).

[0028]

[Equation 8]

However, ω ^* is a set value (estimated value) of the primary magnetic flux vector used for the control calculation.

From the equation (6), the d-axis component I of the primary current I
_The d-axis component command I _ds ^* corresponding to _ds is given by the following equation (11).

[0031]

[Equation 9]

Here, in order to set the set value ω ^* of the primary magnetic flux vector from the equation (8), the value of the rotation frequency ω _m of the induction motor is required. However, this control method is a control method that does not use a speed detector, and it is necessary to calculate the rotation frequency of the induction motor as an estimated value ω ^ _m of the rotation frequency of the induction motor by some alternative method. Since the primary magnetic flux vector is obtained using ω ^ _m , equation (8) is given by the following equation (12)
It replaces like.

[0033]

[Equation 10]

This estimated value ω of the rotation frequency of the induction motor
The calculation principle of ^ _m is as follows.

Now, if control is performed using a PWM inverter so that the primary voltage of the induction motor matches the command value, ω
^{If *} = ω, then I _qs ^* = I _qs and I _ds ^* = I _ds . However, if ω ^* ≠ ω, then I _qs ^* ≠ I _qs and I _ds ^* ≠
I _ds . Therefore, I _qs ^* = I _qs and I _ds ^* = I _ds
By obtaining the holds such ω ^* ω ^* = ω, and the estimated value of the rotation frequency of the induction motor can be matched to the actual value.

Therefore, the q-axis component command I _qs ^{* of the} primary current
And the q-axis component I _qs are integrated, and the integration result is an estimated value ω ^ _m of the rotation frequency of the induction motor, I _qs ^* = I _qs and ω ^* = ω are established in a steady state. At that time, the estimated value ω ^ _m matches the actual value ω _m .

Now, the d-axis component I _ds and the q-axis component I _qs of the primary current can be obtained using the following equations (13a) and (13b).

[0038]

[Equation 11]

However, θ is the phase of the primary magnetic flux vector, and is obtained by integrating the primary frequency ω, that is, the rotation frequency of the primary magnetic flux vector.

The primary voltage commands V _us ^* , V _vs ^* and V _ws
^* Can be obtained using the following equation (14).

[0041]

[Equation 12]

Next, a specific operation of the torque control device for the induction motor shown in FIG. 13 will be described. First, when the torque command τ ^* _m of the induction motor 1 and the estimated value ω ^ _m of the rotation frequency of the induction motor 1 output from the first integrator 8 are input to the current component command calculation means 4, the exciting current component command I ₀ ^* and the torque current component command I _qs ^* are output.

Specifically, as shown in FIG. 14, an absolute value circuit 20 estimates an estimated value ω of the rotation frequency of the induction motor 1.
^ Absolute value of _m | _ω ^ _m | is output. Further, when this absolute value | ω ^ _m | is input to the function generator 21, the following equation (1
The calculation of 5) is performed, and the exciting current component command I ₀ ^* is output.

[0044]

[Equation 13]

Here, I _0-100% is the rated value of the exciting current component,
ω _b is the rated frequency of the induction motor 1.

Next, when the exciting current component command I ₀ ^* is input to the coefficient unit 22, the calculation of the equation (4) is performed and the d-axis component command Φ _ds ^{* of the} primary magnetic flux is obtained. Further, the divider 23
By dividing the torque command τ ^* _m of the induction motor 1 input from the outside by the d-axis component command Φ _ds ^* of the primary magnetic flux, and inputting it to the coefficient unit 24, the torque is calculated from the relationship of equation (3). The current component command I _qs ^* is output.

Next, 1 detected by the current detector 2
When the secondary current I _us (U phase), I _vs (V phase) and the phase θ of the primary magnetic flux vector output from the second integrator 10 are input to the current component computing means 5, the computation of equation (13) is performed. The d-axis component I _ds and the q-axis component I _{qs of the} primary current are output.

Subsequently, the deviation of the torque current component command I _qs ^* output from the current component command calculating means 4 and the q-axis component (torque current component) I _qs of the primary current output from the current component calculating means 5 is calculated. This difference is calculated by the subtractor 7
When input to the integrator 8, the estimated value ω ^ _m of the rotation frequency of the induction motor 1 is output.

Next, the exciting current component command I ₀ ^* output from the current component command calculating means 4 and the d-axis component I _ds and the q-axis component I _qs of the primary current output from the current component calculating means 5 are shown. When input to the slip frequency calculation circuit 6, the slip frequency ω _s
Is output.

Specifically, as shown in FIG. 15, the coefficient units 30, 32, the differentiator 31, and the adder 33 are used to (5)
The numerator on the right side of the expression is calculated. On the other hand, the coefficient unit 34,
The denominator on the right side of the equation (5) is calculated by the 36 and the subtractor 35. Therefore, the adder 3 is added by the divider 37.
When the output of 3 is divided by the output of the coefficient unit 36, the calculation of the equation (5) is performed, and the slip frequency ω _s is obtained.

Subsequently, when the output of the first integrator 8 and the output of the slip frequency operation circuit 6 are added by the adder 9, the operation of the expression (12) is performed, and the set value of the rotation frequency of the primary magnetic flux vector is obtained. ω ^* is required. Further, the set value ω ^* of the rotation frequency of the primary magnetic flux vector is set to the second integrator 1
When input to 0, the phase θ of the primary magnetic flux vector is output.

Next, the exciting current component command I ₀ ^* and the torque current component command I _qs output from the current component command calculating means 4 are outputted.
^* , The set value ω ^* of the rotation frequency of the primary magnetic flux vector output from the adder 9, and the d-axis component I _ds and the q-axis component I _qs of the primary current output from the current component calculating means 5, respectively.
When input to the current component control circuit 11a, the d-axis component command V _ds ^* and the q-axis component command V _qs ^{* of the} primary voltage are output.

Specifically, as shown in FIG. 16, a signal obtained by inputting the d-axis component I _ds of the primary current to the coefficient unit 40 is converted by the subtracter 41 from the exciting current component command I ₀ ^*. When it is reduced, (I ₀ ^* −σI _ds ) is output. Next, multiplier 4
When the q-axis current component (torque current component) I _qs is squared by ^2, and input to the coefficient unit 43, σI _qs ² is output.
Further, the divider 44 subtracts this σI _qs ² from the subtractor 4
After dividing by (I ₀ ^* −σI _ds ) output from 1, the subtractor 45 subtracts from the d-axis component I _{ds of the} primary current.
Expression (7) is calculated and the exciting current component I ₀ is output.

Subsequently, when the d-axis component I _ds of the primary current is input to the coefficient unit 46, the d-axis voltage drop due to the primary resistance is output. Further, in order to match the d-axis component of the primary current with the command value, the subtractor 47 causes the exciting current component command I ₀ ^* and the exciting current component I ₀ output from the subtracter 45 to be output.
Is obtained, the output of the amplifier 48 and the output of the coefficient unit 46 are added by an adder 49, and the obtained signal is the primary voltage expressed by the following equation (16). Output as the d-axis component command V _ds ^* .

[0055]

[Equation 14]

On the other hand, the exciting current component command I ₀ ^* is calculated by the coefficient unit 50.
When input to, the equation (4) is calculated and the d-axis component command Φ _ds ^{* of the} primary magnetic flux vector is output. Furthermore, this Φ
_{When ds} ^* and the setting value ω ^* of the rotation frequency of the primary magnetic flux vector are multiplied by the multiplier 51, ω ^* · Φ _ds ^* is output.
When the torque current component I _qs is input to the coefficient unit 52,
The voltage drop on the q-axis due to the primary resistance is output.

Further, in order to match the torque current component I _qs with the command value, the subtracter 53 obtains the deviation between the torque current component command I _qs ^* and the torque current component I _qs , and the amplifier 54 amplifies this deviation. At the same time, the output of the amplifier 54 and the outputs of the multiplier 51 and the coefficient unit 52 are added to the adder 55.
And the obtained signal is added by 1
Output as the q-axis component command V _qs ^* of the next voltage.

[0058]

[Equation 15]

Next, the d-axis component command V _ds ^* and the q-axis component command V _qs ^{* of the} primary voltage output from the current component control circuit 11a ^.
And the phase θ of the primary magnetic flux vector output from the second integrator 10 is input to the voltage command calculation circuit 12, (14)
The equation is calculated, and the primary voltage commands V _us ^* , V _vs ^* and V _ws
^* Is output. Further, the PWM inverter 3 controls the primary voltage of the induction motor 1 to match the above command.

By the above operation, I ₀ ^* = I ₀ and I _qs ^* =
I _ds and I _qs are controlled so that I _qs holds. Therefore, the d-axis component Φ _ds of the primary magnetic flux matches the command value, and the set value ω ^* of the primary magnetic flux vector output from the adder 9 is ω ^* = ω, which matches the actual value. Furthermore, the second
The phase θ output from the integrator 10 of 1 corresponds to the phase of the primary magnetic flux vector. As a result, the generated torque τ _m of the induction motor 1 is controlled so as to match the torque command τ ^* _m .

[0061]

In the torque control device for an induction motor shown in the above-mentioned prior art, it becomes possible to perform torque control without using a speed detector and a voltage (magnetic flux) detector. However, the above control device has a problem in that the control system becomes unstable and cannot be controlled when a regenerative operation is performed when the induction motor is rotating at a low speed.

That is, when the regenerative operation is performed, the sign of the slip frequency ω _s shown in the equation (12) has the opposite sign to the sign of the estimated value ω ^ _m of the rotation frequency of the induction motor. Especially when trying to perform regenerative operation at low speed rotation, {ω _S nearly equal
ω ^ _m }, the absolute value of the setting value ω ^* of the rotation frequency of the primary magnetic flux vector becomes {| ω ^* | nearly equal 0} as shown in the equation (12). At this time, the sign of the deviation between the q-axis component command I _qs ^{* of the} primary current and the q-axis component I _qs has the opposite sign to the sign of the deviation when the control is normally performed in the power running operation. That is, obtains an estimated value of the rotation frequency of the induction motor to match the I _qs to I _qs ^*, (1
0b), even if the q-axis component command V _qs ^* of the primary voltage is obtained by the control and then controlled, it does not converge in the direction of I _qs = I _qs ^*, and conversely I _qs ≠ I _qs ^* Diversify in the direction.
Therefore, since it cannot be controlled so that ω ^* = ω,
The primary frequency ω cannot be calculated normally and cannot be controlled. From the above, in the conventional control device, it was necessary to limit the operation region of the induction motor to a region excluding the low speed regenerative operation so as not to be out of control as described above.

The present invention has been made in order to solve the above problems, and particularly in a wide operating range including low speed rotation and regenerative operation without using a speed detector and a voltage (magnetic flux) detector. In general, it is an object of the present invention to provide a torque control device for an induction motor that realizes stable torque control performance with high accuracy and high speed response.

[0064]

According to another aspect of the present invention, there is provided a torque control device for an induction motor, wherein the primary component of the induction motor is in-phase with the primary magnetic flux vector by the current component calculating means.
(Exciting current component) and orthogonal component (torque current component), and the primary current is calculated based on the command signal of the inductive torque output by the induction motor and the estimated value of the rotation frequency of the induction motor by the current component command calculation means. Of the excitation current component command and the torque current component command of
The primary current of the induction motor is controlled so that the current components match the respective current component commands, and the torque error component command and the torque current component and the excitation current component command are controlled by the speed error estimating means. An error (speed error) between the actual value of the rotational frequency of the primary magnetic flux vector and the set value of the rotational frequency of the primary magnetic flux vector used in the control calculation is calculated based on the exciting current component, and the speed error is calculated. Is integrated by a first integrator to obtain an estimated value of the rotation frequency of the induction motor to obtain the rotation frequency of the induction motor, and a slip frequency calculation circuit is used to calculate the excitation current component command or the excitation current component. And the slip frequency of the induction motor from the torque current component command or the torque current component, and the addition of the adder to estimate the rotation frequency of the induction motor. Values and the induction motor rotational frequency and without the addition to the primary flux vector slip frequency,
Furthermore, the rotational frequency of the primary magnetic flux vector is integrated by the second integrator to obtain the phase of the primary magnetic flux vector, and the phase of the primary magnetic flux vector is calculated to obtain a high-speed response and It is possible to obtain highly stable torque control characteristics.

In the torque control device for an induction motor according to a second aspect of the present invention, the speed error estimating means according to the first aspect is arranged such that the deviation between the torque current component command and the torque current component, the exciting current component command and the exciting current. The current error vector on the Cartesian coordinates represented by the deviation of the current component is subjected to coordinate conversion to obtain the actual value of the rotation frequency of the primary magnetic flux vector and the set value of the rotational frequency of the primary magnetic flux vector used for the control calculation. Error (speed error) and the deviation of the exciting current component command after the coordinate conversion and the exciting current, and further, the current control means controls the primary current by inputting the deviation to input the low speed regeneration. The control system is always kept stable over a wide range of operation including operation, and high-speed response torque control characteristics are obtained.

In the torque control device for an induction motor according to a third aspect of the present invention, the actual value of the rotation frequency of the primary magnetic flux vector performed by the speed error estimating means and the rotational frequency of the primary magnetic flux vector used for the control calculation. Of the error (speed error) with respect to the set value of, the current error on the Cartesian coordinates expressed by the deviation of the torque current component command and the torque current component and the deviation of the exciting current component command and the exciting current component When performing vector coordinate conversion, the conversion phase angle of the above coordinate conversion is switched by the power running / regenerative operation of the induction motor, so that stable, high-accuracy, high-speed operation is always possible over a wide range of operation including low-speed regenerative operation. The response torque control characteristic is obtained.

According to a fourth aspect of the present invention, there is provided the torque control device for an induction motor according to the fourth aspect, wherein the conversion phase angle of the coordinate conversion performed by the speed error estimating means is switched.
By performing through the buffer circuit, the degree of change in the conversion phase angle at the time of switching is suppressed, and the vibration and shock transiently generated at the time of switching are alleviated.

According to a fifth aspect of the present invention, there is provided the torque control device for an induction motor according to the first aspect, wherein the speed error output from the speed error estimating means is determined according to the absolute value of the rotation frequency of the primary magnetic flux vector. By adjusting, it is possible to obtain equal control response over a wide range of operation areas.

[0069]

DETAILED DESCRIPTION OF THE INVENTION

First Embodiment Prior to describing the embodiments relating to the inventions of claims 1 and 2, a method of estimating a velocity error, which is the basis of the present invention, will be described below.

As described in the control method in the prior art, the q-axis component command I _qs ^* and q-axis component I _qs of the primary current and the d-axis component command I _ds ^* and d-axis component I _{ds of the} primary current.
With respect to, if the setting value ω ^* of the rotation frequency of the primary magnetic flux vector that satisfies I _qs ^* = I _qs and I _ds ^* = I _ds is obtained, then ω ^* = ω and the actual value of the rotational frequency of the primary magnetic flux vector is obtained. ω can be calculated. However, the deviation of the I _qs ^* and I _qs deviation and I _ds ^* and I _ds can be speed error is constant, 1
It varies depending on the actual value ω of the rotation frequency of the secondary magnetic flux vector and the state of the generated torque τ _m . This is shown in FIGS. 4 and 5.

FIG. 4 shows that the induction motor is in a power running operation (rated output torque τ _m100% ) and has a certain speed error Δω (= ω ^*).
-Ω)> 0 exists, the error generated in the primary current due to the velocity error is expressed as a current error vector on the dq coordinate axes while changing the rotation frequency ω of the primary magnetic flux vector. It is a thing. That is, the deviation ΔI _qs (= I _qs ^* − between the q-axis component command I _qs ^{* of the} primary current and the q-axis component I _qs
I _qs ) is the q-axis component of the current error vector, and the deviation ΔI _ds (= d-axis component command I _ds ^* and d-axis component I _ds of the primary current is ΔI _ds (=
I _ds ^* -I _ds ) is the d-axis component of the current error vector.
Similarly, FIG. 5 shows the case where the induction motor is in the regenerative operation.

As shown in FIG. 4, when the induction motor is in the powering operation, the current error vector fluctuates according to the rotation frequency ω of the primary magnetic flux vector, but the fluctuation of the d-axis component ΔI _ds is small and the q-axis component ΔI _qs. The fluctuation of is dominant. Also,
1 for both the d-axis component ΔI _ds and the q-axis component ΔI _qs
The sign does not reverse according to the change of the rotation frequency ω of the next magnetic flux vector, and the sign is the opposite of the sign of the speed error Δω. This indicates that in the control system, the signal relating to the speed error estimation calculation is always negative feedback, and the control system acts to be stable.

On the other hand, as shown in FIG. 5, when the induction motor is in the regenerative operation, the current error vector fluctuates according to the rotation frequency ω of the primary magnetic flux vector, and its d-axis component ΔI _ds fluctuates and q.
The fluctuations of the axial component ΔI _qs are both larger than those in the case of FIG. In addition, there is a portion where the signs of the d-axis component ΔI _ds and the q-axis component ΔI _qs are inverted according to the change of the rotation frequency ω of the primary magnetic flux vector, and the sign is the same as the case of the velocity error Δω. The reverse case exists. This indicates that in the control system, the signals involved in the speed error estimation calculation are both positive feedback and negative feedback. Positive feedback occurs when the rotation frequency ω of the primary magnetic flux vector is low. Further, the locus of the current error vector passes through the origin, but at this time, the current error due to the speed error does not occur.

Here, in the regenerative operation, the signal related to the speed error estimation calculation becomes a positive feedback, and the speed error estimation calculation diverges, which makes the control system unstable. To solve this problem, when the velocity error is estimated and calculated based on the current error vector, d of the current error vector is calculated.
It is necessary to take some measures to invert the signs of the axial component ΔI _ds and the q-axis component ΔI _qs so that positive feedback is not obtained.

Therefore, in the present invention, the current error vector represented on the dq coordinate axes is coordinate-converted, and the speed error is estimated and calculated based on the current error vector after the coordinate conversion. Eliminate points.

From FIGS. 4 and 5, it is desired that the current error vector exists so that the control system is always stable when a certain speed error Δω (= ω ^* −ω)> 0 exists. The area (hereinafter referred to as “desired area”) is represented as shown in FIG. However, as mentioned above, in regenerative operation, the current error vector may lie outside this "desired region". Therefore, as shown in FIG. 7, consider enlarging the "desired region" by coordinate conversion.

In FIG. 7, the d-axis and the q-axis are the coordinate axes before the coordinate conversion, and the d'axis and the q'axis are the coordinate axes after the coordinate conversion. Further, ψ is a conversion phase angle of coordinate conversion. The current error vector changes according to each operating point such as the rotation frequency of the induction motor and the generated torque. Therefore, the conversion phase angle ψ is determined so that the current error vector exists within the “desired region” so that the control system is always stable at any operating point within the desired operating range of the induction motor. It The coordinate transformation of the primary d-axis component command I _ds ^* and q-axis component command of the d-axis component I _ds of the deviation [Delta] I _ds, the primary current I _qs ^* and the deviation [Delta] I _qs q-axis component I _qs currents , The following equation (18
As shown in a) and (18b), they are converted into ΔI ′ _ds and ΔI ′ _qs .

[0078]

[Equation 16]

By performing an estimation calculation of the velocity error based on the current error vector after the coordinate conversion, the signal used for the estimation calculation is not positive feedback at any operating point regardless of power running or regenerative operation. Therefore, the control system can be kept stable.

The first embodiment relating to the inventions of claims 1 and 2 will be described below with reference to the drawings. Figure 1
FIG. 3 is a block diagram showing a configuration of a torque control device for an induction motor according to the first embodiment.

In FIG. 1, the current component control circuit 11b,
The speed error estimating means 13 has a novel configuration according to the first embodiment. In addition, 1 is an induction motor, 2 is a current detector, 3 is a PWM inverter, 4 is a current component command calculating means, 5 is a current component calculating means, 6 is a slip frequency calculating circuit, and 8 and 10.
Is an integrator, 9 is an adder, and 12 is a voltage command calculation circuit, which has the same configuration as in FIG. The current control means is composed of a current component control circuit 11b, a voltage command calculation circuit 12 and a PWM inverter 3.

FIG. 2 is a block diagram showing a detailed structure of the current component control circuit 11b. In FIG. 2, reference numeral 56 is an adder. Incidentally, in FIG. 2, the same reference numerals as those in FIG.
Indicates the same or corresponding part.

FIG. 3 is a block diagram showing a detailed structure of the speed error estimating means 13. In FIG. 3, 60 and 63 are coefficient units, 61, 66, 67, 72 and 75 are subtractors, and 6
5 is an adder, 62, 70, 71, 73 and 74 are multipliers, 64 is a divider, 68 is a cosine function unit, and 69 is a sine function unit.

Next, the operation of the torque control device for the induction motor of the first embodiment will be described. First, the induction motor 1
Torque command τ _m and the estimated value ω ^ _m of the rotation frequency of the induction motor 1 output from the first integrator 8 are input to the current component command calculation means 4, the exciting current component command I ₀ ^* and the torque current component The command I _qs ^* is output.

Further, 1 detected by the current detector 2
When the secondary current I _us (U phase), I _vs (V phase) and the phase θ of the primary magnetic flux vector output from the second integrator 10 are input to the current component computing means 5, the d-axis component of the primary current is input. The I _ds and the q-axis component I _qs are output.

Subsequently, the exciting current component command I ₀ ^* and the torque current component command I output from the current component command calculating means 4 are outputted.
_qs ^* , d of the primary current output from the current component calculation means 5
When the axis phase I _ds , the q axis component (torque current component) I _qs, and the conversion phase angle ψ in the coordinate conversion of the preset current error vector are input to the speed error estimating means 13, the rotation frequency of the primary magnetic flux vector is actually calculated. The error (speed error) Δω between the value ω and the set value (estimated value) ω ^* used for the control calculation and the deviation ΔI ′ _ds of the d ′ axis component of the primary current after coordinate conversion are output.

Specifically, as shown in FIG. 3, the exciting current component command I ₀ ^* and the torque current component command I _qs ^* , the d-axis component I _{ds of the} primary current and the q-axis component (torque current component) I _{The qs} and the conversion phase angle ψ are input to the velocity error estimating means 13. Then, first, the signal obtained by inputting the d-axis component I _ds of the primary current into the coefficient unit 60 is subtracted from the exciting current component command I ₀ ^* by the subtractor 61 to output I ₀ ^* -σI _ds. It Subsequently, after multiplying the q-axis current component (torque current component) I _qs by the multiplier 62 and inputting it to the coefficient unit 63, σI _qs ²
Is output. Further, this σI is calculated by the divider 64.
_{When qs} ² is divided by I ₀ ^* −σI _ds output from the subtractor 61 and the exciting current component command I ₀ ^* is added by the adder 65, the d-axis component command I _ds ^{* of the} primary current is output. It
Then, by the subtractor 66, when reducing the d-axis component I _ds from the primary current of the d-axis component command I _ds ^*, the deviation [Delta] I _ds of the d-axis component of the primary current is output. Also, the primary current of the q-axis component command I _qs ^* When reducing the q-axis component I _qs by subtracter 67, the deviation [Delta] I _qs q-axis component of the primary current is output.

Next, when the conversion phase angle ψ is input to the cosine function unit 68, the cosine function value cos ψ is output. Further, when the output of the cosine function unit 68 and the output of the subtractor 66 are input to the multiplier 73, ΔI _ds · cos ψ is output. When the output of the cosine function unit 68 and the output of the subtractor 67 are input to the multiplier 71, ΔI _qs · cos ψ is output. Next, when the output of the sine function unit 69 and the output of the subtractor 66 are input to the multiplier 70, ΔI _ds · sin ψ is output. When the output of the sine function unit 69 and the output of the subtractor 67 are input to the multiplier 74, ΔI _qs · sin ψ is output.

Subsequently, the subtractor 75 causes the multiplier 73
When the output of the multiplier 74 is subtracted from the output of, the deviation ΔI ′ of the d ′ axis component of the primary current after coordinate conversion based on the equation (18a).
_ds is obtained. Further, when the subtracter 72 subtracts the output of the multiplier 70 from the output of the multiplier 71, the deviation ΔI ′ _{qs of the q′-} axis component of the primary current after coordinate conversion based on the equation (18b) is obtained. Here, the deviation ΔI ′ _qs of the _q′- axis component is defined as an error (speed error) Δω between the actual value ω of the rotation frequency of the primary magnetic flux vector and the set value (estimated value) ω ^* used for the control calculation.

Next, the velocity error Δω is calculated by the first integrator 8
Input to the estimated value ω ^ of the rotation frequency of the induction motor 1.
_m is output. Here, the gain K of the first integrator 8
Is set such that the speed error Δω is sufficiently small even during the acceleration / deceleration operation.

Subsequently, the torque current component command I _qs ^* output from the current component command calculating means 4 and the d-axis component I _ds and the q-axis component I _qs of the primary current output from the current component calculating means 5 are shown. When input to the slip frequency calculation circuit 6, the slip frequency ω _s is output.

Subsequently, when the output ω ^ _{m of} the first integrator 8 and the output ω _s of the slip frequency computing circuit 6 are added by the adder 9, the equation (12) is calculated and the primary magnetic flux vector The set value ω ^{* of the} rotation frequency is obtained. Further, when the set value ω ^* of the rotation frequency of the primary magnetic flux vector is input to the second integrator 10, the phase θ of the primary magnetic flux vector is output.

Next, the exciting current component command I ₀ ^* and the torque current component command I _qs output from the current component command calculating means 4 are outputted.
^* , The setting value ω ^* of the rotation frequency of the primary magnetic flux vector output from the adder 9, the deviation ΔI ' _ds of the d'axis component of the coordinate-converted primary current output from the speed error estimating means 13,
The d-axis component I _ds and the q-axis component I _qs of the primary current output from the current component calculation means 5 are respectively supplied to the current component control circuit 1
When input to 1b, the d-axis component command V _ds ^* and the q-axis component command V _qs ^{* of the} primary voltage are output.

Specifically, as shown in FIG. 2, a signal obtained by inputting the d-axis component I _ds of the primary current into the coefficient unit 40 is converted by the subtracter 41 from the exciting current component command I ₀ ^*. When it is decreased, I ₀ ^* −σI _ds is output. Next, multiplier 4
When the q-axis current component (torque current component) I _qs is squared by ^2, and input to the coefficient unit 43, σI _qs ² is output.
Further, the divider 44 subtracts this σI _qs ² from the subtractor 4
After being divided by I ₀ ^* −σI _ds output from 1, the adder 5
6, the excitation current component command is added, and the d-axis component command I _ds ^{* of the} primary current is output.

Subsequently, the d-axis component command I _ds ^{* of the} primary current is given ^.
Is input to the coefficient unit 46, the voltage drop on the d-axis due to the primary resistance is output. Further, in order to match the d-axis component of the primary current with the command value, d ′ of the primary current after coordinate conversion is changed.
The deviation ΔI ′ _ds of the axial component is amplified by the amplifier 48, and the signal obtained by adding the output of the amplifier 48 and the output of the coefficient unit 46 by the adder 49 is expressed by the following equation (19).
It is output as the d-axis component command V _ds ^* of the primary voltage shown by.

[0096]

[Equation 17]

On the other hand, when the exciting current component command is input to the coefficient unit 50, the equation (4) is calculated and the d-axis component command Φ _ds ^{* of the} primary magnetic flux is output. Further, when this Φ _ds ^* and the set value ω ^* of the rotation frequency of the primary magnetic flux vector are multiplied by the multiplier 51, ω ^* · Φ _ds ^* is output. When the torque current component I _qs is input to the coefficient unit 52, the q-axis voltage drop due to the primary resistance is output.

Further, in order to match the torque current component with the command value I _qs , the subtracter 53 obtains the deviation between the torque current component command I _qs ^* and the torque current component I _qs , and the amplifier 54 amplifies this deviation. At the same time, the output of the amplifier 54 and the outputs of the multiplier 51 and the coefficient unit 52 are added to the adder 55.
The signal obtained by the addition is output as the q-axis component command V _qs ^* of the primary voltage represented by the following equation (20).

[0099]

[Equation 18]

Next, the d-axis component command V _ds ^{* of the} primary voltage output from the current component control circuit 11 b and the q-axis component command V
_{When qs} ^* and the phase θ of the primary magnetic flux vector output from the second integrator 10 are input to the voltage command calculation circuit 12, the primary voltage commands V _us ^* , V _vs ^* and V _ws ^* are obtained. Is output. Further, the PWM inverter 3 controls the primary voltage of the induction motor 1 to match the above command.

According to the above operation, I _ds and I _qs are controlled so that I ₀ ^* = I ₀ and I _qs ^* = I _qs . Therefore, the d-axis component Φ _ds of the primary magnetic flux matches the command value, and the set value ω ^* of the rotation frequency of the primary magnetic flux vector output from the adder 9 matches its actual value. Furthermore, the phase θ output from the second integrator 10 matches the phase of the primary magnetic flux vector. As a result, the generated torque τ _m of the induction motor 1 is controlled so as to match the torque command τ ^* _m .

Second Embodiment Next, a second embodiment relating to the inventions of claims 3 and 4 will be described with reference to the drawings.

FIG. 8 is a block diagram showing the structure of a torque control device for an induction motor according to the second embodiment. still,
In the figure, the same reference numerals as those of the conventional one and FIG.

In FIG. 8, 14 is a phase angle selection circuit, 1
Reference numeral 5 is a phase angle buffer circuit. Here, the current control means
Current component control circuit 11b, voltage command calculation circuit 12 and P
It is composed of a WM inverter 3.

The configuration of the torque controller for the induction motor shown in FIG. 8 is the same as that of the torque controller for the induction motor in the first embodiment shown in FIG. Is added.

FIG. 9 is a block diagram showing a detailed structure of the phase angle selection circuit 14. In FIG. 9, reference numeral 80 is a code determiner, and 81 is a selection switch. FIG. 10 is a block diagram showing the configuration of the phase angle buffer circuit 15. In FIG. 10, reference numeral 82 is a first-order delay circuit.

Next, the operation of the torque control device for the induction motor of the second embodiment will be described. First, the induction motor 1
Of the torque command τ _m ^* , the preset conversion phase angle ψ _P when the induction motor 1 is in the power running operation and the preset conversion phase angle ψ _R when the induction motor 1 is in the regenerative operation, are input to the phase angle selection circuit 14, and ψ _P , ψ
_{One of R} is selected and output as ψ.

Specifically, in FIG. 9, the torque command τ
_{When m} ^* is input to the sign determiner 80, the sign of τ _m ^* is used to determine whether the vehicle is a powering operation or a regenerative operation.
However, in the case of regenerative operation, -1 is output.

Then, when the output of the sign determiner 80 is input to the selection switch 81, ψ _P is output as the phase angle ψ when the input is 1, and ψ _R is output when the input is -1. next,
When the phase angle ψ is input to the phase angle buffer circuit 15, the first-order delay circuit 82 suppresses the change rate of ψ when the above-mentioned switching between ψ _P and ψ _R occurs, but outputs ψ ′. This ψ ′ is input to the speed error estimation means 13 and the operation described in the first embodiment is performed.

Third Embodiment As shown in FIGS. 1 and 2, when a constant speed error Δω (= ω ^* −ω) exists, the absolute amount of the current error caused by the speed error. Varies depending on the rotation frequency of the primary magnetic flux vector. Therefore, the input of the first integrator 8 in FIG. 1 is adjusted according to the rotation frequency of the primary magnetic flux vector.

The third embodiment relating to the invention of claim 5 will be described below with reference to the drawings. FIG. 11 is a block diagram showing the configuration of the torque control device for an induction motor according to the present embodiment. Incidentally, in the figure, the same reference numerals as those in the conventional one and FIG.

In FIG. 11, reference numeral 16 is a rotation frequency estimation arithmetic circuit. Here, the current control means is composed of a current component control circuit 11b, a voltage command calculation circuit 12 and a PWM inverter 3.

The configuration of the torque controller for the induction motor shown in FIG. 11 is the same as the configuration of the torque controller for the induction motor of the first embodiment shown in FIG. The arithmetic circuit 16 is used.

FIG. 12 is a block diagram showing a detailed structure of the rotation frequency estimation calculation circuit 16. In FIG. 12, 9
0 is an absolute value circuit, 91 is a function generator, 92 is a multiplier, 9
3 is an integrator.

Next, the operation of the torque control device for the induction motor of the third embodiment will be described. First, when the set value ω ^* of the rotation frequency of the primary magnetic flux vector output from the adder 9 is input to the absolute value circuit 90, the absolute value | ω ^* | is output. Further, when this absolute value | ω ^* | is input to the function generator 91, an adjustment coefficient (gain) for adjusting the speed error Δω according to | ω ^* | is output. Further, when the adjustment coefficient (gain) and the speed error Δω are multiplied by the multiplier 92 and further input to the integrator 93, the induction motor 1
The estimated value ω ^ _m of the rotation frequency of is output. Here, the gain K of the integrator 93 is set so that the speed error Δω is sufficiently small even during the acceleration / deceleration operation. As described above, it is possible to obtain equal control responsiveness over a wide range of operation regions.

Fourth Embodiment In the first to third embodiments described above, the slip frequency ω _{s is} calculated by using the q-axis component (torque current component) I _qs of the primary current output from the current component calculation means 5.
However, I _qs is controlled by the first integrator 8 so as to match the torque current component command I _qs ^* output from the current component command calculating means 4. Thus may be performed operations to Beri frequency omega _s with I _qs ^* instead of I _qs.

Embodiment 5. In the first to fourth embodiments described above, in the current component control circuit 11b, the d-axis component command I _ds ^{* of the} primary current is used for the d-axis voltage drop due to the primary resistance ^.
The q-axis voltage drop is calculated by using the q-axis component I _qs of the primary current, and the d-axis component I _ds of the primary current is I _ds ^* and I _qs is calculated by the current control means. It is controlled so as to match the q-axis component command I _qs ^* of the primary current. Thus, the I _ds instead of I _ds ^*, by using the I _qs ^* instead of I _qs, may be calculated each voltage drop.

Sixth Embodiment The configuration described in the third embodiment can be used in combination with the configuration described in the second embodiment. In addition, those configured by hardware in the above embodiments may be realized by software processing using a microcomputer.

[0119]

As described above, according to the inventions of claims 1 to 4, the deviation between the command and the actual value of the exciting current component of the primary current, which is in phase with the primary magnetic flux vector of the induction motor, and Coordinate conversion is performed on the current error vector represented by the command of the torque current component of the primary current orthogonal to the primary magnetic flux vector and the deviation of the actual value, and the deviation of the torque current component after coordinate conversion is used. By configuring to estimate the rotation frequency of the induction motor used for control, a wide range of operations including low-speed rotation-regenerative operation range that was previously unstable and uncontrollable without using a current detector There is an effect that torque control characteristics that are always stable, highly accurate, and have high-speed response can be obtained over the region.

According to the fifth aspect of the present invention, in the configuration for performing the estimation calculation of the rotation frequency of the induction motor used for control, the speed error (the rotation frequency of the primary magnetic flux vector of the induction motor By adjusting the deviation between the actual value and the set value used for the control calculation with the set value of the rotation frequency of the primary magnetic flux vector, there is an effect that uniform control response can be obtained regardless of the rotation frequency.

[Brief description of drawings]

FIG. 1 is a block diagram showing an overall configuration of a torque control device for an induction motor according to a first embodiment of the present invention.

FIG. 2 is a block diagram showing a configuration of a current component control circuit according to the first embodiment of the present invention.

FIG. 3 is a block diagram showing a configuration of speed error estimating means according to the first embodiment of the present invention.

FIG. 4 is an explanatory diagram of the operating principle of the first embodiment of the present invention.

FIG. 5 is an explanatory diagram of the operating principle of the first embodiment of the present invention.

FIG. 6 is an explanatory diagram of the operating principle of the first embodiment of the present invention.

FIG. 7 is an explanatory diagram of the operating principle of the first embodiment of the present invention.

FIG. 8 is a block diagram showing an overall configuration of a torque control device for an induction motor according to a second embodiment of the present invention.

FIG. 9 is a block diagram showing a configuration of a phase angle selection circuit according to a second embodiment of the present invention.

FIG. 10 is a block diagram showing a configuration of a phase angle buffer circuit according to a second embodiment of the present invention.

FIG. 11 is a block diagram showing an overall configuration of a torque control device for an induction motor according to a third embodiment of the present invention.

FIG. 12 is a block diagram showing a configuration of a rotation frequency estimation calculation circuit according to a third embodiment of the present invention.

FIG. 13 is a block diagram showing the overall configuration of a conventional torque control device for an induction motor.

FIG. 14 is a block diagram showing a configuration of a current component command calculation means of a conventional induction motor torque control device.

FIG. 15 is a block diagram showing a configuration of a slip frequency calculation circuit of a conventional induction motor torque control device.

FIG. 16 is a block diagram showing a configuration of a current component control circuit of a conventional induction motor torque control device.

[Explanation of symbols]

1 induction motor, 2 current detector, 3 PWM inverter, 4 current component command calculating means, 5 current component calculating means, 6 slip frequency calculating circuit, 7, 35, 41, 4
5, 47, 53, 61, 66, 67, 72, 75 Subtractor, 8 First integrator, 9, 33, 49, 55, 56,
65 adder, 10 2nd integrator, 11b current component control circuit, 12 voltage command calculation circuit, 13 speed error estimation means, 14 phase angle selection circuit, 15 phase angle buffer circuit, 2
0, 90 absolute value circuit, 21, 91 function generator, 2
2, 24, 30, 32, 34, 36, 40, 43, 4
6, 50, 52, 60, 63 coefficient units, 23, 37, 4
4, 64 divider, 31 differentiator, 42, 51, 62,
70, 71, 73, 74, 92 multiplier, 48, 54 amplifier, 68 cosine function unit, 69 sine function unit, 80 sign judging unit, 81 selection switch, 82 first-order delay circuit,
93 Integrator.

─────────────────────────────────────────────────── ─── Continuation of front page (58) Fields surveyed (Int.Cl. ⁷ , DB name) H02P 7/63 H02P 21/00

Claims (5)

(57) [Claims] 1. An induction motor and a command signal of a slip torque output from the induction motor are input, and an exciting current component command in phase with the primary magnetic flux vector of the induction motor and a torque orthogonal to the primary magnetic flux vector. A current component command calculating means for calculating a current component command, a current detector for detecting a primary current of the induction motor, and an exciting current component of the induction motor based on the phases of the primary current and the primary magnetic flux vector. And a current component calculating means for calculating a torque current component, the torque current component command, the torque current component, the exciting current component command, and the exciting current component are input, and the above 1
Error between the actual value of the rotational frequency of the secondary magnetic flux vector and the set value of the rotational frequency of the primary magnetic flux vector used in the control calculation
A speed error estimating means for calculating (speed error); a first integrator for integrating the speed error output from the speed error estimating means to obtain a rotation frequency of the induction motor; and a current component command calculating means. And a signal obtained by adding the rotation frequency of the induction motor and the slip frequency to the slip frequency of the induction motor based on the output of the current component calculation means. And a signal obtained by integrating the set value of the rotation frequency of the primary magnetic flux vector as the phase of the primary magnetic flux vector
And an electric current control means for controlling the primary current of the induction motor so that the exciting current component and the torque current component follow the exciting current component command and the torque current component command, respectively. A torque control device for an induction motor. 2. The velocity error estimating means coordinates a current error vector on a rectangular coordinate system represented by the deviation between the torque current component command and the torque current component and the deviation between the exciting current component command and the exciting current component. Converting the speed error and outputting the deviation between the exciting current component command after coordinate conversion and the exciting current component, the current control means outputs the exciting current component after coordinate conversion output by the speed error estimating means. The torque control device for an induction motor according to claim 1, wherein a deviation between the command and the exciting current component is input. 3. The speed error estimating means is provided with a phase angle selection circuit that switches a conversion phase angle when performing coordinate conversion on the current error vector by a power running operation and a regenerative operation of the induction motor. A torque control device for an induction motor according to claim 2. 4. A phase angle buffer circuit for inputting the conversion phase angle output from the phase angle selection circuit, and suppressing and outputting the degree of change in the phase angle when switching the phase angle. The torque control device for an induction motor according to claim 3. 5. The torque control device for an induction motor according to claim 1, wherein the speed error output by the speed error estimating means and
A rotation frequency estimation arithmetic circuit is provided, which inputs the set value of the rotation frequency of the next magnetic flux vector, adjusts the speed error according to the rotation frequency, and integrates the adjusted speed error to obtain the rotation frequency of the induction motor. A torque control device for an induction motor.