JPH11318094A - Motor - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a motor with reduced power loss and vibration. SOLUTION: An operation controller 51 PWM-controls a switching transistor 61 of a voltage converter 52 and variably controls the converted DC voltage of the voltage converter 52. First power amplifiers 11-13 and second power amplifiers 15-17 supply three-phase coils 2-4 with the DC converted voltage. First field effect type of power transistors 81-83 and second field effect type of power transistors 85-87 perform the switching action of a current path, and reduces the pulsation of the supply current to the coils.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a motor in which a current path is electronically switched by a plurality of transistors.

[0002]

2. Description of the Related Art In recent years, motors for electronically switching a current path using a plurality of transistors have been widely used as driving motors for OA equipment and AV equipment. Examples of such a motor include a PNP type power transistor and an NP
There is a motor that switches a current path to a coil using an N-type power transistor. FIG. 32 shows a conventional motor,
The operation will be described. The rotor 2011 has a field portion made of a permanent magnet, and in response to the rotation of the rotor 2011, the position detector 2041 outputs two sets of three-phase voltage signals K1,
K2, K3 and K4, K5, K6 are generated. The first distributor 2042 generates three-phase lower energization control signals L1, L2, L3 in response to the voltage signals K1, K2, K3, and generates lower NPN-type power transistors 2021, 2022, 2
023 is controlled. The second distributor 2043 generates three-phase upper energization control signals M1, M2, and M3 in response to the voltage signals K4, K5, and K6, and controls energization of the upper PNP power transistors 2025, 2026, and 2027. Thereby, the coils 2012, 2013, 20
14 is supplied with a three-phase drive voltage.

[0003]

However, this conventional motor has the following various problems. (1) Power loss is large. In the conventional configuration, the NPN power transistor 202
1, 2022, 2023 and PNP type power transistors 2025, 2026, 2027
The voltage between the collectors is controlled in an analog
2, 2013 and 2014. For this reason, the residual voltage of each power transistor is large, and a large power loss occurs due to the product of the residual voltage and the current flowing through the power transistor. Especially,
Since the driving current to the motor coil was large, the power loss was extremely large. Therefore, the power efficiency of the motor was extremely poor. (2) High cost. To reduce the cost, one transistor or resistor
It is effective to combine them on a chip integrated circuit (IC). However, the PNP type power transistors 2025 and 2
In order to form 026 and 2027, a large chip area is required, which has been a major factor in causing an increase in cost. In addition, it was difficult to operate the PNP-type power transistor at high speed due to the influence of the parasitic capacitance when integrated. In addition, the power loss of the power transistor
Heat generation was large, and it was difficult to form an integrated circuit. In particular, since the driving current to the motor coil is large, there is a great risk that the integrated circuit will be thermally destroyed by the heat generated by the power transistor. Further, when a heat radiating plate was attached to prevent thermal destruction, the cost increased significantly. (3) The vibration of the motor is large. 2. Description of the Related Art In recent years, in an optical disk device and a magnetic disk device, a motor having small vibration has been demanded in accordance with high-density recording and reproduction of a disk. However, in the conventional configuration, a spike voltage is generated in the coil in accordance with the steep switching of the power transistor, and pulsation of the drive current is generated. As a result, the generated driving force pulsates, causing large motor vibration. It was an important issue to realize a motor that solved these problems individually or simultaneously. The purpose of the present invention is
An object of the present invention is to solve the various problems described above individually or simultaneously, to provide a motor having a configuration suitable for integration into an integrated circuit, and to solve the problems in the prior art.

[0004]

According to the present invention, there is provided a motor comprising: a moving body; a plurality of phase coils; and a field-effect switching transistor for switching a power supply path of a DC power supply at a high frequency. A voltage supply unit that outputs a converted DC voltage obtained by converting a voltage, and a first field-effect power transistor that forms a current path from a negative output terminal side of the voltage supply unit to one of the coils of the plurality of phases. Q (Q is an integer of 2 or more) first power amplifying means, and a second power path forming a current path from the positive output terminal side of the voltage supply means to one of the multi-phase coils.
Q second power amplifying means each including a field effect type power transistor, switching creating means for obtaining a switching signal of a plurality of phases, and the Q first power amplifying means in response to an output signal of the switching creating means. A first distribution control unit that performs distribution control of energization from the power amplification unit and causes at least one of the Q first field-effect power transistors to perform a resistive voltage drop operation; In response to the output signal, distribution of the energization from the Q second power amplifying means is controlled, and at least one of the Q second field-effect power transistors is caused to perform a resistive voltage drop operation. Second distribution control means, and operation control means for changing the converted DC voltage between the positive output terminal side and the negative output terminal side of the voltage supply means in synchronization with the moving operation of the moving body. .

[0005] The above-described resistive voltage drop operation refers to a full-on state of the field-effect power transistor. The operation state of the field-effect power transistor includes three states: a full-on state, a half-on state,
There is an off state. Here, in the half-on state, the field-effect power transistor performs the amplification operation of the active region. In addition, the field-effect power transistor is in an active state or an active state during the full-on state and the half-on state. With the above configuration, the power loss of the first field-effect power transistor, the second field-effect power transistor, and the field-effect switching transistor is significantly reduced, and the power efficiency of the motor is significantly improved. Therefore, power elements such as a field-effect switching transistor and a field-effect power transistor can be formed at a high density in a single-chip integrated circuit together with required semiconductor elements, and an inexpensive motor can be realized. Further, by the operation control means and the voltage supply means, the change DC voltage of the voltage supply means is changed in synchronization with the moving operation of the moving body, so that the pulsation of the generated driving force due to the influence of the back electromotive force waveform of the coil can be reduced. Was. Also, for example, the first power amplifying means or the second power amplifying means
By supplying the smoothly changing first Q-phase current signal and the second Q-phase current signal to the power supply control terminal side of the power amplifying means, a smoothly changing driving current in both directions can be supplied to the coil. In addition, the pulsation of the driving current can be greatly reduced.
As a result, the pulsation of the generated driving force is reduced, and a high-performance motor can be realized.

According to another aspect of the present invention, there is provided a motor comprising a moving body, a single-phase or multi-phase coil, voltage supply means for supplying a DC voltage, and a voltage supply means for supplying the DC voltage from a negative output terminal side of the voltage supply means. Q (Q is an integer of 2 or more) first power amplifying means each including a first field-effect power transistor forming a current path to one of a single-phase or multi-phase coil; and the voltage supply means Q second power-supply transistors each including a second field-effect power transistor forming a current path from the positive output terminal side to one of the single-phase or multi-phase coils.
Power amplifying means, and a switching creating means for obtaining a switching signal,
In response to the output signal of the switching creation means, the Q first
Distribution control means for distributing and controlling energization from the power amplifying means, and causing at least one of the Q first field-effect power transistors to perform a resistive voltage drop operation; In response to the output signal of the second power amplifying means, and controls distribution of the energization from the Q second power amplifying means so that at least one of the Q second field effect power transistors operates in a resistive voltage drop operation. A second distribution control means for causing the voltage supply means to include a conversion inductor means for storing magnetic energy; a conversion capacitor means for storing electric energy;
A current outgoing terminal is connected to a negative terminal of the DC power supply, and a field effect switching transistor having a current inflow terminal connected to one end of the converting inductor is provided. Switching means for high-frequency switching of a power supply path for replenishing energy; and off-on operation complementary to on-off high-frequency switching operation of the field-effect switching transistor to perform conversion from the conversion inductor means to the conversion capacitor means. Current path forming means for forming a current path to the circuit side including: a converted DC voltage is output between one end of the conversion capacitor means and one end of the DC power supply, and the converted DC voltage is Q The first power amplifying means and the Q second power amplifying means. Integrated circuit means for forming the field-effect switching transistor, the first field-effect power transistor, and the second field-effect power transistor together with required semiconductor elements in a single-chip integrated circuit. Having.

As a result, the power loss of the first field-effect power transistor and the second field-effect power transistor is greatly reduced. Further, the power loss of the field effect switching transistor of the voltage supply means is small. As a result, the power efficiency of the motor has been significantly improved. Further, even if these power elements are integrated into a single chip, the heat generation is extremely small, and the temperature of the integrated circuit does not rise excessively. Further, with the above configuration, the operation of the parasitic transistor element formed with the integration of the integrated circuit is prevented. That is, even if the field-effect switching transistor performs a high-frequency on / off switching operation, a malfunction due to the parasitic transistor does not occur. Therefore, power elements such as a field-effect switching transistor and a field-effect power transistor can be formed at a high density in a single-chip integrated circuit together with required semiconductor elements, and an inexpensive motor can be realized. Also, for example, a first Q-phase current signal or a second Q-phase current signal that smoothly changes to the energization control terminal side of the Q first power amplifying means or the Q second power amplifying means. By supplying the coil, a smoothly changing drive current in both directions can be supplied to the coil,
Drive current pulsation has been greatly reduced. Further, since the first field-effect power transistor and the second field-effect power transistor perform a smooth current path switching operation, malfunction of a parasitic transistor of the integrated circuit due to the current path switching can be prevented. As a result, the pulsation of the generated driving force is reduced, and a high-performance motor can be realized.

According to another aspect of the present invention, there is provided a motor including a moving body, a multi-phase coil, voltage supply means for supplying a DC voltage, and the multi-phase coil from a negative output terminal side of the voltage supply means. Q (Q is an integer of 2 or more) first power amplifying means each including a first field-effect power transistor forming a current path to one of the power supply means, and a positive output terminal side of the voltage supply means. Q second power amplifying means each including a second field-effect power transistor forming a current path to one of the coils of the plurality of phases, a switching creating means for obtaining a switching signal of a plurality of phases, In response to the output signal of the switching generating means, the energization of the stages from the Q first power amplifiers is distributed and controlled, and at least one of the Q first field effect power transistors is connected to a resistor. Voltage drop operation In response to the output signal of the first distribution control means and the output signal of the switching generation means, distribution control of energization from the Q number of second power amplifying means is performed to control the Q number of the second field effect type power transistors. Second distribution control means for causing at least one of them to perform a resistive voltage drop operation, and turning off or on a current path between a positive output terminal side of the voltage supply means and a common connection terminal side of the multi-phase coils. A shunt switch means having a shunt transistor to be turned on, a current-carrying means having a current-carrying transistor for performing or stopping current supply from the Q second power amplifying means to the coils of the plurality of phases; Diode means capable of conducting current in one direction from a current outflow terminal side to a current inflow terminal side of at least one field effect type power transistor of the field effect type power transistor; To Bei.

Thus, the first energizing mode in which a bidirectional current is supplied to the coil to obtain a large generated force, and the second energizing mode in which a unidirectional current is supplied to the coil to perform high-speed rotation are provided in a timely manner. A motor that can be switched to operate has been realized. In particular, the diode means connected from the current outflow terminal side to the current inflow terminal side of the second field-effect power transistor tries to flow a backflow current by the back electromotive force generated in the coil in the second conduction mode. However, backflow is prevented by the energization transistor of the energization stop means,
Normal operation is being performed. In addition, for example, when the first field-effect power transistor, the second field-effect power transistor, the shunt transistor, and the conduction transistor are integrated, many parasitic elements (diode means) formed in the integrated circuit are formed. ), And the first and second energization modes are stably operated. Further, for example, a first Q-phase current signal or a second Q-phase current signal that smoothly changes to the energization control terminal side of the Q first power amplifying means or the Q second power amplifying means. By supplying the driving current, a smoothly changing bidirectional or unidirectional driving current can be supplied to the coil, so that the pulsation of the driving current can be significantly reduced. As a result, a high-performance motor with little pulsation of the generated driving force can be realized.

According to another aspect of the present invention, there is provided a motor including a moving body, a multi-phase coil, a voltage supply means for supplying a DC voltage, and a multi-phase coil from a negative output terminal side of the voltage supply means. NMO forming a current path to one of the
Q (Q is an integer of 2 or more) first power amplifying units each including an S-type power transistor, and a current path from a positive output terminal side of the voltage supply unit to one of the coils of the plurality of phases. Q second power amplifying means each including a second PMOS type power transistor to be formed, switching generating means for obtaining a switching signal of a plurality of phases, and the Q number of power generating means in response to an output signal of the switching generating means. First distribution control means for controlling the distribution of the power supply from the first power amplifying means and causing at least one of the Q first power transistors to perform a resistive voltage drop operation; In response to the output signal, distribution control of energization from the Q second power amplifying means is performed, and at least one of the Q second power transistors is operated in a resistive voltage drop operation. Distribution system Means for connecting a power path from a positive terminal of the DC power supply to a current inflow end of the Q second power amplifying means when the DC power supply of the voltage supply means is turned on; A PMOS power path switch for interrupting a power path between a positive terminal of the DC power supply and a current inflow terminal of the Q power amplification means when the DC power supply of the voltage supply means is turned off. Power path switching means having a transistor; and voltage extracting means for extracting a rectified DC voltage obtained by rectifying the back electromotive force of the coil when the DC power supply is turned off.

As a result, a motor capable of extracting a rectified DC voltage when the DC power supply is turned off has been realized. further,
In order to operate this motor, a configuration was required in which a high potential point higher than the output DC voltage of the DC power supply was not required. As a result, in an emergency state where the DC power supply is turned off, it is possible to execute a required emergency evacuation process using the rectified DC voltage, and to obtain a high potential point including such an emergency state. The power loss due to the circuit configuration was eliminated. In addition, an increase in the number of components (particularly, a capacitor) due to a circuit configuration for obtaining a high potential point is eliminated. Therefore, a low-cost motor can be realized by integration into an integrated circuit. These and other configurations and operations will be described in detail in the description of the embodiments.

[0012]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Several preferred embodiments of the present invention will be described below in detail with reference to the accompanying FIGS.

Embodiment 1 FIGS. 1 to 8 show a motor according to Embodiment 1 of the present invention. FIG. 1 shows the overall configuration. Moving object 1
Is, for example, a rotor provided with a field portion that generates a plurality of poles of field magnetic flux by a magnetic flux generated by a permanent magnet.
Here, the field portion of the moving body 1 is shown as a two-pole magnetized permanent magnet. However, in a modified example, the moving body 1 may have multiple poles, or may be configured with many magnetic pole pieces. Three-phase coils 2, 3,
Numeral 4 is electrically displaced by 120 degrees and is disposed on a stator as a fixed body. Three-phase coils 2, 3, and 4 are 3
A three-phase magnetic flux is generated by the phase driving currents I1, I2, and I3, and a driving force is generated by interaction with the field portion of the moving body 1, thereby giving the driving force to the moving body 1. The disk 1b is attached to the moving body 1 and rotates together with the moving body 1.

The voltage converter 52 has an NMOS-type switching transistor 61 for performing a high-frequency switching operation of about 200 kHz. Here, the NMOS-type switching transistor is an N-channel MOS structure field-effect switching transistor. NM
The OS type switching transistor 61 has its current outflow terminal side connected to the negative terminal side (-) of the DC power supply 50,
The current inflow terminal side and one end of the conversion inductor element 63 are connected, and a power supply path for replenishing magnetic energy from the positive terminal side (+) of the DC power supply 50 to the conversion inductor element 63 is subjected to high-frequency switching (on / off). Working).

The flywheel diode 62 connected to one end of the conversion inductor element 63 performs an on / off operation complementary to the on / off high-frequency switching operation of the NMOS type switching transistor 61. A current path forming circuit to the circuit side including the conversion capacitor element 64 is configured. This allows
The flywheel diode 62 forms a current path for supplying current to the circuit including the conversion capacitor element 64 through the conversion inductor element 63 when the NMOS switching transistor 61 is off. That is, NM
When the OS type switching transistor 61 is on,
From the positive terminal side of the DC power supply 50, the conversion inductor element 6
3 is formed to replenish the magnetic energy of the conversion inductor element 63 (increase the magnetic energy of the conversion inductor element 63).

NMOS type switching transistor 61
Is turned off, the terminal voltage of the conversion inductor element 63 rapidly increases, and the flywheel diode 6
2 is changed to the conductive state, and the current path forming circuit from the conversion inductor element 63 to the circuit including the conversion capacitor element 64 operates (reduces the magnetic energy of the conversion inductor element 63). As a result, a converted DC voltage (Vcc-Vg) is output between one end of the conversion capacitor element 64 and one end of the DC power supply 50. Conversion capacitor element 6
Reference numeral 4 denotes a filter circuit that is connected between the positive output terminal side (P) and the negative output terminal side (M) of the voltage converter 52 and smoothes the current and voltage supplied through the conversion inductor element 63. doing. Thus, the potential Vg on the negative output terminal side of the voltage converter 52 is variably controlled by performing the high-frequency PWM operation (pulse width modulation operation) on the NMOS switching transistor 61. As a result, the output DC voltage Vcc supplied from the DC power supply 50 is used as a power supply source to generate a converted DC voltage (Vcc-Vg) between the positive output terminal side and the negative output terminal side of the voltage converter 52. Here, the negative terminal of the DC power supply 50 is set to the ground potential (0 V). The DC power supply 50 and the voltage converter 52 form a voltage supply block for supplying a required DC voltage.

The NMOS switching transistor 61 is formed of, for example, an N-channel field effect transistor having a double diffusion MOS structure, and has a switching diode formed as a parasitic element from the current outflow terminal side to the current inflow terminal side. 61d are connected in reverse and connected in an equivalent circuit (the integrated circuit may be formed so as not to form such a parasitic switching diode 61d).

On the negative output terminal side of the voltage converter 52, 3
The current outflow terminal sides of the first power amplifiers 11, 12, 13 are commonly connected. The first power amplifier 11 has a first NMOS power transistor 81, and amplifies the output current F1 of the first current amplifier 41 input to the conduction control terminal side in a predetermined manner and outputs the result. Here, the NMOS power transistor is an N-channel MOS structure field-effect power transistor.

First NMOS type power transistor 81
And the NMOS transistor 91 constitute a first NMOS power section current mirror circuit. Where N
The MOS-type power section current mirror circuit is a field-effect power section current mirror circuit using an N-channel MOS structure field-effect power transistor.

The cell size of the NMOS power transistor 81 is set to one of the cell size of the NMOS transistor 91.
00, and a current amplification factor of 100 is obtained when operating in the active region. The first NMOS power transistor 81 is formed of a field effect transistor having a double-diffused N-channel MOS structure, and has a first power source formed as a parasitic element from the current outflow terminal side to the current inflow terminal side. It has a diode 81d connected in reverse, and is connected in an equivalent circuit.

Similarly, the first power amplifier 12 has a first NMOS power transistor 82, and the output current F2 of the first current amplifier 42 input to the conduction control terminal side.
Is amplified and output. The first NMOS-type power transistor 82 and the NMOS-type transistor 92 constitute a first NMOS-type power section current mirror circuit.
The cell size of the OS type power transistor 82 is set to NMOS
The cell size of the type transistor 92 is set to 100 times.

The first NMOS power transistor 82 is constituted by a field effect transistor having a double-diffused N-channel MOS structure, and is formed as a parasitic element from the current outflow terminal side to the current inflow terminal side. One power diode 82d is connected in reverse, and is connected in an equivalent circuit.

Similarly, the first power amplifier 13 has a first NMOS power transistor 83, and the output current F3 of the first current amplifier 43 input to the conduction control terminal side.
Is amplified and output. The first NMOS type power transistor 83 and the NMOS type transistor 93 constitute a first NMOS type power section current mirror circuit.
The cell size of the OS type power transistor 83 is set to NMOS
The cell size of the type transistor 93 is set to 100 times. The first NMOS power transistor 83 is formed of a field effect transistor having a double-diffused N-channel MOS structure, and has a first power source formed as a parasitic element from the current outflow terminal side to the current inflow terminal side. It has a diode 83d connected in reverse, and is connected in an equivalent circuit.

First NMOS power transistor 8
The current outflow terminals 1, 82, and 83 are commonly connected to the negative output terminal of the voltage converter 52, and the current inflow terminals are connected to the power supply terminals of the coils 2, 3, and 4, respectively. As a result, the first power amplifiers 11, 12, and 13 output currents obtained by amplifying the input currents to the respective energization control terminals to the respective power supply terminals of the coils 2, 3, and 4, respectively. 4 is supplied to the negative side of the drive currents I1, I2 and I3.

The current output terminals of the three second power amplifiers 15, 16, and 17 are commonly connected to the positive output terminal of the voltage converter 52 via the current detecting resistor 31. The second power amplifier 15 has a second NMOS power transistor 85, and amplifies the output current H1 of the second current amplifier 45, which is input to the conduction control terminal side, by a predetermined amount, and outputs the result. Second NMOS power transistor 85
And the NMOS transistor 95 constitute a second NMOS power section current mirror circuit. The cell size of the NMOS power transistor 85 is set to 100 times the cell size of the NMOS transistor 95, and a current amplification factor of 101 times is obtained when operating in the active region.
The second NMOS power transistor 85 is constituted by a field effect transistor having a double-diffused N-channel MOS structure, and has a second power transistor formed as a parasitic element from the current outflow terminal side to the current inflow terminal side. It has a diode 85d connected in reverse, and is connected in an equivalent circuit.

Similarly, the second power amplifier 16 has a second NMOS power transistor 86, and the output current H2 of the second current amplifier 46 input to the conduction control terminal side.
Is amplified and output. The second NMOS type power transistor 86 and the NMOS type transistor 96 constitute a second NMOS type power section current mirror circuit.
The cell size of the OS type power transistor 86 is set to NMOS
The cell size of the type transistor 96 is set to 100 times. The second NMOS power transistor 86 is constituted by a field effect transistor having a double-diffused N-channel MOS structure, and a second power transistor formed as a parasitic element from the current outflow terminal side to the current inflow terminal side. It has a diode 86d connected in reverse, and is connected in an equivalent circuit.

Similarly, the second power amplifier 17 has a second NMOS type power transistor 87, and the output current H3 of the second current amplifier 47 inputted to the conduction control terminal side.
Is amplified and output. The second NMOS type power transistor 87 and the NMOS type transistor 97 constitute a second NMOS type power section current mirror circuit.
The cell size of the OS type power transistor 87 is set to NMOS
The cell size of the type transistor 97 is set to 100 times. The second NMOS power transistor 87 is formed of a field effect transistor having a double diffusion N-channel MOS structure, and has a second power source formed as a parasitic element from the current outflow terminal side to the current inflow terminal side. It has a diode 87d connected in reverse, and is connected in an equivalent circuit.

Second NMOS power transistor 8
5, 86, 87 are commonly connected to the positive output terminal of the voltage converter 52 via the resistor 31, and the current outflow terminals are connected to the respective power supply terminals of the coils 2, 3, and 4. Have been. Thereby, the second power amplifier 15,
Reference numerals 16 and 17 respectively output amplified currents of the input currents to the respective energization control terminals to the respective power supply terminals of the coils 2, 3 and 4, and drive currents I1 and I1 to the coils 2, 3 and 4, respectively.
2 and I3 are supplied.

The current supply 30 has a resistor 31 for detecting a current.
And a current detecting section and a supply output section 33 each including a level converting circuit 32. Drive currents I1, I
The combined supply current Iv to the coil corresponding to the combined value of the positive side currents I2 and I3 is detected as a voltage drop of the current detection resistor 31. The level conversion circuit 32 outputs the combined supply current I
A current detection signal Bj corresponding to v is output. The supply output section 33 outputs a first supply current signal C1 and a second supply current signal C2 in response to the current detection signal Bj.

FIG. 3 shows a specific configuration of the current supply device 30. The level conversion circuit 32 includes a voltage / current conversion circuit 151. The voltage / current conversion circuit 151 outputs a current detection signal Bj proportional to the voltage drop of the current detection resistor 31 due to the combined supply current Iv. The current detection signal Bj of the voltage-current conversion circuit 151 flows through the transistor 171 and the resistor 174 of the supply output unit 33, and obtains a voltage signal Cg based on the negative terminal side (-) of the DC power supply 50.

The transistors 171 and 1 of the supply output section 33
The current mirror circuit including the resistors 72, 173 and the resistors 174, 175, 176 generates two current signals proportional to the current detection signal Bj on the collector side of the transistors 172, 173. The collector current of transistor 172 is output via a current mirror circuit of transistors 181 and 182. Collector current Bp1 of transistor 182
And the first predetermined current Qq1 of the constant current source 185 are added, and output as a first supply current signal C1.

The collector current Bp2 of the transistor 173
And the second predetermined current Qq2 of the constant current source 186, and the result is output as a second supply current signal C2. Here, by setting the transistors 171, 172, 173, 181, and 182 to predetermined design values, the first supply current signal C1
And the second supply current signal C2 to the composite supply current I to the coil.
The current signal is a response (proportional or substantially proportional) to v. Further, the first supply current signal C1 and the second supply current signal C2 correspond to the current values Qq1, Q2 of the constant current sources 185, 186.
q2 includes a predetermined bias current. Note that the current values Qq1 and Qq2 of the constant current sources 185 and 186 include zero.

The switching generator 34 shown in FIG.
In order to allow a phase current to flow, three-phase switching current signals D1, D2, and D3 that smoothly change are output. FIG. 2 shows the switching generator 3
4 shows a specific configuration. In this example, the switching generator 34
Is composed of a position detection unit 100 and a switching signal unit 101.

The position detecting section 100 is a magneto-electric conversion element (for example, a Hall element) for detecting a magnetic flux generated in a field portion of the moving body 1.
And the position detecting elements 111 and 112 composed of The position detecting elements 111 and 112 are electrically
Two-phase position detection signals Ja1 and Jb having a phase difference of 0 ° and changing in a smooth sine wave shape with the movement of the moving body 1.
1, Ja2 and Jb2 are output. Where Ja
1 and Ja2 have an opposite phase relationship (electrically 180 ° phase difference), and Jb1 and Jb2 have an opposite phase relationship. It should be noted that the signal of the opposite phase is not counted in the new phase number.

The position detection signals Ja2 and Jb2 are connected to the resistor 11
3, 114 and the third phase position detection signal Jc
1 and the position detection signals Ja1 and Jb1
5, 116 and the third phase position detection signal Jc
Create 2. Accordingly, the position detection unit 100 electrically detects the three-phase position detection signal Ja having a phase difference of 120 °.
1, Jb1, Jc1 (Ja2, Jb2, Jc2).

The switching signal section 101 has a sinusoidal switching current signal D which smoothly changes in response to the three-phase position detection signal.
Create 1, D2 and D3. Transistors 122 and 12
3 shunts the current of the constant current source 121 to the collector side in response to the difference voltage between the first phase position detection signals Ja1 and Ja2. The collector current of the transistor 123 is doubled by the current mirror circuit of the transistors 124 and 125, and is output from the collector of the transistor 125. The collector current of the transistor 125 is the constant current source 1
26, and the difference current between the two is output as the first-phase switching current signal D1. Therefore, the switching current signal D1 changes smoothly in response to the position detection signal Ja1, and the current flows out in the 180 ° section (positive current) and the current flows in the next 180 ° section (in the electrical angle) ( Negative current).

Similarly, the switching current signal D2 smoothly changes in response to the position detection signal Jb1, and the current flows out in the 180 ° electrical angle section (positive current) and the current in the next 180 ° section. Flows in (negative current). Similarly, the switching current signal D3 changes smoothly in response to the position detection signal Jc1, and the current flows out in the 180 ° section (positive current) and the current flows in the next 180 ° section. (Negative current). Thereby, the switching current signals D1, D2, D3
Is a sinusoidal three-phase current signal. FIG.
4 shows waveform examples of phase switching current signals D1, D2, and D3.

The distribution creator 36 shown in FIG.
7 and a second distributor 38. The first distributor 37 receives the three-phase switching current signal D of the switching generator 34.
The first supply current signal C1 of the current supply unit 30 is distributed in response to the first, current, and second supply current signals E1, E2, and E3. The second distributor 38 includes a three-phase switching current signal D1,
In response to D2 and D3, the second supply current signal C2 of the current supply unit 30 is distributed to generate three-phase second distribution current signals G1, G2 and G3 that smoothly change.

FIG. 4 shows a specific configuration of the distribution creator 36. The first distributor 37 includes three first input transistors 201, 202, and 203 and three first distribution transistors 205, 206, and 207.
Each of the first input transistors 201, 202, 2
03, the three-phase switching current signals D1, D2,
D3 is connected to the supplied current inflow / outflow terminal side.

The first input transistors 201, 202,
The signal output terminals of the current path terminal pair 203 are commonly connected. First distribution transistors 205, 206, 20
7 are commonly connected to the current signal input terminal side, and the first supply current signal C1 of the current supply 30 is input to the common connection terminal side. First distribution transistors 205, 206, 207
Is a three-phase switching current signal D
1, D2 and D3 are connected to the side of the current inflow / outflow terminal to be supplied, respectively. As a result, the three first distribution transistors 205, 206, and 207 output the three-phase first distribution current signals E1, E2, and E from their current signal output terminals.
3 is output. Also, the first input transistors 201,
202, 203 and first distribution transistors 205, 20
6,207 use the same type of transistor.

Here, the first input transistor 20
1, 202, 203 and the first distribution transistor 205,
PNP type bipolar transistors are used for 206 and 207. The conduction control terminal of the first input transistor is a base terminal, the signal input terminal of the current path terminal pair is a collector terminal, and the signal output terminal of the current path terminal pair is an emitter terminal. The conduction control terminal of the first distribution transistor is a base terminal, the current signal input terminal is an emitter terminal, and the current signal output terminal is a collector terminal.

The second distributor 38 comprises three second input transistors 211, 212, 213 and three second distribution transistors 215, 216, 217. Each second input transistor 211,
The signal input terminals of the current control terminal and the current path terminal pair of the switching current generators 12 and 213 are connected to the three-phase switching current signal D 1 of the switching generator 34.
D2 and D3 are respectively connected to the supplied current inflow / outflow terminal side. Second input transistors 211, 21
The signal output terminals of the current path terminal pairs 2213 are commonly connected.

The second distribution transistors 215, 216,
217, the current signal input terminal side is commonly connected, and the second supply current signal C2 of the current supply 30 is input to the common connection terminal side. Second distribution transistors 215, 216, 2
Numeral 17 indicates that each of the energization control terminals is connected to a current inflow / outflow terminal to which three-phase switching current signals D1, D2, and D3 are supplied. As a result, the three second distribution transistors 215, 216, and 217 output the three-phase second distribution current signals G1, G from their current signal output terminals.
2 and G3 are output. Also, the second input transistor 2
11, 212, 213 and second distribution transistor 21
5,216,217 use the same type of transistor.

Further, the first input transistor 201,
The polarity of the transistors 202 and 203 is different from the polarity of the transistors of the second input transistors 211, 212 and 213. Here, NPN-type bipolar transistors are used for the second input transistors 211, 212, 213 and the second distribution transistors 215, 216, 217. The conduction control terminal of the second input transistor is a base terminal, the signal input terminal of the current path terminal pair is a collector terminal, and the signal output terminal of the current path terminal pair is an emitter terminal.

The conduction control terminal of the second distribution transistor is a base terminal, the current signal input terminal is an emitter terminal, and the current signal output terminal is a collector terminal. Further, the reference voltage source 220 and the transistors 221 and 222 constitute a predetermined voltage supply unit, and the first input transistors 201 and 20
A first DC voltage is supplied to a common connection terminal of the second input transistors 211, 212, and 213, and a second DC voltage is supplied to a common connection terminal of the second input transistors 211, 212, and 213.

Thus, when the switching current signal D1 is a negative current, a current flows through the first input transistor 201 and no current flows through the second input transistor 211. When the switching current signal D1 is a positive current, a current flows through the second input transistor 211 and no current flows through the first input transistor 201. That is,
A smooth current is supplied complementarily to the first input transistor 201 and the second input transistor 211 in accordance with the polarity of the switching current signal D1, and the current is simultaneously supplied to the first input transistor 201 and the second input transistor 211. It does not flow.

Similarly, when the switching current signal D2 is a negative current, a current is supplied to the first input transistor 202, and when the switching current signal D2 is a positive current, a current is supplied to the second input transistor 212. Similarly, when the switching current signal D3 is a negative current, a current is supplied to the first input transistor 203, and when the switching current signal D3 is a positive current, a current is supplied to the second input transistor 213.

The first distribution transistors 205, 206, and 207 of the first distributor 37 respond to three-phase currents flowing through the first input transistors 201, 202, and 203,
The first supply current signal C1 is distributed to the respective current signal output terminals, and three-phase first distribution current signals E1, E2, E
Produce 3. Therefore, the three-phase first distribution current signal E
1, E2, E3 are three-phase switching current signals D1, D2, D3
, And smoothly changes in response to the negative current on the negative side, and the composite value of the distribution current signals E1, E2, and E3 becomes equal to the first supply current signal C1.

Similarly, the second distribution transistors 215, 216, and 217 of the second distributor 38 respond to the three-phase currents flowing through the second input transistors 211, 212, and 213 to generate the second supply current. The signal C2 is distributed to the respective current signal output terminals, and a three-phase second distributed current signal G1,
Create G2 and G3. Therefore, the three-phase second distribution current signals G1, G2, G3 are three-phase switching current signals D1, D
2, and smoothly changes in response to the positive current of D3, and the composite value of the distribution current signals G1, G2, and G3 becomes equal to the second supply current signal C2. FIG. 29B shows a waveform example of the three-phase first distribution current signals E1, E2, and E3.
(C) shows a waveform example of the three-phase second distribution current signals G1, G2, G3. These current signals change smoothly in the rising slope portion and the falling slope portion.

The first distribution current signals E1, E2, E3 are 1
The second distributed current signals G1 and G have a phase difference of 20 °.
2 and G3 have a phase difference of 120 °. The first distributed current signal E1 and the second distributed current signal G1 have a phase difference of 180 ° and smoothly change complementarily and smoothly.
Is always zero on one side. Similarly, the first distribution current signal E
The second and second distributed current signals G2 have a phase difference of 180 ° and smoothly change complementarily, and one of E2 and G2 is always zero. Similarly, the first distribution current signal E3 and the second distribution current signal G3 have a phase difference of 180 ° and smoothly change complementarily, and one of E3 and G3 always becomes zero.

The first distribution current signals E1, E2, E3 of the first distributor 37 of FIG.
1, 42 and 43. A first current amplifier 41,
42, 43 are the first distribution current signals E1, E, respectively.
2 and E3 are amplified a predetermined number of times to obtain a first amplified current signal F
1, F2, F3, and the first power amplifier 11,
12 and 13 are supplied to the respective energization control terminals. The first power amplifiers 11, 12, and 13 respectively amplify the current of the three-phase first amplified current signals F1, F2, and F3, and drive currents I1, I2 to the coils 2, 3, and 4 from the respective current inflow terminals. , I
3 is supplied.

FIG. 5 shows the first current amplifiers 41, 42, 43
The following shows a specific configuration. The first current amplifier 41 includes a current mirror circuit of a preceding stage including transistors 231 and 232, transistors 233 and 234, and resistors 235 and 236.
, And a first amplification unit current mirror circuit in which the current mirror circuits of the preceding and succeeding stages are cascaded. Transistor 2
The emitter area ratio between 31 and 232 is set to 1, and the current amplification factor of the current mirror circuit in the preceding stage is set to 1.

The emitter area ratio of the transistors 233 and 234 is 50 times, the resistance ratio of the resistors 236 and 235 is 50 times, and the current amplification ratio of the current mirror circuit in the subsequent stage is 50 times. As a result, the first amplifying unit current mirror circuit of the first current amplifier 41 amplifies the current amplification factor by 50 times. Similarly, the first current amplifier 42 includes transistors 241, 242, 243, 244 and a resistor 24.
5, 246, a first amplification unit current mirror circuit, and amplifies the current amplification factor by 50 times.

Similarly, the first current amplifier 43 includes transistors 251, 252, 253, 254 and a resistor 255,
The first amplifying unit is constituted by a first current mirror circuit of 256 and amplifies the current amplification factor by 50 times. As a result, the first current amplifiers 41, 42, and 43 convert the three-phase first amplified current signals F1, F2, and F3 obtained by amplifying the three-phase first distributed current signals E1, E2, and E3 by 50 times. It is supplied to the respective power supply control terminals of the first power section current mirror circuits of the first power amplifiers 11, 12, and 13.

The second distribution current signals G1, G2, G3 of the second distributor 38 of FIG.
5, 46 and 47 are input. A second current amplifier 45,
46 and 47 are second distributed current signals G1 and G, respectively.
2, a second amplified current signal H obtained by amplifying G3 by a predetermined amount.
Create 1, H2, H3. The high voltage output unit 53 responds to the high frequency pulse signal and charges and accumulates the voltage in the boosting capacitor, thereby generating a high potential point Vu higher than the positive terminal side potential Vcc of the DC power supply 50.

The second current amplifiers 45, 46, 47 supply the second amplified current signals H1, H2, H3 to the high voltage output 53.
From the high potential point Vu of the second power amplifiers 15, 16, 1
7 is supplied to each energization control terminal side. The second power amplifiers 15, 16, 17 are provided with three-phase second amplified current signals H1,
The currents H2 and H3 are respectively amplified, and the positive currents of the drive currents I1, I2 and I3 are supplied to the coils 2, 3 and 4 from the respective current outflow terminals.

FIG. 6 shows the second current amplifiers 45, 46, 47.
And a specific configuration of the high voltage output unit 53. The high-voltage output device 53 includes a pulse generation circuit 421 that outputs a high-frequency pulse signal Pa of about 100 kHz, a first boosting capacitor 411, a second boosting capacitor 412, and a first A voltage limiting circuit;
The second voltage limiting circuit including the diode 429 is included. The pulse signal P of the pulse generation circuit 421
In response to a, the inverter circuit 422 digitally changes.

When the inverter circuit 422 is at “L” (for example, the potential on the negative terminal side of the DC power supply 50), the first boosting capacitor 411 is charged via the diode 423. When the inverter circuit 422 changes to “H” (for example, the potential on the positive terminal side of the DC power supply 50), the electric charge accumulated in the first boosting capacitor 411 is
4 and is transferred to the second boosting capacitor 412 to charge and accumulate the second boosting capacitor 412. As a result, a high potential point potential Vu which is higher than the potential of the positive electrode output terminal of the voltage converter 52 is output to the terminal of the second boosting capacitor 412. The high potential point Vu is connected to the second current amplifiers 45, 46, 47.

When the charging of the second boosting capacitor 412 is continued, the voltage Vu at the high potential point becomes very high,
There is a possibility that breakdown voltage of a transistor or a diode integrated may be caused. Therefore, the first voltage limiting circuit including the diodes 425 to 428 limits the high potential point voltage Vu so as not to exceed a predetermined value. Note that the first voltage limiting circuit may be eliminated if there is no concern about breakdown voltage.

Further, the second amplified current signals H1, H2, H
3 acts to discharge the charge of the second boosting capacitor 412. If a large current operation such as when starting the motor continues for a long time, the charge / accumulated charge of the second boosting capacitor 412 becomes insufficient, and the potential Vu at the output voltage point of the high voltage output unit 53 may drop significantly. is there. For this reason, the circuit operation may be temporarily unstable, and the starting operation may be hindered. Therefore, a second voltage limiting circuit using a diode 429 is provided, and the high potential point potential V
u was restricted so as not to be too small. In the normal control state where the current level is small, the second voltage limiting circuit does not operate. When the fluctuation of the potential Vu is small, the second voltage limiting circuit may be omitted.

The second current amplifier 45 includes the transistor 2
61, 262 and resistors 263, 264 and a second amplifier current mirror circuit. The emitter area ratio of the transistors 261 and 262 is 50 times, and the resistance 2
The resistance ratio between 64 and 263 is increased by 50 times, and the current amplification factor is increased by 5
It is 0 times. As a result, the second amplification section current mirror circuit of the second current amplifier 45 performs amplification by 50 times at the current amplification factor. Similarly, the second current amplifier 46
Are transistors 271 and 272 and resistors 273 and 274
, And amplifies the current amplification factor by 50 times.

Similarly, the second current amplifier 47 comprises a second current amplifier 47 comprising transistors 281 and 282 and resistors 283 and 284.
, And amplifies the current amplification factor by 50 times. As a result, the second current amplifiers 45, 46, and 47 amplify the three-phase second distribution current signals G1, G2, and G3 by 50 times, respectively, and three-phase second amplified current signals H1, H2, and H3. And supplies it from the high potential point Vu of the high voltage output unit 53 to each energization control terminal side of the second power section current mirror circuits of the second power amplifiers 15, 16, and 17.

The operation controller 51 shown in FIG. 1 outputs an operation control signal Vd in response to the command signal Ad. Command signal Ad
Is generated by, for example, a command block including a speed control circuit, and is supplied to the operation controller 51 in response to a speed control signal corresponding to a result of comparison between the speed of the moving body 1 and the target speed. The voltage converter 52 generates a predetermined high-frequency PWM signal Sw (pulse width modulation signal) having a pulse width corresponding to the operation control signal Vd of the operation controller 51 in the PWM unit 65, and outputs the NMOS switching transistor 61
Is subjected to a high-frequency switching operation (including a case where the NMOS switching transistor 61 is always on).

That is, the PWM switching operation of the NMOS switching transistor 61 of the voltage converter 52 is controlled in response to the operation control signal Vd of the operation controller 51. The voltage converter 52 is connected to the DC voltage Vc of the DC power supply 50.
By using c as a power supply source, a converted DC voltage (Vcc-Vg) corresponding to the PWM switching operation of the NMOS switching transistor 61 is generated.

FIG. 7 shows a specific configuration of the operation controller 51 and the voltage converter 52. Differential amplifier circuit 2 of operation controller 51
92 compares the command signal Ad with the voltage value of the reference voltage source 291 and amplifies the difference voltage to output the operation control signal Vd. Note that the capacitor 293 forms a filter in the differential amplifier circuit 292.

The PWM unit 65 of the voltage converter 52 includes a triangular wave generation circuit 301 and a comparison circuit 302. The triangular wave generation circuit 301 outputs a triangular wave signal Vh of about 200 kHz. The comparison circuit 302 compares the triangular wave signal Vh of the triangular wave generation circuit 301 with the operation control signal Vd of the operation controller 51, and the PW responding to the operation control signal Vd
Generate an M voltage signal Sw. PWM signal Sw is NMOS
The NMOS switching transistor 61 is supplied to the conduction control terminal side of the switching transistor 61 and turns on and off in response to the PWM signal Sw. NMOS
The type switching transistor 61 performs high-frequency switching on a power supply path for supplementing magnetic energy from the positive terminal side of the DC power supply 50 to the conversion inductor element 63.
The flywheel diode 62 constituting the current path forming circuit performs an on / off operation complementary to the on / off high-frequency switching operation of the NMOS type switching transistor 61, and includes a conversion inductor element 63 to a conversion capacitor element 64. Form a current path to the circuit side.

NMOS switching transistor 61
Conversion inductor element 63 associated with high-frequency switching of
As a result, the converted DC voltage (Vcc-Vg) is output between one end of the conversion capacitor element 64 and one end of the DC power supply 50.

As described above, using the DC voltage of the DC power supply 50 as a power supply source, the NMOS switching transistor 61 performs a high-frequency PWM operation (pulse width modulation operation) in response to the operation control signal Vd. The potential Vg on the output terminal side is variably controlled, and the converted DC voltage value (Vc) is applied between the positive output terminal side and the negative output terminal side of the voltage converter 52.
c-Vg). Conversion DC voltage (Vcc-Vg)
Are the first power amplifiers 11, 12, 1 connected in parallel.
3, the second power amplifiers 15, 16, 1 connected in parallel
7 is supplied.

The first power amplifiers 11, 12, 1 in FIG.
3, the first power transistors 81, 82, 83 and the second
The second power transistors 85, 86, 87 of the power amplifiers 15, 16, 17 and the switching transistor 61 of the voltage converter 52 are composed of the current supplier 30 and the switching generator 3.
4, the distribution generator 36 and the first current amplifiers 41, 42, 4
3, the second current amplifiers 45, 46, 47 and the operation controller 5
1 and the required transistors and resistors of the voltage converter 52 and the high-voltage output device 53 are joined and separated on a single silicon substrate to form an integrated circuit.

FIG. 8 shows an example of the structure of the integrated circuit. The required N + layer, N- layer, P + layer and P-
Various transistors are formed by diffusing layers and the like.
Numeral 191 is an example of a double-diffused NMOS field effect transistor, which is used as a first NMOS power transistor, a second NMOS power transistor, and an NMOS switching transistor. Number 192 is
This is an example of an NPN type bipolar transistor, and is used as a signal amplification transistor. Numeral 193 is an example of a PNP type bipolar transistor, which is used as a signal amplification transistor. Numeral 194 is an example of P-channel and N-channel CMOS field effect transistors, which is used for logic signal processing. The transistors are separated from each other by a P layer having the same potential as the silicon substrate connected to the ground potential (0 V).

A junction-separated integrated circuit uses a low-cost manufacturing process as compared with a dielectric-separated integrated circuit, and a large number of power transistor elements and signal transistors can be formed on a small one-chip substrate at a high density. Can be integrated.
That is, an integrated circuit can be formed at low cost. Note that the specific mask arrangement is a design matter, and a detailed description is omitted.

Next, the overall operation of the motor shown in FIG. 1 will be described. The switching generator 34 generates three-phase switching current signals D1, D2, and D3 that smoothly change, and supplies them to the distribution generator 36. The first distributor 37 includes the current supplier 3
0 of the first supply current signal C1 to the three-phase switching current signal D
1, D2, and D3, and outputs three-phase first distribution current signals E1, E2, and E3. The first current amplifiers 41, 42, and 43 respectively provide a first distribution current signal E
The first power amplifier 1 outputs first amplified current signals F1, F2, and F3 obtained by amplifying currents 1, 1, and 2 by a predetermined number.
1, 12 and 13 are supplied to the respective energization control terminals.

The first power amplifiers 11, 12, 13 are:
Each of the first amplified current signals F1, F2, F3 is current-amplified, and drive currents I1, I2,
A negative current of I3 is supplied. The current supply 30, the first distributor 37, and the first current amplifiers 41, 42, 43
Forms a first distribution control block, and in response to the output signal of the switching generator 34, the first power amplifiers 11, 12, 1
Distribution control of the power supply from 3 is performed.

On the other hand, the second distributor 38 is connected to the current supplier 3
0 of the second supply current signal C2 to the three-phase switching current signal D
1, D2, and D3, and outputs three-phase second distribution current signals G1, G2, and G3. The second current amplifiers 45, 46, 47 respectively provide a second distributed current signal G
, G2, and G3 are amplified by a predetermined number of times to output second amplified current signals H1, H2, and H3.
5, 16 and 17 are supplied to the respective energization control terminals.

The second power amplifiers 15, 16, 17 are:
The currents of the second amplified current signals H1, H2, H3 are respectively amplified, and drive currents I1, I2,
A positive current of I3 is supplied. The current supply 30, the second distributor 38, and the second current amplifiers 45, 46, 47
Forms a second distribution control block, and in response to the output signal of the switching generator 34, the second power amplifiers 15, 16, 1
Distribution control of the power supply from 7 is performed.

The current detecting resistor 31 of the current supplier 30
Detects a combined supply current Iv that is a combined value of the positive side currents of the drive currents I1, I2, and I3, and generates a current detection signal Bj corresponding to the combined supply current Iv via the level conversion circuit 32. The supply output unit 33 includes a first supply current signal C1 and a second supply current signal C2 responsive to the current detection signal Bj.
Is output. The first supply current signal C1 and the second supply current signal C2 change in proportion or substantially in proportion to the combined supply current signal Iv.

The first distribution control block (current supply 30
And the first distributor 37 and the first current amplifiers 41, 42, 4
3) is a three-phase first amplified current signal F1, F in which the rising slope portion and the falling slope portion smoothly change.
2 and F3 (a first three-phase current signal), and outputs the first amplified current signals F1, F2 and F3 to the first power amplifier 1
1, 12, and 13 are supplied to the respective energization control terminals. This ensures that at least one of the three first NMOS power transistors 81, 82, 83 is operated to have a resistive voltage drop, while the three first NMOS power transistors 81, 82 are being operated. , 83 smoothly switch the current path. Where M
The resistive voltage drop operation of the OS-type transistor means an operation in a full-on state in which the operating voltage on the current inflow terminal side and the current outflow terminal side becomes a resistive voltage drop.

Therefore, each of the first NMOS type power transistors performs a current amplifying operation (operation in a half-on state) in the active region in the first period and the last period of the switching of the current path, so that the current value is smoothly changed. Is changed, a resistive voltage drop operation (operation in a full-on state) is performed in a period after the current path is formed. As a result, no spike voltage is generated due to the switching of the current path, and the drive currents I1, I2, I
3 changes smoothly.

The second distribution control block (current supply 30
, A second distributor 38 and second current amplifiers 45, 46, 4
7) is a three-phase second amplified current signal H1, H in which the rising slope portion and the falling slope portion smoothly change.
2, H3 (second three-phase current signal), and the second amplified current signals H1, H2, H3 are supplied to the second power amplifier 1
5, 16 and 17, respectively. This ensures that at least one of the three second NMOS power transistors 85, 86, 87 is operated with a resistive voltage drop, while the three second NMOS power transistors 85, 86 are being operated. , 87 smoothly perform the current path switching operation.

Accordingly, each of the second NMOS type power transistors performs a current amplifying operation in the active region in the first period and the last period of the switching of the current path to smoothly change the current value. Is formed, a resistive voltage drop operation is performed. As a result, no spike voltage is generated due to the switching of the current path, and the drive currents I1, I2, I
3 changes smoothly.

The first distribution control block and the first power amplifier form a positive feedback loop, and their combined transfer gain (the current supply 30, the first distributor 37, and the first current amplifiers 41 and 42). , 43 and the first power amplifiers 11, 12,.
13 forward gain) is increased to 1 or more.
Also, the second distribution control block and the second power amplifier form a positive feedback loop, and the combined transfer gain thereof (the current supply 30, the second distributor 38, and the second current amplifiers 45 and 4).
6, 47 and the second power amplifiers 15, 16, 17) are increased to 1 or more.

Thus, the overall circuit operation is stabilized. That is, at least one of the three first NMOS power transistors 81, 82, 83 reliably performs a resistive voltage drop operation, and the three second NMOS transistors 81, 82, 83 operate.
At least one of the S-type power transistors 85, 86, 87 reliably performs a resistive voltage drop operation, and supplies drive currents I1, I2, I3 to the coils 2, 3, 4. Further, the loop transmission gain of the two positive feedback loops is set as small as possible so that the current path switching operation is performed smoothly.

The first distribution current signal E1 and the second distribution current signal G1 of the same phase flow complementarily with a phase difference of 180 °. Therefore, the first power amplifier 11 and the second power amplifier 15 operate complementarily, and the drive current I1 in both directions, which changes smoothly and continuously, is supplied to the coil 2.
Similarly, the first distributed current signal E2 and the second distributed current signal G2 flow complementarily with a phase difference of 180 °, and the first power amplifier 12 and the second power amplifier 16 operate complementarily, Drive current I in both directions that changes smoothly and continuously
2 is supplied to the coil 3.

Similarly, the first distribution current signal E3 and the second distribution current signal G3 flow complementarily with a phase difference of 180 °, and the first power amplifier 13 and the second power amplifier 1
Reference numeral 7 operates complementarily, and the drive current I3 in both directions, which changes smoothly and continuously, is supplied to the coil 4. in this way,
Since the first power amplifier and the second power amplifier of the same phase are not simultaneously energized, no short-circuit current occurs between the positive output terminal side and the negative output terminal side of the voltage converter 52. As a result, excessive heat generation and overcurrent of the power transistor do not occur. In addition, since the smoothly changing continuous drive currents I1, I2, and I3 are supplied to the coils 2, 3, and 4, no spike voltage is generated in the coils 2, 3, and 4, and the first parasitic element, which is a parasitic element, is generated. Power diode 81
d, 82d, 83d and the second power diode 85d,
No abnormal current flows through 86d and 87d.
Therefore, the pulsation of the driving force generated by the motor is significantly reduced.

The operation controller 51 generates an operation control signal Vd in response to the command signal Ad. The voltage converter 52 operates the NMOS switching transistor 61 in high-frequency PWM in response to the operation control signal Vd. The potential Vg is variably controlled. The converted DC voltage (Vcc-Vg) of the voltage converter 52 is supplied to the first power amplifiers 11, 12, 13 connected in parallel and the second power amplifiers 15, 16, 17 connected in parallel. And the coils 2, 3, via the selected first NMOS power transistor and the selected second NMOS power transistor.
4 to form a current path. As a result, coils 2, 3, 4
Is controlled by the converted DC voltage (Vcc-Vg) of the voltage converter 52 in response to the command signal Ad.

In this embodiment, the motor configuration is suitable for integration into an integrated circuit. First, since a MOS switching transistor and a MOS power transistor are used as power elements, an integrated circuit can be formed on a small chip. In particular, according to recent reviews,
There has been a prospect that MOS power devices can be realized at low cost by being integrated on the same chip. Further, the current supplier 30, the switching generator 34, the distribution generator 36, the first current amplifiers 41, 42, 43, and the second current amplifiers 45, 4
6, 47, the operation controller 51, the voltage converter 52, and the high-voltage output device 53. The required transistors, semiconductor elements such as diodes and resistors are mounted on the same chip as the MOS power transistor and the MOS switching transistor. It is integrated and integrated into a circuit.

The integrated circuit separated from the junction can be integrated at a higher density on a small chip substrate than the integrated circuit separated from the dielectric. As a result, it can be realized at low cost. A field effect transistor having a double-diffused MOS structure is used as the first NMOS power transistor and the second NMOS power transistor, so that an integrated circuit is formed in a small chip size. When a field effect transistor having a double diffusion MOS structure is used, a parasitic power diode is formed from the current outflow terminal to the current inflow terminal. However, since the switching of the current path is performed smoothly, the operation of the parasitic power diode is prevented, and the pulsation of the driving current is reduced.

Further, in this embodiment, the operation of the parasitic transistor element formed in the junction isolation portion is prevented, and the configuration is suitable for integration into an integrated circuit. As shown in FIG. 8, an integrated circuit using the junction separation technology can realize a low-cost IC suitable for high-density integration. However, there is a drawback in that a large number of parasitic transistor elements having a junction terminal connected to the negative terminal side (ground potential) of the DC power supply as a base terminal are formed. Usually, these parasitic transistors are reverse-biased so as not to operate. However, when the terminal potential of the integrated transistor becomes lower than the ground potential by the forward voltage of the diode, the parasitic transistor operates,
A phenomenon occurs in which current is extracted from other integrated transistors.

In an application such as a motor that supplies a large current to a coil having an inductance function or a conversion inductor element, when a parasitic transistor operates, the function of an integrated transistor may be significantly impaired. The NMOS type switching transistor 61 of the present embodiment has its current outflow terminal side connected to the negative terminal side of the DC power supply 50, its current inflow terminal side connected to one end of the conversion inductor element 63, and the positive terminal of the DC power supply 50. The power supply path for replenishing magnetic energy from the terminal side to the conversion inductor element 63 is switched at high frequency.

The flywheel diode 62, which is a current path forming circuit, is connected between one end of the conversion inductor element 63 and the positive terminal side of the DC power supply 50, and the on / off high-frequency switching operation of the NMOS switching transistor 61 is performed. , The current path from the conversion inductor element 63 to the conversion capacitor element 64 is formed.

A converted DC potential (Vcc-Vg) is output between one end of the conversion capacitor element 64 and one end of the DC power supply 50, and the three first power amplifiers 1 connected in parallel are output.
The converted DC voltage is supplied to three second power amplifiers 15, 16, and 17 connected in parallel with 1, 12, and 13.
Thereby, the NMOS type switching transistor 61
The potential of each terminal of the flywheel diode 62 does not fall below the negative terminal side potential of the DC power supply 50. Therefore, even if the NMOS switching transistor 61 performs high-frequency switching, the parasitic transistor does not operate.

Since the current paths of the first NMOS power transistor and the second NMOS power transistor are switched smoothly, the potential of their terminals does not fall below the negative terminal side potential of the DC power supply 50. Therefore,
Even if the current path is switched by the first power transistor or the second power transistor, the parasitic transistor does not operate. As a result, even if the switching transistor, the flywheel diode, the first power transistor, and the second power transistor are integrated into one chip with other transistors, the operation of the parasitic transistor in the integrated circuit is completely prevented. it can.

Further, in this embodiment, the heat generation in each power element is extremely reduced, and the configuration is suitable for integration into an integrated circuit. Since the first NMOS power transistors 81, 82, and 83 of the first power amplifiers 11, 12, and 13 perform a resistive voltage drop operation when energized, the power loss in the first power amplifier is very small. . Since the second NMOS power transistors 85, 86, 87 of the second power amplifiers 15, 16, 17 perform a resistive voltage drop operation when energized, the power loss in the second power amplifier is very small. .

Since the voltage converter 52 performs the voltage conversion by causing the NMOS switching transistor 61 to perform the high-frequency PWM operation, the power loss accompanying the voltage conversion is very small. Further, the NMOS type switching transistor 6
1 performs the PWM operation by the voltage swing of the voltage signal Sw applied to the conduction control terminal (gate terminal), so that the current supply to the conduction control terminal side is extremely small, and almost no power loss occurs. Therefore, power loss and heat generation in the first power amplifier, the second power amplifier, and the voltage converter are extremely small, and the power transistor and the switching transistor can be integrated into one chip.
Also, it is not necessary to take measures against heat generation such as a heat sink.

In the present embodiment, the current supply 30
A current detection unit (resistor 31 and level conversion circuit 32) for obtaining a current detection signal Bj in response to a combined supply current Iv that changes in response to the command signal Ad; and a first supply current signal C1 in response to the current detection signal Bj And a supply output section 33 for outputting a second supply current signal C2. First amplified current signals F1, F responsive to the first supply current signal C1
2 and F3, the first power amplifiers 11, 12, 13
Of the second power amplifiers 15, 16 and 17 using the second amplified current signals H1, H2 and H3 responsive to the second supply current signal C2.

Thus, the rising slope portion and the falling slope portion of the first amplified current signal, and the second
The rising slope portion and the falling slope portion of the amplified current signal change in slope in response to the command signal Ad and the combined current signal Iv. Therefore, even when the combined supply current Iv is large, such as when the motor is started, or when the combined supply current Iv is small, as in the steady control state, the current path switching operation is performed smoothly. It is possible,
The drive currents I1, I2, I3 change smoothly. As a result, the pulsation of the driving current and the pulsation of the driving force due to the switching of the current path become extremely small.

In this embodiment, the first power amplifier is constituted by a first field effect type power section current mirror circuit using a field effect transistor, and the second power amplifier is constituted by a second field effect transistor using a field effect transistor. The first power amplifiers 11, 12, and 13 and the second power amplifier 1 are constituted by a field effect type power section current mirror circuit.
Variations in the current amplification factors of 5, 16, and 17 were reduced.
In addition, the first of three phases that smoothly changes in response to the switching signal.
Of the first three-phase current signals F1, F2, and F3 that smoothly change at least in the rising slope portion and the falling slope portion.
F3 was supplied to each of the conduction control terminals of the three first power amplifiers 11, 12, and 13, respectively.

In addition, three-phase second amplified current signals H1, H2, and H3 that smoothly change in response to the switching signal are generated, and the second three-phase current signals H2 and H3 that change smoothly at least in the rising slope portion and the falling slope portion. The phase current signals H1, H2, H3 are supplied to three second power amplifiers 15,
16 and 17 were supplied to the respective energization control terminals. As a result, the current path switching operation by the three first field-effect power transistors 81, 82, 83 and the three second field-effect power transistors 85, 86, 87 was smoothly performed. As a result, the pulsation of the driving current and the motor vibration were significantly reduced. By integrating the field-effect power transistor into an integrated circuit, the variation in the current amplification factor of the field-effect power section current mirror circuit can be further reduced.

There is also an advantage that variations in the combined transmission gain between the first power amplifier and the first distribution control block and the combined transmission gain between the second power amplifier and the second distribution control block are reduced. As a result, the first supply current signal C1 and the second supply current signal C2 of the current supply 30 are changed in proportion to or approximately in proportion to the combined supply current Iv to the coil that changes in response to the command signal Ad. While 3
Ensuring that at least one of the first MOS-type power transistors is in a resistive voltage drop operation, and that at least one of the three second MOS-type power transistors is of a resistive type. A configuration capable of voltage drop operation has been realized.

With this configuration, the first three-phase current signal having an appropriate slope portion can be provided regardless of whether a large current is supplied at the time of startup or a small current is supplied during steady-state control. Can be supplied to the conduction control terminal side of the first power amplifier, and a second three-phase current signal having an appropriate slope portion can be supplied to the conduction control terminal side of the second power amplifier. as a result,
A pulsating drive current having a smoothly inclined portion can always be supplied to the coil, and the pulsation of the generated driving force is significantly reduced. In order to smoothly switch the current paths, the first three-phase current signals F1, F2, F3 and the second three-phase
It is preferable that the rising slope portion or the falling slope portion of the phase current signals H1, H2, H3 is set to an electrical angle of 15 ° or more.

Further, the current signal F1 and the current signal H1 forming the same phase are such that the start of the slope of F1 coincides or substantially coincides with the end of the slope of H1, and the start of the slope of H1 is the slope of F1. Most preferably, they flow complementarily so as to match or substantially match the end portion. Current signal F
2 and the current signal H2, and the current signal F3 and the current signal H
The same applies to No. 3.

Further, in the present embodiment, the first current amplifier is constituted by the first current mirror circuit of the first amplification section,
Is constituted by the second current mirror circuit of the second amplifying unit to reduce the variation of the current amplification factor. Therefore, the configuration is suitable for integration into an integrated circuit. Thus, the first supply current signal C1 and the second supply current signal C2 of the current supply 30 are changed in proportion to the combined supply current to the coil in response to the command signal Ad, and the three first MOSs
At least one and three of the type power transistors
At least one of the second MOS power transistors was reliably operated in a resistive voltage drop operation. As a result, the first MOS power transistor and the second MOS
The smooth switching operation of the current path by the type power transistor becomes extremely stable.

Further, in the present embodiment, the distribution generator 36 is devised so that the first amplified current signal and the second amplified current signal of the same phase have a phase difference of 180 °, and are complementarily and smoothly cut. It was changed so that one of the first amplified current signal and the second amplified current signal was always zero. This allows
The first power amplifier and the second power amplifier in the same phase do not simultaneously become energized. As a result, no short-circuit current is generated, so that current destruction or thermal destruction of the power transistor does not occur.

In this embodiment, the first power amplifiers 11, 12, 13 and the second power amplifiers 15, 16, 1, 1
7, current supply 30, switching generator 34, distribution generator 36
(The first distributor 37 and the second distributor 38), the first current amplifiers 41, 42, 43, and the second current amplifiers 45, 4
6, 47, the operation controller 51, the voltage converter 52, and the high voltage output unit 53 form a drive circuit for supplying a drive current to the three-phase load (coils 2, 3, 4). The DC power supply 50 and the voltage converter 52 form a voltage supply block that supplies a required converted DC voltage (Vcc−Vg) between the positive output terminal side and the negative output terminal side of the voltage converter 52. . These configurations can be changed as needed. Note that the switching creator 34 of the present embodiment is configured to include the position detection unit 100 using a magnetoelectric conversion element. However, without using such an element, for example, the coil 2,
A three-phase switching signal may be created by using the back electromotive force generated in 3, 4.

The three-phase first amplified current signals F1, F
2, F3 or three-phase second amplified current signals H1, H
2 and H3 may be switched with a temporal gradient in the rising slope portion and the falling slope portion.
As a result, the drive currents I1, I2, and I3 also switch the current paths smoothly with a temporal gradient in the rising slope portion and the falling slope portion. Further, it is preferable that the current value be continuously changed when the polarity of the drive current is changed. However, there is a period in which the first amplified current signal and the second amplified current signal of the same phase become zero at the same time. There may be a time for reducing the drive current of the drive to zero.

In the present embodiment, the first power amplifiers 11, 12, 13 and the second power amplifiers 15, 1, 1
6 and 17 are not limited to the configuration shown in FIG. 1 and various modifications are possible. For example, the first power amplifier 11, 1
2 and 13 and the second power amplifiers 15, 16, and 17, instead of the power amplifier 1 having the configuration shown in FIG.
000 may be used. Power amplifier 1000 is NM
The OS-type power transistor 1010, the NMOS-type transistor 1011 and the resistor 1012 constitute a field-effect type power section current mirror circuit.

Field effect type power section current mirror circuit 1
000, the control terminal side of the field-effect power transistor 1010 is connected to the control terminal side of the field-effect transistor 1011 (directly or through some element such as a resistor), and the current path terminal pair of the field-effect transistor 1011 One terminal side is connected to one terminal side of the current path terminal pair of the field effect type power transistor 1010 by a resistor 10.
12 and a field-effect transistor 101
The other terminal side of the current path terminal pair 1 is a power amplifier 1
000 (directly or through some element) and the field-effect transistor 10
The control terminal side of the power amplifier 1000 is connected (directly or through some element) to the control terminal side of the power amplifier 1000. This field effect type power section current mirror circuit has a current amplification ratio relatively larger than the cell size ratio. Thereby, there is an advantage that the input current to the power amplifier can be reduced.

For example, power amplifier 1100 having the configuration shown in FIG. 23 may be used. Power amplifier 1
100 is an NMOS type power transistor 1110 and NM
The OS-type transistor 1111 and the resistor 1112 constitute a field-effect power section current mirror circuit.
In the field effect type power section current mirror circuit 1100, the control terminal side of the field effect type power transistor 1110 is connected (directly or through some element) to the control terminal side of the field effect type transistor 1111, and the current of the field effect type transistor 1111 is changed. One terminal of the power path terminal pair is connected to the conduction control terminal of the power amplifier 1100 via the resistor 1112, and the other terminal of the current path terminal pair of the field effect transistor 1111 is connected to the current path of the field effect power transistor 1110. The control terminal side of the field-effect transistor 1111 is connected (directly or through some element) to the one terminal side of the terminal pair (directly or through some element), and the control terminal side of the field effect transistor 1111 is connected to the power supply control terminal side of the power amplifier 1110. It is configured to:

This field effect type power section current mirror circuit has a predetermined current amplification factor while the input current to the conduction control terminal side is small, and the current amplification factor sharply increases as the input current increases. . The NMOS power transistor 1010 and the NMOS power transistor 1
110 can be constituted by a field effect type power transistor having a double diffusion N-channel MOS structure, and is easily integrated.

In the present embodiment, the voltage converter 52
Flywheel diode 6 which is the current path forming circuit of FIG.
Various modifications can be made to the portion 2. For example, an NMOS type synchronous rectification transistor 1400 having the configuration shown in FIG. The rectifying transistor 1400 can be turned on and off. The synchronous rectification transistor 1400 is constituted by a field effect type power transistor having a double-diffused N-channel MOS structure, and a parasitic diode 1400d is reversely connected between the current input and output terminals of the synchronous rectification transistor 1400.
With the parasitic diode 1400d, the effect of the flywheel diode can also be obtained.

Further, for example, the P of the configuration shown in FIG.
Using MOS type synchronous rectification transistor 1500, PW
By the signal from the M unit 65, the synchronous rectification transistor 1500 can be turned on and off in a complementary manner to the on / off switching operation of the switching transistor 61. Synchronous rectification transistor 1
Reference numeral 500 denotes a field effect type power transistor having a double-diffused P-channel MOS structure, and a parasitic diode 1500d is reversely connected between the current input and output terminals of the synchronous rectification transistor 1500. The effect of the diode for the flywheel can be obtained by the parasitic diode 1500d. Further, it is easy to integrate the synchronous rectification transistors 1400 and 1500 into an integrated circuit.

The configuration of the current supply 30 is not limited to that shown in FIG. 1, and various modifications are possible. The current supply device 30 in FIG. 1 may be replaced with, for example, a current supply device 950 having the configuration shown in FIG. Current supply 950
Converts a current signal Bk proportional to or substantially proportional to the difference voltage between the command signal Ad and the voltage of the voltage source 951 into a level converter 952.
And a first supply current signal C1 proportional or substantially proportional to the current signal Bk by the supply output unit 953.
And the second supply current signal C2. That is, the first supply current signal C1 directly responding to the command signal Ad and the second
Is obtained. Note that the current supply 95
0 is a specific configuration of the current supply unit 3 shown in FIG.
0 and the detailed description is omitted.

Embodiment 2 FIGS. 9 to 12 show a motor according to Embodiment 2 of the present invention. FIG. 9 shows the overall configuration. In the second embodiment, the modulation section 300 is provided in the operation controller 310, and the converted DC voltage of the voltage converter 52 is changed in response to the modulated signal. In other configurations, the same components as those in the first embodiment are denoted by the same reference numerals, and detailed description thereof will be omitted.

The operation controller 310 shown in FIG.
And an operation control signal V corresponding to a modulated current signal Pm to be described later.
Output d. The voltage converter 52 causes the NMOS switching transistor 61 to perform a high-frequency switching operation in response to the operation control signal Vd. The voltage converter 52 uses the DC voltage Vcc of the DC power supply 50 as a power supply
A converted DC voltage (Vcc-Vg) corresponding to the PWM switching operation of the S-type switching transistor 61 is generated.

FIG. 10 shows the operation controller 310 and the voltage converter 5.
2 shows a specific configuration. The differential amplifier circuit 292 of the operation controller 310 compares the command signal Ad with the voltage value of the reference voltage source 291 and amplifies the difference voltage to output an amplified signal Vf. The modulator 300 outputs a modulated current signal Pm that changes in an analog manner in response to the moving operation of the moving body 1.
The modulated current signal Pm is supplied to the resistor 296 of the synthesizing unit 295, and generates an operation control signal Vd at a terminal of the resistor 296. That is, the operation control signal Vd is the command signal Ad.
And in response to the modulated current signal Pm. The operation control signal Vd is supplied to the voltage converter 52. Note that the capacitor 293 forms a filter in the differential amplifier circuit 292.

FIG. 11 shows a specific configuration of the modulation section 300. The modulation unit 300 includes a modulation generation circuit 3 that obtains modulation signals R1, R2, and R3 that change in synchronization with the rotational movement of the moving body 1.
91, an amplitude circuit 392 that generates an amplitude current signal Lm, and a modulated output circuit 393 that outputs a modulated current signal Pm obtained by modulating the amplitude current signal Lm in response to the modulation signals R1, R2, and R3. ing.

Transistor 32 of modulation creating circuit 391
1, 322 are the position detection signals Ja1,
In response to Ja2, the current of the constant current source 317 is distributed to the collector side. The collector currents of the transistors 321 and 322 are compared by the current mirror circuit of the transistors 324 and 325. The absolute value of the difference current between the two is determined by transistors 325, 326, 327, 328, 329, 33
It is output through an absolute value circuit by 0 and produces a voltage signal R1 at the resistor 361. That is, the voltage signal R1 corresponds to the absolute value of the position detection signal Ja1.

Similarly, the transistors 331 to 340, the constant current source 318, and the resistor 362 generate a voltage signal R2 corresponding to the absolute value of the position detection signal Jb1 at the terminal of the resistor 362. Similarly, the transistors 341 to 350, the constant current source 319, and the resistor 363 generate a voltage signal R3 corresponding to the absolute value of the position detection signal Jc1 at the terminal of the resistor 363.
That is, the voltage signals R1, R2, and R3 are three-phase absolute value signals responsive to the three-phase position detection signals Ja1, Jb1, and Jc1.

The amplitude circuit 392 outputs an amplitude current signal Lm that determines the amplitude of the modulated current signal Pm (a specific configuration will be described later). The transistors 371, 372, 373, and 374 of the modulated output circuit 393 and the diode 375
376 compares the three-phase absolute value voltage signals R1, R2, and R3 with a predetermined voltage (here, the voltage of the common connection terminal of the resistors 361, 362, and 363), and responds to the comparison result to generate the amplitude current signal Lm. Transistors 371, 372, 373, 3
74 is diverted to the collector side. Transistors 371, 3
The collectors of the transistors 72 and 373 are connected in common.
77,378 compared by the current mirror circuit,
The difference current is output as the modulated current signal Pm via the current mirror circuit of the transistors 379 and 380.

Thus, the amplitude of the modulated current signal Pm is changed in an analog manner in synchronization with the rotational movement of the moving body. In particular, transistors 371, 372, 373, 37
4 and the diodes 375 and 376, the minimum value of the three-phase absolute value voltage signals R1, R2 and R3 and the amplitude circuit 38.
The modulated current signal Pm changes in accordance with the result of multiplication of the amplitude current signal Lm of No. 9. Three-phase absolute value voltage signals R1, R2,
The minimum value of R3 is a harmonic signal that changes six times for one cycle of the position detection signal. Therefore, the modulated current signal Pm has a peak amplitude proportional to the amplitude current signal Lm, and is 6 times per cycle of the position detection signal (360 electrical degrees).
Times, it becomes a harmonic signal that changes in an analog manner. This number of times corresponds to the number of times the first power transistor and the second power transistor switch the current paths to the coils 2, 3, and 4.

FIG. 12 shows an example of a specific configuration of the amplitude circuit 392. Voltage-current converter 401 of amplitude circuit 392
Outputs a current signal L1 proportional to the converted DC voltage between the positive output terminal side and the negative output terminal side of the voltage converter 52 in the outflow direction. The voltage-current conversion circuit 402 outputs a current signal L2 proportional to the combined supply current Iv in the inflow direction.

The constant current source 403 outputs a current signal L3 having a predetermined value in the inflow direction. Therefore, as for the amplitude current signal Lm of the amplitude circuit 392, the current signals L1, L2 and L3 are arithmetically synthesized including the inflow and outflow directions, and Lm = | L1 |-| L2 |
− | L3 |. Here, | A | means the absolute value of signal A. The current signal | L1 | corresponds to the converted DC voltage of the voltage converter 52, the current signal | L2 | corresponds to the internal resistance of the coil and the voltage drop at the resistor 31, and the current signal | L3 | This corresponds to a resistive voltage drop (on-voltage) in the formed first power transistor and the second power transistor.

As a result, the amplitude circuit 392 constitutes a back electromotive force estimating mechanism for estimating the magnitude of the back electromotive force generated in the coil through which the current is supplied. The current signal responds to the magnitude. Therefore, the modulated current signal Pm in FIG. 11 changes in an analog manner in response to the moving operation of the moving body 1 and is a harmonic signal having a peak amplitude corresponding to the magnitude of the back electromotive force of the coil. . The operation control signal Vd in FIG. 10 changes in response to both the command signal Ad and the modulated current signal Pm.

The PWM section 65 of the voltage converter 52 shown in FIG.
Includes a triangular wave generation circuit 301 and a comparison circuit 302. The comparison circuit 302 includes the triangular wave generation circuit 30
1 and the operation control signal V of the operation controller 51.
d is compared to generate a PWM voltage signal Sw corresponding to the operation control signal Vd. The PWM signal Sw is supplied to the conduction control terminal side of the NMOS switching transistor 61,
The NMOS switching transistor 61 is turned on / off in response to the PWM signal Sw.

NMOS switching transistor 61
The current outflow terminal side is connected to the negative terminal side of the DC power supply 50, the current inflow terminal side is connected to one end of the conversion inductor element 63, and from the positive terminal side of the DC power supply 50 to the conversion inductor element 63. The high frequency switching of the power supply path for replenishing the magnetic energy is performed.

The flywheel diode 62 is connected between one end of the conversion inductor element 63 and the positive terminal of the DC power supply 50, and is turned off in a complementary manner to the on / off high-frequency switching operation of the NMOS switching transistor 61. It turns on to form a current path from the conversion inductor element 63 to the conversion capacitor element 64. NMO
The S-type switching transistor 61 increases / decreases the magnetic energy of the conversion inductor element 63 accompanying the high-frequency switching, so that the converted DC voltage (Vcc-V) is applied between one end of the conversion capacitor element 64 and one end of the DC power supply 50.
g) is output.

Accordingly, the voltage converter 52 of FIG. 9 operates the NMOS switching transistor 61 in a high-frequency PWM operation in response to the command signal Ad and the modulated current signal Pm, and operates between the positive output terminal side and the negative output terminal side. Conversion DC voltage (V
cc-Vg). In addition, N for PWM operation
Since voltage conversion is performed using the MOS switching transistor 61 and the conversion inductor element 63, power loss in the voltage converter 52 is small. In particular, the PWM signal S
Since the NMOS-type switching transistor 61 is completely turned on / off by w, the heat generation of the switching transistor 61 is extremely small.

Here, the operation controller 310 controls the modulation section 300
And generates an operation control signal Vd corresponding to the modulated current signal Pm. In response to the operation control signal Vd, the voltage converter 52 causes the NMOS switching transistor 61 to perform a high-frequency PWM operation, and the potential V on the negative output terminal side.
g is variably controlled. Therefore, the potential Vg on the negative output terminal side of the voltage converter 52 and the converted DC voltage (Vcc-Vg)
Changes in response to the modulated current signal Pm.

The converted DC voltage (Vcc-
Vg) is the first power amplifier 1 connected in parallel.
The first NM is supplied to second power amplifiers 15, 16, 17 connected in parallel with
OS-type power transistor and selected second NMOS
The type power transistor performs a resistive voltage drop operation to form a current path to the coils 2, 3, 4. The modulated current signal Pm of the modulation section 300 changes in response to the converted DC voltage of the voltage converter 52, and has a peak amplitude corresponding to the magnitude of the back electromotive force of the energized coil. The modulated current signal Pm is a harmonic signal that changes in synchronization with the moving operation of the moving body 1. As a result, the effect of the ripple of the back electromotive force generated in the coil in the energized state is offset and compensated. This will be described.

A sine-wave-like back electromotive force having a predetermined phase difference is generated in each of the coils 2, 3, and 4 of each phase, but the coils having a current path to which the converted DC voltage of the voltage converter 52 is applied are formed. The combined back electromotive force has a ripple component as the current path is switched. Therefore, it was found that the drive current to the coil pulsated under the influence of the ripple of the combined back electromotive force, and a large pulsation occurred in the generated drive force. Therefore,
The converted DC voltage (Vcc-Vg) of the voltage converter 52 is changed in response to the modulated current signal Pm of the modulating unit 300, thereby canceling out and compensating for the effect of the ripple of the back electromotive force generated in the energized coil. Was. In particular, by changing the modulated current signal Pm in response to the converted DC voltage of the voltage converter 52 and making the peak amplitude of the modulated current signal Pm respond to the magnitude of the back electromotive force, the influence of the back electromotive force is reduced. Accurate compensation. As a result, the pulsation of the driving force is significantly reduced, and a high-performance motor with less vibration can be realized.

In particular, in the amplitude circuit 392 shown in FIG. 12, the amplitude current signal Lm is changed according to the magnitude of the back electromotive force of the energized coil, and the peak of the modulated current signal Pm of the modulation section 300 is changed. The amplitude is changed according to the magnitude of the back electromotive force. This makes it possible to include a harmonic component having an appropriate amplitude in the converted DC voltage of the voltage converter 52 even if the back electromotive force changes significantly due to a change in the rotational movement speed of the moving body 1. As a result, the pulsation of the generated driving force is always small.

FIG. 13 shows another configuration example of the amplitude circuit 392. Here, the current signal L1 corresponding to the converted DC voltage of the voltage converter 52 is directly used as the amplitude current signal Lm. As a result, the modulated current signal Lm also changes in response to the converted DC voltage. Even with such a configuration, pulsation of the driving force can be reduced.

FIG. 14 shows another example of the configuration of the amplitude circuit 392. Here, the voltage-current conversion circuit 411 and the constant current source 413 arithmetically combine the current signal L4 responsive to the amplified voltage signal Vf of the operation controller 51 responsive to the command signal Ad and a predetermined current value L5 to obtain an amplitude current The signal Lm is obtained. Therefore, the amplitude current signal Lm and the modulated current signal Pm change in response to the command signal Ad. Even with such a configuration, pulsation of the generated driving force can be reduced. When the change in the magnitude of the back electromotive force is small, the amplitude current signal Lm can be set to a constant value.

Other configurations and operations are the same as those of the first embodiment.
The detailed description is omitted. In this embodiment,
The power loss and heat generation of the power element are reduced. Since the first power amplifier and the second power amplifier perform a resistive voltage drop operation when energized, the power loss in the first NMOS power transistor and the second NMOS power transistor is very small. In addition, the voltage converter 5
The power loss in the two NMOS switching transistors 61 is also small.

As described above, since the heat generation in the power element is small, the motor configuration is suitable for integration into an integrated circuit. Therefore, the current supplier 30, the switching generator 34, the distribution generator 36, the first current amplifiers 41, 42, 43, the second current amplifiers 45, 46, 47, the operation controller 310, the voltage converter 52, It becomes easy to integrate necessary transistors, diodes and resistors of the voltage output device 53 on the same chip as the power transistor and the switching transistor.

Further, in this embodiment, the operation controller 310 is provided with the modulation section 300 for obtaining the modulated current signal Pm which changes in synchronization with the moving operation of the moving body 1, and responds to the output signal Pm of the modulation section 300. Thus, the converted DC voltage of the voltage converter 52 was changed. As a result, the pulsation of the driving force has been significantly reduced. Further, the modulated current signal Pm of the modulator 300 is changed in response to the converted DC voltage of the voltage converter 52, and
The pulsation of the driving force can always be reduced even when the rotational speed of the motor changes.

Further, the switching creator 34 of the present embodiment includes a position detecting section 100 using a magneto-electric conversion element. However, instead of using such an element, for example, a three-phase switching signal may be generated using the back electromotive force generated in the coils 2, 3, and 4. At this time, a zero-crossing point of the back electromotive force is used as a timing signal to obtain a modulated signal of the modulation unit that changes in synchronization with the moving operation of the moving body 1, and the voltage converter responds to the modulated signal. Can be changed.

Embodiment 3 FIGS. 15 to 17 show a motor according to Embodiment 3 of the present invention. FIG. 15 shows the overall configuration. In the third embodiment, when the DC power supply 50 is turned off, the power path switch device 54 is turned off, and a rectified DC voltage of the back electromotive force of the coils 2, 3, 4 is taken out to the terminal Xf of the voltage extractor 490. Further, a high voltage output device 450 having a different configuration was used. In other configurations, the same components as those in the above-described second or first embodiment are denoted by the same reference numerals, and detailed description thereof will be omitted. The high-voltage output device 450 of FIG. 15 includes a boosting inductor and a boosting capacitor, generates a high potential point Vu higher than the positive terminal potential Vcc of the DC power supply 50, and generates the second current amplifiers 45 and 46. , 47
To supply.

FIG. 16 shows a specific configuration of the high voltage output device 450. The high-voltage output device 450 includes a pulse generation circuit 461 that outputs a high-frequency pulse signal Pa of about 100 kHz.
, A boost inductor 451 and a boost capacitor 45
2, a first voltage limiting circuit including diodes 475 to 478, and a second voltage limiting circuit including diode 479. In response to the pulse signal Pa of the pulse generation circuit 461, the inverter circuit 462 digitally changes.

When the pulse signal Pa is “L”, the transistor 464 is turned on, a current flows through the boost inductor 451 through the transistor 464, and the boost inductor 451 is charged with magnetic energy. Pulse signal Pa
Changes to "H", the transistor 464 is turned off,
Due to the magnetic energy stored in the boost inductor 451, the boost capacitor 45
2, a charging path for flowing current is formed.
52 is charged to accumulate charge. As a result, a high potential point potential Vu which is higher than the positive terminal side potential Vcc of the DC power supply 50 is output to the terminal of the boosting capacitor 452.

Further, if the charging of the boosting capacitor 452 is continued, the voltage Vu at the high potential point becomes extremely high, causing breakdown of the integrated circuit transistors and diodes. The first voltage limiting circuit including the diodes 475 to 478 limits the high-potential point voltage Vu so that it does not exceed a predetermined value, thereby preventing breakdown voltage breakdown. In addition, the second voltage limiting circuit using the diode 479 limits the high-potential point voltage Vu of the high-voltage output device 450 so as not to be significantly lower than the positive terminal side potential Vcc of the DC power supply 50.
As a result, even when a large current is supplied, such as when the motor is started, the high potential point potential Vu does not become excessively small, and the operation of the integrated circuit is stabilized. Note that the first voltage limiting circuit and the second voltage limiting circuit need not be connected if unnecessary.

The DC power supply 50 shown in FIG. 15 comprises, for example, a DC voltage source 70 and a switch circuit 71. When the DC power supply 50 is on, the switch circuit 71 is connected to the Ta terminal side and outputs the DC voltage of the DC voltage source 70 between the positive terminal side and the negative terminal side. When the DC power supply 50 is off, the switch circuit 71 is connected to the Tb terminal side, and equivalently, the positive terminal side and the negative terminal side of the DC power supply 50 are short-circuited. The DC power supply 50 is normally in an on state, but is turned off when the power supply is off, in an emergency, or in an abnormal state.

The power path switch 54 of FIG. 15 includes a PMOS type power path transistor 72 which is turned on / off in response to the output voltage of the DC power supply 50. When the DC power supply 50 supplies a predetermined output voltage, the PMO
The S-type power path transistor 72 is turned on, and the DC power supply 5
0, the second power amplifiers 15, 16, 1
7 is connected to a power path to the current inflow terminal side. DC power supply 5
When the output voltage of 0 becomes zero or becomes smaller than a predetermined value, P
The MOS power path transistor 72 is turned off, and the positive terminal side of the DC power supply 50 and the second power amplifiers 15, 16,
The current path to the current inflow terminal 17 is opened.

The PMOS power path transistor 72 has a current inflow terminal connected to the positive terminal of the DC power supply 50 and a current outflow terminal connected to the positive output terminal of the voltage converter 52 and the second power amplifier 15. , 16 and 17 are connected to the current inflow terminal side. The operation of the PMOS power path transistor 72 is switched by the switch control unit 73, and is turned on / off in response to the output voltage of the DC power supply 50.

FIG. 17 shows a specific configuration of the power path switch unit 54. The switch control unit 73 includes an NMOS transistor 311 and a resistor 312. When the switch circuit 71 of the DC power supply 50 is on the Ta side and the DC power supply 50 is outputting a predetermined voltage, the NMOS
The type transistor 311 is turned on, and the PMOS type power path transistor 72 is turned on. When the switch circuit 71 of the DC power supply 50 is switched to the Tb side and the DC power supply 50 is turned off, the NMOS transistor 311 is turned off and the PMOS power path transistor 72 is also turned off.
Here, the PMOS power path transistor 72 is constituted by a field effect transistor having a double-diffused P-channel MOS structure connected in reverse.

That is, the drain terminal is on the current inflow terminal side, the source terminal is on the current outflow terminal side, and the switch diode 72d formed as a parasitic element is connected in an equivalent circuit from the current inflow terminal side to the current outflow terminal side. Have been. When the PMOS power path transistor 72 is on, both ends of the switch diode 72d are short-circuited,
When the S-type power path transistor 72 is off, the switch diode 72d functions as a backflow prevention diode. However, even if the switch diode 72d does not exist, there is no problem in operation.

The voltage extractor 490 shown in FIG. 15 includes a first extractor diode 491 and a second extractor diode 492 whose output sides are commonly connected. The input side of the first extraction diode 491 is connected to the common connection terminal side of the second NMOS power transistor of the second power amplifier. The input side of the second extraction diode 492 is connected to the positive terminal side of the DC power supply 50. The positive output terminal Xf of the voltage extractor 490 is a common connection terminal side of the first extraction diode 491 and the second extraction diode 492. As a result, the DC voltage generated at the common connection terminal side of the second NMOS power transistor is compared with the output DC voltage Vcc of the DC power supply 50, and the DC voltage corresponding to the larger voltage value of the two is output to the voltage extractor. 490 to the output terminal Xf.

Normally, DC power supply 50 is on, and a DC voltage corresponding to output DC voltage Vcc of DC power supply 50 is output to output terminal Xf of voltage extractor 490. However, when the DC power supply 50 is turned off in an emergency, the output voltage Vcc of the DC power supply 50 becomes zero and the power path switch 5
The fourth PMOS power path transistor 72 is turned off.

When the DC power supply 50 is turned off, the first supply current signal C 1 and the second supply current signal C
2 becomes zero or very small, and the first distribution current signal of the first distributor 37 and the second distribution current signal of the second distributor 38 become zero or very small. As a result, the first NMOS power transistors 81, 82, 83 of the first power amplifiers 11, 12, 13 and the second PMOS power transistors 85, 86 of the second power amplifiers 15, 16, 17 are obtained. , 87 are stopped. At this time, the coils 2, 3, 4
Generates a three-phase back electromotive force.

The first power diodes 81d, 82d,
83d and second power diodes 85d, 86d, 87
d and the switching diode 61d are connected to the coils 2, 3,
4 rectifies the back electromotive force and outputs a rectified DC voltage to the common connection terminal side of the second power amplifier. Since the power path transistor 72 of the power path switch 54 is off, the rectified DC voltage is output to the output terminal Xf via the first extraction diode 491 of the voltage extractor 490. That is, when the DC power supply 50 is turned off, the voltage extractor 490 extracts the rectified DC voltage obtained by rectifying the three-phase back electromotive force generated in the coils 2, 3, and 4 by the power diode to the output terminal Xf. Using the output voltage of the voltage extractor 490, it is possible to perform various types of emergency evacuation processing.

Other configurations and operations are the same as those of the first embodiment.
Alternatively, it is similar to the second embodiment, and a detailed description is omitted. In the present embodiment, when the DC power supply 50 is turned off, a rectified DC voltage obtained by rectifying the three-phase back electromotive force generated in the coil via the first power diode and the second power diode is output to the voltage extractor 490. To the terminal Xf. Using the output voltage of the voltage extractor 490, it is possible to perform various types of emergency evacuation processing. For example,
When the motor of this embodiment is used as a spindle motor of a hard disk drive, when the DC power supply 50 is turned off, the output voltage of the voltage extractor 490 is used to electrically store the memory contents of the microcomputer. This makes it possible to mechanically retreat the reproducing head.

When the DC power supply 50 is turned off,
If the first supply current signal C1 of the current supply 30 is set to a predetermined value, the first NMOS power transistors 81, 82 of the first power amplifiers 11, 12, 13 along with the rotational movement of the moving body 1. , 83 can be sequentially turned on. If the PWM signal Sw of the voltage converter 52 is set to a high voltage, the NMOS switching transistor 61 can be turned on. As a result, the negative voltage of the three-phase back electromotive force of the coil is rectified by activating the power transistor and the switching transistor, and the positive voltage is changed to the second power diode 85d,
Rectification can be performed by 86d and 87d.

Also, if the second supply current signal C2 is not a predetermined value and the second power amplifiers 15, 16, 17 and the high voltage output device 450 are operated, the positive polarity of the three-phase back electromotive force of the coil The side voltage can also be rectified by the power transistor. Note that the voltage extractor 490 may be configured using a MOS transistor that performs a switching operation instead of the diode. Further, in this embodiment, various advantages similar to those of the above-described embodiment can be obtained.

Fourth Embodiment FIGS. 18 and 19 show a motor according to a fourth embodiment of the present invention. FIG. 18 shows the overall configuration. In the fourth embodiment, the second power amplifiers 615, 616, 61
7 shows the second PMOS type power transistors 685, 68
6,687 and no high voltage output device. Accordingly, the second current amplifiers 645, 646,
647 was changed. In other configurations, the same components as those in the above-described third, second, or first embodiment are denoted by the same reference numerals, and detailed description thereof will be omitted.

The current inflow terminals of the three second power amplifiers 615, 616, 617 are commonly connected to the positive output terminal side of the voltage converter 52 of FIG. 18 via the current detecting resistor 31. I have. The second power amplifier 615 has the second
And a second field effect type power section current mirror circuit including a PMOS type power transistor 685 and a PMOS type transistor 695, and amplifies the output current H1 of the second current amplifier 645 input to the conduction control terminal side in a predetermined manner. Output. Here, the PMOS transistor means a field-effect transistor having a P-channel MOS structure.

Second PMOS Power Transistor 68
The cell size of the cell No. 5 is 100 times the cell size of the PMOS transistor 695, and a current amplification factor of 100 times is obtained when operating in the active region. Also, the second PMO
The S-type power transistor 685 is constituted by a field effect transistor having a double-diffused P-channel MOS structure, and its current outflow terminal side to the current inflow terminal side.
Second power diode 68 formed as a parasitic element
5d is reversely connected in an equivalent circuit.

Similarly, the second power amplifier 616 is connected to the second power amplifier 616.
And a second field effect type power section current mirror circuit including a PMOS type power transistor 686 and a PMOS type transistor 696, and amplifies the output current H2 of the second current amplifier 646 input to the conduction control terminal side in a predetermined manner. And output (the cell size is 100 times). The second PMOS power transistor 686 is formed of a field effect transistor having a double-diffused P-channel MOS structure, and has a second power diode formed as a parasitic element from the current outflow terminal side to the current inflow terminal side. 686d are reversely connected in an equivalent circuit.

Similarly, the second power amplifier 617 is connected to the second power amplifier 617.
And a second field effect type power section current mirror circuit including a PMOS power transistor 687 and a PMOS transistor 697, and amplifies the output current H3 of the second current amplifier 647 inputted to the conduction control terminal side in a predetermined manner. And output (the cell size is 100 times). The second PMOS power transistor 687 is formed of a field effect transistor having a double-diffused P-channel MOS structure, and has a second power diode formed as a parasitic element from the current outflow terminal side to the current inflow terminal side. 687d are reversely connected in an equivalent circuit.

Second PMOS type power transistor 68
5, 686, 687 are commonly connected to the positive output terminal side of the voltage converter 52 via the resistor 31;
Each current outflow terminal side is connected to each power supply terminal of the coils 2, 3, and 4. Thereby, the second power amplifier 6
Reference numerals 15, 616, 617 respectively output amplified currents of the input currents to the respective energization control terminals to the respective power supply terminals of the coils 2, 3, 4 and drive currents I1, I2 to the coils 2, 3, 4 respectively. , I3.

The second distribution current signals G1, G2, G3 of the second distributor 38 in FIG. 18 are input to the second current amplifiers 645, 646, 647, respectively. The second current amplifiers 645, 646, and 647 respectively amplify the second distribution current signals G1, G2, and G3 by a predetermined amount to generate second amplified current signals H1, H2, and H3, and generate a second power The power is supplied to the respective energization control terminals of the amplifiers 615, 616, and 617. Second power amplifiers 615, 616, 617
Amplifies the current of the three-phase second amplified current signals H1, H2, H3, respectively, and outputs the coils 2, 3,
4 are supplied with the positive currents of the drive currents I1, I2 and I3.

FIG. 19 shows second current amplifiers 645 and 64.
6,647 shows a specific configuration. Second current amplifier 6
45 is a current mirror circuit of the preceding stage by transistors 651 and 652, transistors 653 and 654 and a resistor 6
A current mirror circuit at the subsequent stage by 55,656,
It is composed of a second amplifier current mirror circuit in which the current mirror circuits of the preceding and succeeding stages are cascaded. The emitter area ratio of the transistors 651 and 652 is set to 1,
The current amplification factor of the current mirror circuit in the preceding stage is set to 1. The emitter area ratio of the transistors 653 and 654 is 5
The current amplification ratio of the current mirror circuit at the subsequent stage is set to 50 times by making the resistance ratio between the resistors 656 and 655 50 times. As a result, the second amplifying section current mirror circuit of the second current amplifier 645 amplifies the current amplification factor by 50 times.

Similarly, the second current amplifier 646 includes transistors 661, 662, 663, 664 and a resistor 66.
5,666, which is a second amplifier current mirror circuit, and amplifies the current amplification factor by 50 times. Similarly, the second current amplifier 647 includes transistors 671, 672, 673, 674 and resistors 675, 676.
, And amplifies the current amplification factor by 50 times. As a result, the second current amplifiers 645, 646, and 647 output the three-phase second distribution current signals G1, G2, and G3 by 50, respectively.
Double amplified three-phase second amplified current signals H1, H2, H
3 and the second power amplifiers 615, 616, 6
The power is supplied to each energization control terminal side of the 17th second power section current mirror circuit.

The second current amplifiers 645, 646,
In the second amplifying unit current mirror circuit 647, the current outflow terminal side of the output NPN transistors 654, 664, 674 is connected to the negative terminal side of the DC power supply 50, and the output transistors 654, 664, 674 are connected. The current is supplied to the second power amplifiers 615, 616, and 617 via the respective power amplifiers. Thereby, the second power amplifier 6
15, 616, 617 second PMOS type power transistors 685, 686, 687 are sufficiently controlled to be energized.

Next, the overall operation of the motor shown in FIG. 18 will be described. The switching creator 34 outputs a smoothly changing 3
The phase switching current signals D1, D2, D3 are supplied to the distribution generator 36. The first distributor 37 is connected to the first
Of the three-phase switching current signals D1, D2,
The first distribution current signals E1, E2, and E3 of three phases that distribute in response to D3 and change smoothly are output. The first current amplifiers 41, 42, and 43 output first amplified current signals F1, F2, and F3 obtained by amplifying the first divided current signals E1, E2, and E3 by a predetermined number, respectively, and output the first power The power is supplied to the respective energization control terminals of the amplifiers 11, 12, and 13. The first field-effect-type power section current mirror circuits of the first power amplifiers 11, 12, and 13 current-amplify the first amplified current signals F1, F2, and F3, respectively, and perform three-phase coils 2, 3, and 4 respectively. Are supplied with the negative currents of the drive currents I1, I2 and I3.

The second distributor 38 converts the second supply current signal C2 of the current supply 30 into three-phase switching current signals D1, D
2, and distributes in response to D3, and outputs three-phase second distribution current signals G1, G2, and G3 that change smoothly. The second current amplifiers 645, 646, and 647 respectively output second amplified current signals H1, H2, and H3 obtained by amplifying the second divided current signals G1, G2, and G3 by a predetermined number, and output the second power The power is supplied to the respective energization control terminals of the amplifiers 615, 616, and 617. Second power amplifiers 615, 616, 6
The second field effect type power section current mirror circuit 17 amplifies the currents of the second amplified current signals H1, H2, and H3, respectively, and supplies drive currents I1, I2 to the three-phase coils 2, 3, and 4.
2 and I3 are supplied.

The current detecting resistor 31 of the current supplier 30
Detects a combined supply current Iv that is a combined value of the positive side currents of the drive currents I1, I2, and I3, and generates a current detection signal Bj corresponding to the combined supply current Iv via the level conversion circuit 32. The supply output unit 33 includes a first supply current signal C1 and a second supply current signal C2 responsive to the current detection signal Bj.
Is output. The first supply current signal C1 and the second supply current signal C2 change in proportion or substantially in proportion to the combined supply current Iv.

The first distribution control block (current supply 30
And the first distributor 37 and the first current amplifiers 41, 42, 4
3) is a three-phase first amplified current signal F1, F in which the rising slope portion and the falling slope portion smoothly change.
2 and F3 (a first three-phase current signal), and outputs the first amplified current signals F1, F2 and F3 to the first power amplifier 1
1, 12 and 13 are supplied to the respective energization control terminals. This ensures that at least one of the three first NMOS power transistors 81, 82, 83 performs a resistive voltage drop operation (full-on operation) while maintaining the three first NMOS power transistors 81, 82, 83. Transistors 81, 82, 8
3 smoothly switches the current path.

Therefore, each first NMOS power transistor smoothly changes the current value by performing a current amplifying operation (half-on operation) in the active region in the first period and the last period of the current path switching. Perform a resistive voltage drop operation in a period after the current path is formed. As a result, no spike voltage is generated due to the switching of the current path, and the drive currents I1, I2, and I3 to the coils 2, 3, and 4 change smoothly.

The second distribution control block (current supply 30
, The second distributor 38 and the second current amplifiers 645, 64
6,647) generates three-phase second amplified current signals H1, H2, and H3 (second three-phase current signals) in which the rising slope portion and the falling slope portion smoothly change, and generates the second amplified current signal. The current signals H1, H2, H3 are supplied to the respective power supply control terminals of the second power amplifiers 615, 616, 617. This ensures that at least one of the three second PMOS power transistors 685, 686, 687 performs a resistive voltage drop operation (full-on operation), while ensuring that the three second PMOS power transistors The switching operation of the current path by the transistors 685, 686, and 687 is smoothly performed.

Accordingly, each of the second PMOS type power transistors performs a current amplifying operation (half-on operation) in the active region in the first period and the last period of the switching of the current path to smoothly change the current value. Perform a resistive voltage drop operation in a period after the current path is formed. As a result, no spike voltage is generated due to the switching of the current path, and the drive currents I1, I2, and I3 to the coils 2, 3, and 4 change smoothly.

A positive feedback loop is formed by the first distribution control block and the first power amplifiers 11, 12, and 13, and positive feedback is formed by the second distribution control block and the second power amplifiers 615, 616, and 617. A loop is formed. As a result, at least one of the three field-effect power transistors 81, 82, and 83 reliably operates a resistive voltage drop, and the three second PMOS-type power transistors 685 and 68.
Out of 6,687, at least one power transistor is reliably operated in a resistive voltage drop operation. This stabilizes the overall circuit operation.

Since the first amplified current signal F1 and the second amplified current signal H1 having the same phase flow complementarily with a phase difference of 180 °, the first power amplifier 11 and the second power amplifier 615 are complementary. Works. Therefore, the drive current I1 in both directions, which changes smoothly and continuously, is supplied to the coil 2. Similarly, since the first amplified current signal F2 and the second amplified current signal H2 flow complementarily with a phase difference of 180 °, the first power amplifier 12 and the second power amplifier 616 operate complementarily. . Therefore, the drive current I2 in both directions, which changes smoothly and continuously, is supplied to the coil 3.

Similarly, since first amplified current signal F3 and second amplified current signal H3 flow complementarily with a phase difference of 180 °, first power amplifier 13 and second power amplifier 617 are complementary. Works. Therefore, the drive current I3 in both directions, which changes smoothly and continuously, is supplied to the coil 4. Thus, the first power amplifier and the second
Are not energized at the same time, the first NMOS power transistor and the second PM
No short-circuit current occurs between the OS type power transistors.

Further, since the continuous drive currents I1, I2, and I3 that change smoothly are supplied to the coils 2, 3, and 4,
No spike voltage is generated in coils 2, 3, and 4.
First power diodes 81d and 82 as parasitic elements
d, 83d and the second power diodes 685d, 686
d, 687d does not flow. Therefore, the pulsation of the driving force generated by the motor is significantly reduced.

The voltage converter 52 operates the NMOS type switching transistor 61 at a high frequency PWM, and converts the converted DC voltage (Vcc-V) between the positive output terminal side and the negative output terminal side.
Vg) is variably controlled. The operation controller 310 outputs an operation control signal Vd corresponding to the command signal Ad and the modulated current signal Pm of the modulator 300. The switching operation of the NMOS switching transistor 61 is controlled in response to the output signal Vd of the operation controller 310, and the converted DC voltage (Vcc-Vg) of the voltage converter 52 is variably controlled. Therefore, the converted DC voltage (Vcc-Vg) of the voltage converter 52
Changes in response to the modulated current signal Pm of the modulating unit 300 to reduce the pulsation of the driving force.

Further, when the DC power supply 50 is turned off in an emergency, the three-phase counter electromotive force generated in the coils 2, 3, and 4 is converted into the first power diodes 81d, 82d, 83d and the second power diodes 685d, The rectified DC voltage rectified through 686d and 687d is supplied to terminal X of voltage extractor 490.
Output to f. Using the output voltage of the voltage extractor 490, various emergency evacuation processes are performed.

In the present embodiment, a motor configuration suitable for integration into an integrated circuit is obtained by joining and separating. First, heat generation of the power element was reduced to prevent thermal destruction when the power transistor and the switching transistor were integrated. Further, a field effect transistor having a double-diffused MOS structure was used as the first power transistor and the second power transistor, and the chip size was reduced. Further, a parasitic diode formed from the current outflow terminal side to the current inflow terminal side of the power transistor is used as the power diode, and the chip area for the power diode is made substantially zero.

Further, a second PMOS power transistor or a PMO
The use of S-type power path transistors eliminates the need for a separate power supply to operate these power elements (requires capacitors, etc., and eliminates the high voltage output that consumes power). This eliminates the need for a separate power supply other than the DC power supply 50 and the voltage converter 52, and significantly reduces the overall configuration.

The NMOS switching transistor whose current outgoing terminal is connected to the negative terminal of the DC power supply performs PWM operation, and a converted DC voltage is obtained by the NMOS switching transistor and the conversion inductor element. As a result, the potentials on the current inflow terminal side and the current outflow terminal side of the NMOS switching transistor do not become lower than the negative terminal side potential (ground potential) of the DC power supply 50. As a result, the operation of the parasitic transistor element having the junction isolation portion as the base terminal can be prevented, and the entire circuit operation has been stabilized.

Further, in the present embodiment, the current supply unit outputs the first supply current signal C1 and the second supply current signal C2 in response to the combined supply current Iv. By using the first amplified current signals F1, F2, and F3 responsive to the first supply current signal C1, energization of the first power section current mirror circuits of the first power amplifiers 11, 12, and 13 is controlled. 2 in response to the second supply current signal C2.
The power supply to the second power section current mirror circuits of the second power amplifiers 15, 16, 17 is controlled using H3. Accordingly, even when the combined supply current Iv to the coil changes in response to the command signal Ad, the current path switching operation can be smoothly performed in an analog manner, and the driving accompanying the current path switching can be performed. The pulsation of the current and the pulsation of the driving force are extremely small.

Further, in this embodiment, the first power amplifiers 11, 12, and 13 are constituted by a first field-effect type power section current mirror circuit including first NMOS type power transistors, and the second power amplifier 615 , 616
617 is constituted by a second field effect type power section current mirror circuit using a second PMOS type power transistor to reduce the variation of the current amplification factor.

Normally, an NMOS power transistor and P
MOS power transistors have significantly different nonlinear voltage amplification characteristics. However, in this embodiment, the variation in the current amplification factor of the first and second field effect type power section current mirror circuits can be significantly reduced.

Therefore, at least in the rising slope portion and the falling slope portion, the first portion which smoothly changes.
Are supplied to the respective conduction control terminals of the three first power amplifiers 11, 12, and 13 at least in the rising slope portion and the rising slope portion. The second amplified current signals H1, H2, and H3 (second three-phase current signals) that smoothly change in the falling slope portion are converted into three second power amplifiers 615 and 615.
The current path switching operation is performed by the three first NMOS power transistors 81, 82, 83 and the three PMOS power transistors 685, 686, 687 by supplying the current to the respective conduction control terminals 616, 617. Can be smoothed. Further, at least one of the three first NMOS power transistors can reliably perform a resistive voltage drop operation, and at least one of the three second PMOS power transistors can reliably perform a resistive voltage drop. Voltage drop operation is possible. Therefore, the pulsation of the driving force is small,
A motor with low heat generation can be realized.

In this embodiment, when the DC power supply 50 is turned off, a rectified DC voltage obtained by rectifying the three-phase back electromotive force generated in the coil via the first power diode and the second power diode. , And a voltage extractor to the terminal Xf. By using the output voltage of the voltage extractor, it is possible to electrically save the contents of the memory of the microcomputer and to mechanically move the reading head when the emergency DC power is turned off. Further, in the present embodiment, the first power amplifiers 11, 12, and 13 are not limited to the configuration shown in FIG. 18, and various modifications are possible. For example, instead of each of the first power amplifiers 11, 12, and 13, the power amplifier 1000 having the configuration shown in FIG. 22 described above may be used. Further, for example, power amplifier 1100 having the configuration shown in FIG. 23 described above may be used.

Further, in the present embodiment, the second power amplifiers 615, 616, and 617 are not limited to the configuration shown in FIG. 18, and various modifications are possible. For example, a power amplifier 1200 having the configuration shown in FIG. 24 may be used instead of each of the second power amplifiers 615, 616, and 617. The power amplifier 1200 comprises a PMOS power transistor 1210 and a PMOS transistor 121
1 and the resistor 1212 constitute a field effect type power section current mirror circuit. In the field effect type power section current mirror circuit 1200, the control terminal of the field effect type power transistor 1210 is connected to the field effect type transistor 1210.
11 (directly or through some element), and one terminal side of the current path terminal pair of the field effect transistor 1211 is connected to one terminal side of the current path terminal pair of the field effect power transistor 1210. Resistance 121
2 and a field effect transistor 1211
The other terminal side of the current path terminal pair
00 (directly or through some element) and the field effect transistor 121
1 is connected (directly or through some element) to the power supply control terminal side of the power amplifier 1200. This field effect type power section current mirror circuit has a relatively large current amplification factor.

Further, for example, power amplifier 1300 having the configuration shown in FIG. 25 may be used. Power amplifier 1
300 is a PMOS power transistor 1310 and PM
The OS-type transistor 1311 and the resistor 1312 constitute a field-effect power section current mirror circuit.

Field effect type power section current mirror circuit 1
Reference numeral 300 denotes a control terminal of the field-effect power transistor 1310 connected to the control terminal of the field-effect transistor 1311 (directly or through some element);
One terminal side of the current path terminal pair of the field effect transistor 1311 is connected to the conduction control terminal side of the power amplifier 1310 via the resistor 1312, and the other terminal side of the current path terminal pair of the field effect transistor 1311 is connected to the field effect transistor. Type power transistor 1310 is connected (directly or through some element) to one terminal side of a current path terminal pair, and the control terminal side of the field effect transistor 1311 is connected to the conduction control terminal side of the power amplifier 1310 (directly or through some element). (Via elements).

This field effect type power section current mirror circuit has a predetermined current amplification factor while the input current to the conduction control terminal side is small, and the current amplification factor sharply increases as the input current increases. . The PMOS power transistor 1210 and the PMOS power transistor 1
Reference numeral 310 denotes a field effect type power transistor having a double-diffused P-channel MOS structure, which can be easily integrated.

Further, the operation controller and the voltage converter are not limited to the above-mentioned configuration. Further, the operation controller and the voltage converter are eliminated, and the first power amplifier and the first power amplifier of the second power amplifier are eliminated. The field-effect power transistor and the second field-effect power transistor can be directly operated at a high frequency to perform the role of the switching transistor of the voltage converter. Further, in this embodiment, various advantages similar to those of the above-described embodiment can be obtained.

Embodiment 5 FIG. 20 shows a motor according to Embodiment 5 of the present invention. FIG. 20 shows the overall configuration. In the fifth embodiment, a shunt switch device 700 and an energization stop device 701 are provided.
A first energizing mode for supplying a bidirectional driving current to the coils 2, 3, and 4 and a second energizing mode for supplying a unidirectional driving current to the coils 2, 3, and 4 can be switched and supplied as appropriate. did. Also, a voltage extractor 74 is provided so as to extract a rectified DC voltage from the common connection terminal side of the coils 2, 3, and 4.
0 was changed. In another configuration, the fourth embodiment is described.
Alternatively, the same components as those of the third embodiment, the second embodiment, or the first embodiment are denoted by the same reference numerals, and the detailed description is omitted.

First, the shunt switch 700 is turned off.
A case where the power stoppage device 701 is not stopped (first power supply mode) will be described. That is, the shunt switch 700
The first switch unit 711 is off, and the second switch unit 721 and the third switch unit 731 of the power supply stop unit 701 are on. Since the first switch section 711 is off, the PMOS shunt transistor 710 of the shunt switch 700 is off. The PMOS shunt transistor 710 is constituted by a field effect transistor having a double-diffused P-channel MOS structure, and a diode 710d formed as a parasitic element thereof has an equivalent circuit from the current outflow terminal side to the current inflow terminal side. Reverse connection.

Second switch section 72 of energization stop device 701
Since 1 is on, the second supply current signal C2 of the current supply 30 is supplied to the second distributor 38. Since the third switch unit 731 of the power supply stop unit 701 is on, P
The MOS conduction transistor 730 is turned on to supply current to the second power amplifiers 615, 616, and 617.

The PMOS conduction transistor 730 is reversely connected, and its current inflow terminal is connected to the positive terminal of the DC power supply 50 via the resistor 31 and the power path switch 54.
The current outflow terminal side is connected to the second power amplifiers 615, 61
6,617 are connected to the common connection terminal side. PMOS
Transistor 730 is a double diffused P-channel MO
It is composed of an S-structure field effect transistor, and has a PM
Since the OS-type conducting transistor 730 is reversely connected, the diode 730d formed as a parasitic element thereof
Are connected in an equivalent circuit from the current inflow terminal side to the current outflow terminal side.

Therefore, when the first switch unit 711 of the shunt switch device 700 is turned off and the second switch unit 721 and the third switch unit 731 of the conduction stop unit 701 are turned on, the above-described operation is performed. The configuration is substantially the same as that of Example 4. Therefore, the first and second power amplifiers 11, 12, and 13 and the second power amplifiers 615, 616, and 617 supply the coils 2, 3, and 4 with drive currents I1, I2, and I3 in both positive and negative directions. (First energization mode). The specific configuration and operation are the same as those in the fourth embodiment, and a detailed description thereof will be omitted.

Next, the shunt switch 700 is turned on,
A case where the power supply stop device 701 is stopped (second power supply mode) will be described. That is, the first switch unit 711 of the shunt switch device 700 is turned on, and the power supply stop device 701 is turned on.
In this case, the second switch unit 721 and the third switch unit 731 are turned off. Since the first switch section 711 is on, the PMOS shunt transistor 710 of the shunt switch 700 turns on. Therefore, the voltage converter 5
2 and a current detection resistor 31 and a PMO
Coil 2, 3, 4 via S type shunt transistor 710
A current path to the common connection terminal side is formed.

On the other hand, the second supply current signal C 2 of the current supply 30 is not supplied to the second distributor 38 because the second switch section 721 of the power supply stop 701 is off. Therefore, the second distribution current signals G1, G2, G3 and the second
Of the amplified current signals H1, H2 and H3 become zero. As a result, the second power amplifiers 615, 616, 617
PMOS power transistors 685, 686, 68
7 are all off. Further, the third switch unit 73
Since 1 is off, the energizing transistor 730 is turned off, and the current supply to the second power amplifier is stopped.

At this time, the energizing transistor 730 and its parasitic diode 730d cut off the current path of the back electromotive force generated in the coils 2, 3, and 4, thereby preventing the formation of an unnecessary current path. Therefore, the current paths to the coils 2, 3, and 4 are connected to the first power amplifiers 11, 1 connected in parallel between the positive output terminal side and the negative output terminal side of the voltage converter 52.
2, 13 first NMOS power transistors 81,
82, 83 are formed. That is, coil 2,
Negative unidirectional drive currents I1, I2, I3 are supplied to 3, 4 (second energization mode). The resistance 712 of the shunt switch 700 and the resistance 732 of the
Is a pull-up resistor, which may be connected as needed.

The overall operation in the second energization mode will be described. The current supply 30 includes a first supply current signal C1 and a second supply current signal C2 responsive to the current detection signal Bj.
Is output. The second supply current signal C2 is supplied to the power supply stop 70
It is blocked by the first second switch unit 721 and is not supplied to the second distributor 38. Therefore, the second power amplifier 61
5, 616, 617 stop supplying electricity and do not supply current to the coils 2, 3, and 4.

On the other hand, the first supply current signal C 1 is supplied to the first distributor 37. The first distributor 37 distributes the first supply current signal C1 in response to the three-phase switching current signals D1, D2, and D3 of the switching generator 34, and smoothly changes the three-phase first distribution. It outputs current signals E1, E2, E3. The first current amplifiers 41, 42, and 43 respectively provide first amplified currents of the first distributed current signals E1, E2, and E3.
And outputs the amplified current signals F1, F2, and F3 to the first power amplifiers 11, 12, and 13 to the conduction control terminals. The first power section current mirror circuits of the first power amplifiers 11, 12, and 13 current-amplify and supply the first amplified current signals F1, F2, and F3 to the coils 2, 3, and 4, respectively. Thus, the first distribution control block (the current supply unit 30, the first distributor 37, and the first current amplifier 4
1, 42, 43), at least one of the three first NMOS power transistors 81, 82, 83 is operated by a resistive voltage drop.

The operation controller 310 outputs an operation control signal Vd in response to the command signal Ad. The voltage converter 52
In response to the operation control signal Vd, the NMOS switching transistor 61 performs a high-frequency switching operation. As a result, the converted DC voltage (Vcc-Vg) of the voltage converter 52
Is variably controlled. The converted DC voltage of the voltage converter 52 is
Three first power amplifiers 11, 1 connected in parallel
2, 13 and the coils 2, 3, 4 and the shunt switch 700, a current path for supplying a unidirectional drive current to the coil is formed by the first power transistor of the selected first power amplifier. You.

The voltage extractor 740 compares the potential on the positive terminal side of the DC power supply 50 with the potential on the common connection terminal side of the coil, and outputs the higher voltage. Thereby, the DC power supply 5
When 0 is turned off, a rectified voltage signal of three-phase back electromotive force generated in the coil is taken out to the terminal Xf. Other configurations and operations are the same as those in the above-described fourth embodiment, and a detailed description thereof will be omitted.

In the present embodiment, the first energizing mode for supplying a bidirectional driving current to the coil and the second energizing mode for supplying a unidirectional driving current to the coil are switched and supplied as needed to improve the motor performance. Made it possible to change. In the first energizing mode and the second energizing mode, power loss and heat generation in the first power transistor, the second power transistor, and the power elements such as the switching transistor, the shunt transistor, and the energizing transistor are small. Accordingly, these power elements can be bonded and separated on a single silicon substrate as required to form an integrated circuit.

Further, in the first energizing mode in which the drive current is supplied to the coil in both directions, there is an advantage that the generated force can be increased. In the second energizing mode for supplying a one-way driving current to the coil, the back electromotive force generated in the coil can be increased, so that there is an advantage that the motor can be rotated at high speed. Therefore, it is possible to realize a motor having a large generating force and capable of rotating at high speed.

The second power transistors 685, 686, and 687 formed as integrated circuits are reversely connected to second power diodes 685d, 686d, and 687d, which are parasitic elements. When the unidirectional drive current is supplied to the coils 2, 3, and 4 by turning on the shunt transistor 710, the potential at the power supply terminal side increases in an alternating current due to the back electromotive force generated in the coils, and the second power Diode 685
The current tends to flow backward through d, 686d, and 687d. However, in the present embodiment, since the energizing transistor 730 is reversely connected, the reverse current path can be reliably shut off by turning off the energizing transistor 730.

It is to be noted that the conducting transistor 730 is not limited to a PMOS transistor connected in reverse, but may be an NMOS transistor connected in reverse.
It can also be configured by an S-type transistor. In these reverse-connected field-effect transistors, a parasitic diode is formed from the current inflow terminal side to the current outflow terminal side, and the parasitic diode also prevents reverse current flow when the energizing transistor is turned off.

Also, in this embodiment, various advantages similar to those of the above embodiments can be obtained. In the present embodiment, the first power amplifiers 11, 12, and 13, the second power amplifiers 615, 616, and 617, the current supply 30, the switching generator 34, and the distribution generator 36 (the first distributor 37) And the second distributor 38) and the first current amplifiers 41, 42, 43
, A second current amplifier 645, 646, 647, an operation controller 310, a voltage converter 52, a shunt switch 700, an energization stop 701, and a voltage extractor 740. ) To form a drive circuit for supplying a drive current to the drive circuit.

A high-voltage output device is provided, and NMOS transistors are used for the second power transistors 685, 686, 687, the shunt transistor 710, and the energizing transistor 730. These transistors are used from the high potential point of the high-voltage output device. It is also possible to control the energization of the elements.

Embodiment 6 FIG. 21 shows a motor according to Embodiment 6 of the present invention. FIG. 21 shows the overall configuration. In the sixth embodiment, as the voltage converter 752, for example, the high-voltage output device 4
A voltage conversion having a configuration such as 50 is performed. In other configurations, the same components as those in the above-described fifth embodiment, the fourth embodiment, the third embodiment, the second embodiment, or the first embodiment are denoted by the same reference numerals, and detailed description is omitted.

The voltage converter 752 shown in FIG.
An NMOS switching transistor 761 that performs a high-frequency switching operation of about z is provided. The NMOS-type switching transistor 761 has its current outflow terminal side connected to the negative terminal side (−) of the DC power supply 50, is connected to its current inflow terminal side and one end of the conversion inductor element 763, and has the positive terminal of the DC power supply 50. The power supply path for replenishing magnetic energy from the side (+) to the conversion inductor element 763 performs high-frequency switching (on / off operation). The flywheel diode 762 connected to one end of the conversion inductor element 763 turns off and on in a complementary manner to the on / off high-frequency switching operation of the NMOS switching transistor 761, and the conversion inductor element 763 outputs the conversion capacitor. A circuit for forming a current path to the circuit including the element 764 is formed.

That is, when the NMOS switching transistor 761 is on, a power supply path is formed from the positive terminal of the DC power supply 50 through the conversion inductor element 763 to replenish the magnetic energy of the conversion inductor element 763. NMOS switching transistor 76
1 changes to the off state, the terminal voltage of the conversion inductor element 763 rapidly increases, and the flywheel diode 762 changes to the conductive state.
A current path from 3 to the circuit side including the conversion capacitor element 764 is formed. Thereby, the conversion capacitor element 7
DC voltage V between one end of DC power supply 64 and one end of DC power supply 50
Output m.

The conversion capacitor element 764 is connected between the positive output terminal side (P) and the negative output terminal side (M) of the voltage converter 752, and is supplied with a current / voltage supplied via the conversion inductor element 763. Are formed in a filter circuit for smoothing the signal. As a result, the potential Vm on the positive output terminal side of the voltage converter 752 is variably controlled by performing the high-frequency PWM operation (pulse width modulation operation) on the NMOS switching transistor 761. As a result, using the DC voltage Vcc supplied from the DC power supply 50 as a power supply source, a converted DC voltage value Vm is created between the positive output terminal side and the negative output terminal side of the voltage converter 752. Here, the negative terminal of the DC power supply 50 is set to the ground potential (0 V).

The NMOS switching transistor 761 is formed of, for example, an N-channel field effect transistor having a double diffusion MOS structure, and has a switching diode formed as a parasitic element from the current outflow terminal side to the current inflow terminal side. 761d are reversely connected in an equivalent circuit. The operation controller 310 outputs an operation control signal Vd in response to the command signal Ad. The voltage converter 752 controls the operation control signal Vd in the PWM unit 765.
A predetermined high-frequency PWM signal Sw having a pulse width corresponding to the
1 is subjected to a high-frequency switching operation. That is, the voltage converter 7 responds to the operation control signal Vd of the operation controller 310.
52 of the NMOS switching transistor 761
The WM switching operation is controlled.

The voltage converter 752 uses the DC voltage Vcc of the DC power supply 50 as a power supply source, and converts the converted DC voltage Vm responsive to the PWM switching operation of the NMOS switching transistor 761 between the positive output terminal side and the negative output terminal side. Output to Note that the specific configuration of the PWM unit 765 of the voltage converter 752 is the same as that shown in FIG. 7 described above, and detailed description will be omitted.

The voltage extractor 790 of FIG. 21 includes an output diode 791. The output diode 791 has an input terminal side connected to the second PMOS power transistor 685,
686, 687 are connected to the current inflow terminal side, and the output terminal side is connected to the output terminal Xf of the voltage extractor 790. Thus, when the DC power supply 50 is turned off, the voltage extractor 790 obtains a rectified DC voltage obtained by rectifying the back electromotive force of the coils 2, 3, and 4 at the output terminal Xf via the output diode 791.

At this time, when the DC power supply 50 is turned off, the flywheel diode 762 of the voltage converter 752 reversely flows from the coil side to the DC power supply 50 due to the back electromotive force of the three-phase coils 2, 3, and 4. You are blocking the current. Accordingly, the flywheel diode 762 substantially also serves as the power path switch 54 shown in FIGS. Therefore, various protection operations can be performed using the rectified DC voltage of the voltage extractor 790 when the DC power supply 50 is turned off.

Other configurations and operations are the same as those of the fifth embodiment.
The detailed description is omitted. In this embodiment,
The motor configuration is suitable for integration into an integrated circuit. First, heat generation of the power element was reduced to prevent thermal destruction when the power transistor and the switching transistor were integrated. Further, a field effect transistor having a double diffusion MOS structure is used as the first power transistor, the second power transistor, and the switching transistor,
Reduced chip size.

Also, the NMOS switching transistor whose current outgoing terminal is connected to the negative terminal of the DC power supply performs PWM operation, and a converted DC voltage is obtained by the NMOS switching transistor and the conversion inductor element. As a result, the potentials on the current inflow terminal side and the current outflow terminal side of the NMOS switching transistor do not become lower than the negative terminal side potential (ground potential) of the DC power supply 50. As a result, the operation of the parasitic transistor element having the junction isolation portion as the base terminal can be prevented, and the entire circuit operation has been stabilized.

Further, a second PMOS power transistor, a PMOS shunt transistor, or a PMOS conduction transistor is used for the second power amplifier, the shunt switch, and the conduction stop device, and these power elements are operated. To eliminate the need for a separate power supply (no need for a high-voltage output device). In this embodiment, the motor performance is changed by appropriately switching between a first energizing mode for supplying a bidirectional driving current to the coil and a second energizing mode for supplying a unidirectional driving current to the coil. Made it possible. Even when such switching is performed, power loss and heat generation in power elements such as the first power transistor, the second power transistor, the switching transistor, the shunt transistor, and the conduction transistor are small. Therefore, these power elements can be integrated on a single silicon substrate as required.

In this embodiment, the DC power supply 50 and the voltage converter 752 form a voltage supply block for supplying a required converted DC voltage Vm between the positive output terminal side and the negative output terminal side of the voltage converter. doing. This voltage supply block converts the converted DC voltage Vm to the output DC voltage V
cc, and the supply voltage level to the coils 2, 3, and 4 was increased. Thereby, high-speed rotation of the motor can be easily realized. When the motor is started, for example, the switching transistor 761 is turned off, and the output DC voltage Vcc of the DC power supply 50 is output via the flywheel diode 762, and the voltage converter 7
52 may be the converted DC voltage.

In the present embodiment, the voltage converter 75
The flywheel diode 762, which is a current path forming circuit of No. 2, may form a current path from the conversion inductor element 763 to the circuit including the conversion capacitor element 764 when the switching transistor 761 is off, and may be variously modified. Is possible. For example, the NMO having the configuration shown in FIG.
The S-type synchronous rectifier transistor 1400 is used, and a signal from the PWM unit 765 complements the on / off switching operation of the switching transistor 761 by NM.
The OS-type synchronous rectification transistor 1400 can be turned on and off.

NMOS synchronous rectification transistor 1400
Is constituted by a field effect type power transistor having a double-diffused N-channel MOS structure, and a parasitic diode 1400d is reversely connected between the current input and output terminals of the NMOS type synchronous rectification transistor 1400. With the parasitic diode 1400d, the effect of the flywheel diode can also be obtained.

Also, for example, the P of the configuration shown in FIG.
Using MOS type synchronous rectification transistor 1500, PW
By the signal from the M unit 765, the PMOS synchronous rectification transistor 1500 can be turned on / off in a complementary manner to the on / off switching operation of the switching transistor 761. PMOS
The synchronous rectifier transistor 1500 is constituted by a field effect type power transistor having a double-diffused P-channel MOS structure, and a parasitic diode 1500d is reversely connected between the current input and output terminals of the PMOS synchronous rectifier transistor 1500. The effect of the diode for the flywheel can be obtained by the parasitic diode 1500d.

Further, the diode 76 for the flywheel
PMOS type synchronous rectification transistor 150 in which 2 is replaced
0 indicates that when the DC power supply 50 is turned off, the reverse current generated from the coil side toward the DC power supply 50 due to the back electromotive force of the coil can be prevented. It serves as a path transistor 72. Further, a new high potential point is not required to operate the PMOS synchronous rectification transistor 1500 (no high voltage output device is required), and the configuration of the motor is simplified.

Further, in this embodiment, the first supply current signal C1 responding to the combined supply current Iv by the current supply 30 is provided.
And the second supply current signal C2. First amplified current signals F1, F2, F responsive to the first supply current signal C1
3 and the first of three phases that smoothly change at least in the rising slope portion and the falling slope portion.
To the first power amplifiers 11, 12, 13
To generate the second amplified current signals H1, H2, and H3 in response to the second supply current signal C2, at least in the rising slope portion and the falling slope portion. The second amplified current signal is supplied to the conduction control terminals of the second power amplifiers 615, 616, and 617. This allows the three first
NMOS power transistors 81, 82, 83 and three second PMOS power transistors 685,
The switching operation of the current paths to the coils 2, 3, and 4 by the 686 and 687 could be performed smoothly.

Therefore, the drive currents I1, I2, and I3 to the coils 2, 3, and 4 change smoothly, and the pulsation of the drive current and the pulsation of the drive force due to the switching of the current path become extremely small. Such an effect can be obtained even in the case of unidirectional current supply with the shunt transistor turned on. Also, the current supply unit 950 shown in FIG. 28 is used in place of the current supply unit 30 in FIG. 21, and the first supply current signal C1 and the second supply current signal C2 are changed in direct response to the command signal Ad. It is also possible to make it. In addition, in this embodiment,
Various advantages similar to those of the above embodiment can be obtained.

The specific configuration of each of the above embodiments can be variously modified. For example, each phase coil may be configured by connecting a plurality of partial coils in series or in parallel. The three-phase coils are not limited to the star connection, but may be a delta connection. Generally, a motor having a single-phase or multi-phase coil can be configured. Further, the field portion of the moving body is not limited to the illustrated one. The number of magnetic poles is not limited to two, and a multi-pole configuration is generally possible.

Further, the field portion may be any as long as it supplies a magnetic flux, which changes with the moving operation of the moving body, to the coil.
Various known configurations are possible. According to the present invention,
Brushless motors, permanent magnet field type stepping motors, reluctance type stepping motors, hybrid type stepping motors, and other various motors can be configured, and it goes without saying that they are included in the present invention. Further, the moving body is not limited to the rotational movement, and may move straight.
Further, the operation controller and the voltage converter are not limited to the above-described configuration. Further, the functions of the operation controller and other required functions may be executed digitally by a microprocessor.

Also, for example, a motor having a single-phase coil such as a focus actuator or a tracking actuator of a disk drive can be configured. The motor having such a single-phase coil may have a configuration in which, for example, the first power amplifier 13, the second power amplifier 17, and the coil 4 in FIG. At this time, coil 3
Is replaced by a zero ohm resistor, the first amplified current signals F1 and F2 change complementarily, and the second amplified current signal H1
And H2 change complementarily. Therefore, if a single-phase switching signal is generated in the switching generator, the current path switching to the single-phase coil can be smoothly performed.

[0229] In the formation of an integrated circuit, various one-chip integrated circuit technologies based on a known semiconductor process can be used. For example, there are various one-chip integrated circuit technologies that can use a single type or a plurality of types of double diffusion MOS type field effect transistors, CMOS type field effect transistors, and bipolar transistors. In any case, the substrate of the integrated circuit is connected to the potential (earth potential) on the negative terminal side of the DC power supply, and a high-density integrated circuit can be realized. Since the specific transistor arrangement in one chip differs depending on the design of each integrated circuit, a detailed description is omitted.

Further, the first power amplifier and the second power amplifier may forcibly turn off the conduction control terminal during the off period. That is, the power control terminal side of the power amplifier which is turned off by the off signal is electrically connected to the negative terminal side or the positive terminal side of the DC power supply via a resistor or the like, and the power amplifier is reliably turned off.

Further, the first power amplifier and the second power amplifier are not limited to the configuration shown in the above-described embodiment, and various modifications may be made as long as the operation substantially follows the gist of the present invention. It is possible. In the above-described example, a power amplifier having a power section current mirror circuit using a field-effect power transistor has been described as a preferable example, but the present invention is not limited to such a configuration. For example, IGBT transistor (InsulatedGate bipolar Transistor) or COMFET transistor (Conductivity modulated)
Field Effect Transistor) is a composite power transistor having a non-linear voltage amplification characteristic, and is used as an on / off switching element because of its large variation in amplification characteristic.

However, since the IGBT transistor is a composite field-effect power transistor having a field-effect transistor on the input side, a field-effect power section current mirror circuit using the IGBT transistor can be formed. Can be used to configure a power amplifier having a current amplification characteristic. By supplying a current signal that smoothly changes at least in the rising slope portion and the falling slope portion to the power supply control terminal side of such a power amplifier, the current path can be switched smoothly.

As a result, although the compound field effect transistor has many disadvantages (large on-voltage, large variation in amplification gain), it is possible to obtain various effects shown in the present invention. Therefore, the field-effect power transistor of the present invention includes an IGBT transistor or a composite field-effect transistor having a field-effect transistor on the input side. FIG. 30 shows an example of a power amplifier 1900 using a composite field-effect power transistor 1910 having a field-effect transistor on the input side such as an IGBT transistor. The connection of the composite field effect transistor 1910 and the field effect transistor 1911 equivalently constitutes a field effect power section current mirror circuit. Thereby, the power amplifier 1
900, amplify the input current to the conduction control terminal side of 900,
A driving current is output to a conduction current path of the composite field effect transistor 1910.

The power diode 1910d is a parasitic diode reversely connected in an equivalent circuit in parallel to the current path of the composite field effect transistor 1910. It is to be noted that the composite field effect transistor 1910 in the ON state performs a resistive voltage drop operation including a bias value of a required voltage. FIG. 31 shows another example of a power amplifier 1950 having a composite field-effect power transistor 1960 such as an IGBT transistor. Further, the field effect switching transistor of the present invention has an IGB
A composite field effect transistor having a T transistor or a field effect transistor on the input side is also included.

Also, the DC power supply 50 shown in the above-described embodiment is used.
Various configurations are possible as long as they can supply a DC voltage or a DC current. For example, a battery power supply or a rectified power supply for an AC line is used. Further, according to the present invention, a high-performance disk device with small vibration can be realized. In addition, various modifications are possible without changing the gist of the present invention, and it goes without saying that the present invention is included in the present invention.

[0236]

According to the motor of the present invention, the power loss of the power transistor is greatly reduced by the operation of the voltage converter. This results in a motor with high power efficiency. Further, since the current path to the coil is smoothly switched by the first power transistor and the second power transistor, the pulsation of the driving current is significantly reduced. This results in a motor with low vibration. In addition, malfunctions due to parasitic transistors when integrated are prevented.
As a result, a highly reliable and low-cost motor can be realized.

[Brief description of the drawings]

FIG. 1 is a diagram illustrating an entire configuration according to a first embodiment of the present invention.

FIG. 2 is a circuit diagram of a switching generator according to the first embodiment.

FIG. 3 is a circuit diagram of a control generator 30 according to the first embodiment.

FIG. 4 is a circuit diagram of a distribution creator according to the first embodiment.

FIG. 5 shows first current amplifiers 41 and 4 in the first embodiment.
FIGS. 2 and 43 are circuit diagrams.

FIG. 6 shows second current amplifiers 45 and 4 in the first embodiment.
6 and 47 are circuit diagrams of the high voltage output device 53.

FIG. 7 is a circuit diagram of an operation controller 51 and a voltage converter 52 according to the first embodiment.

FIG. 8 is a cross-sectional view of a part of the integrated circuit according to the first embodiment.

FIG. 9 is a diagram illustrating an entire configuration according to a second embodiment of the present invention.

FIG. 10 is a circuit diagram of an operation controller 310 and a voltage converter 52 according to the second embodiment.

FIG. 11 is a circuit diagram of a modulation unit 300 according to the second embodiment.

FIG. 12 is a circuit diagram of an amplitude circuit 392 according to the second embodiment.

FIG. 13 is a circuit diagram of another configuration of the amplitude circuit 392 according to the second embodiment.

FIG. 14 is a circuit diagram of another configuration of the amplitude circuit 392 according to the second embodiment.

FIG. 15 is a diagram illustrating an entire configuration according to a third embodiment of the present invention.

FIG. 16 is a circuit diagram of a high voltage output device 450 according to the third embodiment.

FIG. 17 is a circuit diagram of a power path switch unit 54 and a voltage converter 52 according to the third embodiment.

FIG. 18 is a diagram illustrating an overall configuration according to a fourth embodiment of the present invention.

FIG. 19 is a diagram illustrating a second current amplifier 645 and a second current amplifier 645 according to the fourth embodiment.
It is a circuit diagram of 646,647.

FIG. 20 is a diagram illustrating an overall configuration according to a fifth embodiment of the present invention.

FIG. 21 is a diagram illustrating an overall configuration according to a sixth embodiment of the present invention.

FIG. 22 is a diagram illustrating another configuration of the power amplifier according to the embodiment of the present invention.

FIG. 23 is a diagram illustrating another configuration of the power amplifier according to the embodiment of the present invention.

FIG. 24 is a diagram showing another configuration of the power amplifier according to the embodiment of the present invention.

FIG. 25 is a diagram illustrating another configuration of the power amplifier according to the embodiment of the present invention.

FIG. 26 is a diagram showing another configuration of the current path forming circuit of the voltage converter according to the embodiment of the present invention.

FIG. 27 is a diagram showing another configuration of the current path forming circuit of the voltage converter according to the embodiment of the present invention.

FIG. 28 is a diagram showing another configuration of the current supply device in the embodiment of the present invention.

FIG. 29 is a waveform chart for explaining the operation of the example of the present invention.

FIG. 30 is a diagram showing another configuration of the power amplifier according to the embodiment of the present invention.

FIG. 31 is a diagram illustrating another configuration of the power amplifier according to the embodiment of the present invention.

FIG. 32 is a diagram showing a configuration of a conventional motor.

[Explanation of symbols]

DESCRIPTION OF SYMBOLS 1 Moving body 2,3,4 Coil 11,12,13 1st power amplifier 15,16,17,615,616,617 2nd power amplifier 30 Current supply device 34 Switching creation device 36 Distribution creation device 37 First Distributor 38 second distributor 41, 42, 43 first current amplifier 45, 46, 47, 645, 646, 647 second current amplifier 50 DC power supply 51, 310 operation controller 52, 752 voltage converter 53,450 High voltage output device 54 Power switch device 490,740,790 Voltage extractor 700 Shunt switch device 701 De-energization device

Claims (44)

[Claims] A voltage supply for outputting a converted DC voltage obtained by converting a DC voltage of the DC power supply, comprising: a moving body; a multi-phase coil; And Q means (Q is an integer of 2 or more) each including a first field-effect power transistor forming a current path from the negative output terminal side of the voltage supply means to one of the coils of the plurality of phases. A first power amplifying means, and Q second power effect transistors each including a second field effect type power transistor forming a current path from a positive output terminal side of the voltage supply means to one of the coils of the plurality of phases. Power amplifying means, switching creating means for obtaining a switching signal of a plurality of phases, and the first Q switching means in response to an output signal of the switching creating means.
Distribution control means for controlling the distribution of current from the power amplifying means, and causing at least one of the Q first field-effect power transistors to perform a resistive voltage drop operation; In response to the output signal of
Distribution control means for distributing and controlling energization from the power amplifying means, and causing at least one of the Q second field-effect power transistors to perform a resistive voltage drop operation; and the voltage supply means. An operation control means for changing the converted DC voltage between the positive output terminal side and the negative output terminal side in synchronization with the moving operation of the moving body. 2. The operation control means includes a modulation means for obtaining a modulated signal which changes in synchronization with a movement operation of the moving body, and the positive output of the voltage supply means in response to an output signal of the modulation means. The conversion DC voltage between a terminal side and the negative output terminal side is changed, and the modulation unit is configured to change the modulated signal in response to the conversion DC voltage of the voltage supply unit. Item 2. The motor according to Item 1. 3. The operation control means includes a modulation means for obtaining a modulated signal which changes in synchronization with the movement of the moving body, and the positive output of the voltage supply means in response to an output signal of the modulation means. Changing the converted DC voltage between the terminal side and the negative electrode output terminal side, wherein the modulating means changes the modulated signal in response to a command signal for commanding supply of power to the coil. The motor according to claim 1, wherein: 4. The first distribution control means controls the energization of the Q first power amplifying means with a first Q-phase current signal which smoothly changes at least in a rising slope part and a falling slope part. The motor according to any one of claims 1 to 3, wherein the motor includes means for supplying the motor to a terminal. 5. The first distribution control means includes means for changing the first Q-phase current signal in response to a command signal for commanding supply of power to the plurality of coils. The motor according to claim 4. 6. The second distribution control means controls the energization of the Q second power amplifying means with a second Q-phase current signal that smoothly changes at least in a rising slope part and a falling slope part. The motor according to any one of claims 1 to 5, comprising means for supplying the motor to a terminal. 7. The second distribution control means includes means for changing the second Q-phase current signal in response to a command signal for commanding supply of power to the plurality of coils. The motor according to claim 6. 8. The voltage supply means includes a conversion inductor means for storing magnetic energy, a conversion capacitor means for storing electric energy, a current outflow terminal connected to a negative terminal side of the DC power supply, and a current inflow terminal. A switching means for switching the power supply path for replenishing magnetic energy of the conversion inductor means from the DC power supply at a high frequency, the switching means including the field-effect switching transistor connected to one end of the conversion inductor means; ON of the effect type switching transistor
Current path forming means for forming a current path from the conversion inductor means to a circuit side including the conversion capacitor means by performing an off-on operation complementarily to an off high-frequency switching operation; And a converted DC voltage is output between one end of the DC power supply and one end of the DC power supply, and the converted DC voltage is supplied to the Q first power amplifiers and the Q second power amplifiers. The motor according to any one of claims 1 to 7. 9. The voltage supply means supplies a converted DC voltage to the Q first power amplifying means and the Q second power amplifying means, and supplies the converted DC voltage to an output of the DC power supply. The motor according to any one of claims 1 to 8, wherein the motor is configured to be higher than a DC voltage. 10. The first power amplifying means includes a first power diode reversely connected as a parasitic element between current input / output terminals of the first field-effect power transistor, and the second power amplifying means. The power amplifying means has a second power diode reversely connected as a parasitic element between current input / output terminals of the second field-effect power transistor, and further comprising, when the DC power supply is turned off, Power path switching means for interrupting a power path between a positive terminal side of the DC power supply and a current inflow terminal side of the Q second power amplifying means; The motor according to any one of claims 1 to 9, further comprising voltage extracting means for extracting a rectified DC voltage obtained by rectifying the back electromotive force of the coil. 11. A shunt switch means for turning off or on a current path between a positive output terminal side of the voltage supply means and a common connection terminal side of the plurality of phase coils, and a second power amplifier. The motor according to any one of claims 1 to 10, further comprising an energization stopping unit configured to execute or stop supplying current to the coils of the plurality of phases by a unit. 12. The first power amplifying means comprises a first power amplifying means.
A first field effect type power section current mirror circuit having the field effect type power transistor, and the second power amplifying means includes a second field effect type power section having the second field effect type power transistor. The motor according to any one of claims 1 to 11, further comprising a current mirror circuit. 13. A moving body, a single-phase or multi-phase coil, a voltage supply means for supplying a DC voltage, and one of the single-phase or multi-phase coils from a negative output terminal side of the voltage supply means. Q (Q is an integer of 2 or more) first power amplifying means each including a first field-effect power transistor forming a current path of Q second power amplifying means each including a second field-effect power transistor forming a current path to one of the coils of a plurality of phases; switching generating means for obtaining a switching signal; In response to the output signal, the Q first
Distribution control means for controlling the distribution of current from the power amplifying means, and causing at least one of the Q first field-effect power transistors to perform a resistive voltage drop operation; In response to the output signal of
And a second distribution control means for controlling at least one of the Q second field-effect power transistors to perform a resistive voltage drop operation. The voltage supply means is a conversion inductor means for storing magnetic energy, a conversion capacitor means for storing electric energy, a current outflow terminal side is connected to the negative terminal side of the DC power supply, and the current inflow side is A switching unit that includes a field-effect switching transistor connected to one end of the conversion inductor unit, and that performs high-frequency switching of a power supply path that replenishes magnetic energy of the conversion inductor unit from the DC power supply; Off-on operation complementary to the on-off high-frequency switching operation of Current path forming means for forming a current path from the converting inductor means to the circuit side including the converting capacitor means, wherein a converted DC voltage is applied between one end of the converting capacitor means and one end of the DC power supply. And outputs the converted DC voltage to the Q first
And a motor for supplying power to the Q second power amplifying means. The motor further comprises: a field-effect switching transistor, the first field-effect power transistor, and the second field-effect power transistor. A motor having integrated circuit means for forming a type power transistor together with required semiconductor elements in a single chip integrated circuit. 14. The motor according to claim 13, further comprising an operation control unit that changes the converted DC voltage of the voltage supply unit in synchronization with a movement operation of the moving body. 15. The first distribution control means and the second distribution control means.
Means for supplying a current signal to one of the energization control terminals of the Q first power amplifying means and the Q second power amplifying means, wherein the current signal has at least a rising edge. The motor according to any one of claims 13 and 14, wherein the motor changes smoothly in an inclined portion and a falling inclined portion. 16. The first power amplifying means has a first power diode reversely connected as a parasitic element between current input / output terminals of the first field-effect power transistor, and the second power amplifying means. The power amplifying means has a second power diode reversely connected as a parasitic element between current input / output terminals of the second field-effect power transistor, and further comprising, when the DC power supply is turned off, Power path switching means for interrupting a power path between a positive terminal of the DC power supply and a current inflow terminal of the Q second power amplifying means; and when the DC power supply is turned off, The motor according to any one of claims 13 to 15, further comprising a voltage extracting means for extracting a rectified DC voltage obtained by rectifying the back electromotive force of the coils of the plurality of phases. 17. The apparatus according to claim 17, wherein the voltage supply means is configured to make a converted DC voltage output between one end of the conversion capacitor means and one end of the DC power supply larger than an output DC voltage of the DC power supply. The motor according to any one of claims 13 to 16. 18. The first power amplifying means according to claim 1, wherein:
A first field effect type power section current mirror circuit using the field effect type power transistor, and the second power amplifying means comprises a second field effect type power transistor using the second field effect type power transistor. The motor according to any one of claims 13 to 17, further comprising a power unit current mirror circuit. 19. A moving body, a multi-phase coil, voltage supply means for supplying a DC voltage, and a current path from a negative output terminal side of the voltage supply means to one of the multi-phase coils. Q (Q is an integer of 2 or more) first power amplifying units each including a first field-effect type power transistor; Q second power amplifying means each including a second field-effect power transistor forming a current path, switching generating means for obtaining a switching signal of a plurality of phases, and in response to an output signal of the switching generating means The Q first
A first distribution control means for performing distribution control of current supply to the stage from the power amplifying unit, and causing at least one of the Q first field-effect power transistors to perform a resistive voltage drop operation; In response to the output signal of the creation means, the Q second
Distribution control means for distributing and controlling energization from the power amplifying means, and causing at least one of the Q second field-effect power transistors to perform a resistive voltage drop operation; and the voltage supply means. Shunt switch means having a shunt transistor for turning off or on a current path between the positive output terminal side of the plurality of coils and the common connection terminal side of the coils of the plurality of phases; and Current-carrying means having a current-carrying transistor for performing or stopping current supply to the coils of the plurality of phases; and a current outflow terminal side to a current inflow terminal side of at least one field-effect power transistor of the second field-effect power transistor. A diode means that can be energized in one direction toward the motor. 20. A parasitic diode element which is formed by a reverse-connected field-effect conduction transistor, and which can conduct current in one direction from a current inflow terminal to a current outflow terminal of the field-effect conduction transistor. 20. The motor according to claim 19, wherein the motor is formed. 21. An operation control means for changing a converted DC voltage between the positive output terminal side and the negative output terminal side of the voltage supply means in synchronization with a movement operation of the moving body. A motor according to claim 19 or claim 20. 22. The first distribution control means controls energization of the Q first power amplifying means with a first Q-phase current signal which smoothly changes at least in a rising slope part and a falling slope part. The second distribution control means is configured to supply a second Q-phase current signal that smoothly changes at least in a rising slope portion and a falling slope portion to the Q number. 22. The motor according to any one of claims 19 to 21, comprising means for supplying the power to the power supply control terminal side of the second power amplifying means. 23. The voltage supply means includes a field-effect switching transistor that performs high-frequency switching on a power supply path of a DC power supply, and converts the DC voltage obtained by converting the DC voltage of the DC power supply into the Qth power supply. 1 power amplification means and said Q second power amplification means,
23. The motor according to claim 19, wherein the converted DC voltage is configured to be higher than an output DC voltage of the DC power supply. 24. A moving body, a multi-phase coil, voltage supply means for supplying a DC voltage, and a current path from a negative output terminal side of the voltage supply means to one of the multi-phase coils. Q (Q is an integer of 2 or more) first power amplifying units each including a first NMOS type power transistor; Q second power amplifying means each including a second PMOS type power transistor forming a current path; switching creating means for obtaining a switching signal of a plurality of phases; and Q in response to an output signal of the switching creating means. First
Distribution control means for performing distribution control of energization from the power amplifying means, and causing at least one of the Q first power transistors to perform a resistive voltage drop operation; and an output signal of the switching generation means. In response to the Q second
Distribution control means for distributing and controlling energization from the power amplifying means, and causing at least one of the Q second power transistors to perform a resistive voltage drop operation; and a DC power supply for the voltage supply means. Is turned on, a power path is connected from the positive terminal side of the DC power supply to the current inflow ends of the Q second power amplification means, and the DC power supply of the voltage supply means is turned off. Power path switching means having a PMOS type power path switching transistor for interrupting a power path between the positive terminal side of the DC power supply and the current inflow terminal sides of the Q second power amplification means. And a voltage extracting means for extracting a rectified DC voltage obtained by rectifying the back electromotive force of the coil when the DC power supply is turned off. 25. An operation control means for changing a converted DC voltage between the positive output terminal side and the negative output terminal side of the voltage supply means in synchronization with a movement operation of the moving body. 25. The motor according to 24. 26. The first distribution control means controls the energization of the Q first power amplification means with a first Q-phase current signal that smoothly changes at least in a rising slope part and a falling slope part. The second distribution control means is configured to supply a second Q-phase current signal that smoothly changes at least in a rising slope portion and a falling slope portion to the Q number. 26. The motor according to claim 24, further comprising means for supplying a power to a power supply control terminal of the second power amplifying means. 27. The voltage supply means includes a field-effect switching transistor that performs high-frequency switching on a power supply path of a DC power supply, and converts the converted DC voltage obtained by converting the DC voltage of the DC power supply into the Qth power supply. 1 power amplification means and said Q second power amplification means,
The motor according to any one of claims 24 to 26, wherein the conversion DC voltage is configured to be higher than the output DC voltage of the DC power supply. 28. The power path switch means includes the PMOS type power path switch transistor connected in reverse, and the PMOS type power path switch transistor is connected in one direction from a current input terminal side to a current output terminal side of the PMOS type power path switch transistor. The motor according to any one of claims 24 to 27, wherein an energizable parasitic diode element is formed. 29. A moving body, a multi-phase coil, voltage supply means for supplying a DC voltage, and a current path from a negative output terminal side of the voltage supply means to one of the multi-phase coils. Q (Q is an integer of 2 or more) each including a first field-effect power transistor and having a first field-effect power section current mirror circuit using the first field-effect power transistor
A first power amplifying means, and a second field effect type power transistor forming a current path from a positive output terminal side of the voltage supply means to one of the coils of the plurality of phases. Q second power amplifying means having a second field effect type power section current mirror circuit using an effect type power transistor; switching generating means for obtaining a switching signal of a plurality of phases; output signal of the switching generating means When a current path responding to the above is formed, the first Q-phase current signal that smoothly changes at least in the rising slope portion and the falling slope portion is supplied to the energization control terminals of the Q first power amplifying means, respectively. A first distribution control unit that causes at least one of the Q first field-effect power transistors to perform a resistive voltage drop operation; When a current path corresponding to the output signal is formed, the current signal of the second Q phase, which smoothly changes at least in the rising slope portion and the falling slope portion, is supplied to the energization control terminals of the Q second power amplifying means. And a second distribution control means for supplying at least one of the Q second field-effect power transistors to a resistive voltage drop operation. 30. The voltage supply means includes a field effect switching transistor that performs high-frequency switching on a power supply path of a DC power supply, and converts the DC voltage obtained by converting the DC voltage of the DC power supply into the Qth power supply. 1 power amplification means and said Q second power amplification means,
30. The motor according to claim 29, wherein the conversion DC voltage is configured to be higher than an output DC voltage of the DC power supply. 31. An operation control means for changing a converted DC voltage between the positive output terminal side and the negative output terminal side of the voltage supply means in synchronization with a movement operation of the moving body. A motor according to claim 29 or claim 30. 32. The first power amplifying means has a first power diode reversely connected as a parasitic element between current input / output terminals of the first field-effect power transistor, and the second power amplifying means. The power amplifying means has a second power diode reversely connected as a parasitic element between current input / output terminals of the second field-effect power transistor, and further comprising, when the DC power supply is turned off, Power path switching means for interrupting a power path between a positive terminal side of the DC power supply and a current inflow terminal side of the Q second power amplifying means; The motor according to any one of claims 29 to 31, further comprising a voltage extracting means for extracting a rectified DC voltage obtained by rectifying the back electromotive force of the coil. 33. A moving body, a single-phase or multiple-phase coil, a voltage supply means for outputting a DC voltage, and one of the single-phase or multiple-phase coils from one output terminal side of the voltage supply means. (Q is 2) including first field-effect power transistors each forming a current path to
A first power amplifying means, and a second field-effect power transistor forming a current path from the other output terminal of the voltage supply means to one of the single-phase or multi-phase coils. Q second power amplifying means each including: a switching generating means for obtaining a switching signal; and the Q first power amplifying means in response to an output signal of the switching generating means.
First distribution control means for distributing and controlling the energization from the power amplifying means, and said Q second control means in response to an output signal of said switching creation means.
And a second distribution control means for performing distribution control of power supply from the power amplifying means, wherein the first distribution control means forms a current path in response to an output signal of the switching creating means. Supplying a first Q-phase current signal that smoothly changes at least in the rising slope portion and the falling slope portion to the energization control terminals of the Q first power amplifying means, respectively, A motor comprising means for causing at least one of the field effect type power transistors to perform a resistive voltage drop operation. 34. The first distribution control means includes means for changing the first Q-phase current signal in response to a command signal for commanding power supply to the single-phase or multi-phase coils. The motor according to claim 33, wherein the motor is configured as follows. 35. The second distribution control means according to claim 27, wherein the second Q-phase current signals that smoothly change at least in the rising slope portion and the falling slope portion are converted into the Q second current signals.
To the energization control terminal side of the power amplification means of
35. The motor according to claim 33, further comprising means for causing at least one of the Q second field-effect power transistors to perform a resistive voltage drop operation. 36. The second distribution control means includes means for changing the second Q-phase current signal in response to a command signal for commanding power supply to the single-phase or multi-phase coils. 36. The motor according to claim 35, configured as follows. 37. The voltage supply means includes a field-effect switching transistor that performs high-frequency switching on a power supply path of a DC power supply, and converts the converted DC voltage obtained by converting the DC voltage of the DC power supply into the Qth power supply. 1 power amplification means and said Q second power amplification means,
The motor according to any one of claims 33 to 36, wherein the converted DC voltage is configured to be higher than the output DC voltage of the DC power supply. 38. An operation control means for changing a converted DC voltage between the positive output terminal side and the negative output terminal side of the voltage supply means in synchronization with a movement operation of the moving body. The motor according to any one of claims 33 to 37. 39. The first power amplifying means has a first power diode reversely connected as a parasitic element between current input / output terminals of the first field-effect power transistor, and the second power amplifying means. The power amplifying means has a second power diode reversely connected as a parasitic element between current input / output terminals of the second field-effect power transistor, and further comprising, when the DC power supply is turned off, Power path switching means for interrupting a power path between a positive terminal of the DC power supply and a current inflow terminal of the Q second power amplifying means; and when the DC power supply is turned off, 39. The motor according to any one of claims 33 to 38, further comprising: voltage extracting means for extracting a rectified DC voltage obtained by rectifying a back electromotive force of a coil of a plurality of phases. 40. A shunt switch for turning off or on a current path on a positive output terminal side of the voltage supply means and a common connection terminal side of the coils of the plurality of phases, and the Q second power amplifiers. 40. The motor according to any one of claims 33 to 39, further comprising: a power supply stopping unit configured to perform or stop supplying current to the plurality of coils by a unit. 41. A moving body, a single-phase or multi-phase coil, a voltage supply means for supplying a DC voltage, and one of the single-phase or multi-phase coils from one output terminal side of the voltage supply means. (Q is 2) including first field-effect power transistors each forming a current path to
A first power amplifying means, and a second field-effect power transistor forming a current path from the other output terminal of the voltage supply means to one of the single-phase or multi-phase coils. Q second power amplifying means respectively comprising: a switching signal generating means for generating a switching signal; and the Q first power amplifying means in response to an output signal of the switching signal generating means.
First distribution control means for distributing and controlling the energization from the power amplifying means, and said Q second control means in response to an output signal of said switching creation means.
And a second distribution control means for distributing and controlling the energization from the power amplifying means, wherein the Q first power amplifying means and the Q second power amplifying means are provided. Wherein at least one power amplifying means has a field-effect power transistor, a field-effect power section current mirror circuit including a field-effect transistor and a resistor, and the field-effect power section current mirror circuit. Connects the control terminal side of the field-effect power transistor to the control terminal side of the field-effect transistor, and connects one end of a current path of the field-effect transistor to the field-effect power transistor via the resistor. The other end of the current path of the field effect transistor is connected to one end of the current path, and the other end of the current path is connected to the conduction control terminal side of the power amplifier. The control terminal side effect transistor is configured to connect to the conduction control terminal side of the power amplifier, motor. 42. The first distribution control means and the second distribution control means
Wherein the distribution control means includes means for supplying a current signal that smoothly changes in at least a rising slope part and a falling slope part to a power supply control terminal side of the at least one power amplifying means. 42. The motor according to 41. 43. A moving body, a single-phase or multiple-phase coil, a voltage supply means for supplying a DC voltage, and one of the single-phase or multiple-phase coils from one output terminal side of the voltage supply means. (Q is 2) including first field-effect power transistors each forming a current path to
A first power amplifying means, and a second field-effect power transistor forming a current path from the other output terminal of the voltage supply means to one of the single-phase or multi-phase coils. Q second power amplifying means respectively comprising: a switching signal generating means for generating a switching signal; and the Q first power amplifying means in response to an output signal of the switching signal generating means.
First distribution control means for distributing and controlling the energization from the power amplifying means, and said Q second control means in response to an output signal of said switching creation means.
And a second distribution control means for distributing and controlling the energization from the power amplifying means, wherein the Q first power amplifying means and the Q second power amplifying means are provided. Wherein at least one power amplifying means has a field-effect power transistor, a field-effect power section current mirror circuit including a field-effect transistor and a resistor, and the field-effect power section current mirror circuit. Connects the control terminal side of the field-effect power transistor to the control terminal side of the field-effect transistor, and connects one end of a current path of the field-effect transistor to the field-effect power transistor via the resistor. The other end of the current path of the field effect transistor is connected to one end of the current path, and the other end of the current path is connected to the conduction control terminal side of the power amplifier. The control terminal side effect transistor is configured to connect to the conduction control terminal side of the power amplifier, motor. 44. The first distribution control means and the second distribution control means
44. The distribution control means includes a means for supplying a current signal that smoothly changes at least in a rising slope part and a falling slope part to an energization control terminal side of the at least one power amplifying means. A motor according to claim 1.