JPH1093376A - Surface acoustic wave filter - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a surface acoustic wave filter of a ladder type in which an extension of a passing bandwidth and an improvement of suppression degree outside the passing band are simultaneously attained. SOLUTION: This surface acoustic wave filter is formed by connecting a 1st one-terminal pair surface acoustic wave resonator providing a prescribed resonance frequency to a parallel arm and a 2nd one-terminal pair surface acoustic wave resonator providing a resonance frequency almost coincident with an anti-resonance frequency of a 1st resonator to a series arm. In this case, 2nd one-terminal pair surface acoustic wave resonators R2 , R2 are connected in series with the series arm 61 and an inductance (LS) is connected in series to the connection.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a surface acoustic wave filter, and more particularly to a ladder type surface acoustic wave filter applicable to an RF (high frequency section) filter of a small mobile radio device such as a mobile phone and a mobile phone. . One example of the current specification of automobiles and mobile phones in Japan has a transmission band in the range of ± 8.5 MHz centered at 933.5 MHz. In terms of fractional bandwidth, it is about 2%.

A surface acoustic wave filter needs to have characteristics satisfying the above-mentioned specifications. Specifically, a pass band width is as wide as 2% or more as a fractional bandwidth, and a loss is 1.5. ~ 2dB or less, 20dB suppression
It is required to be as high as ３０30 dB or more. In order to satisfy this requirement, a surface acoustic wave filter using a surface acoustic wave element as a resonator instead of the conventional transversal type and having a ladder type is desired.

[0003]

2. Description of the Related Art FIG. 70 shows an equivalent circuit of a surface acoustic wave filter 1 described in JP-A-52-19044. This filter 1 includes a surface acoustic wave resonator 3
And the surface acoustic wave resonator 5 is disposed on the parallel arm 4, and the equivalent parallel capacitance C _OB of the resonator 5 of the parallel arm 4 is larger than the equivalent parallel capacitance C _OA of the resonator 3 of the series arm 2. Configuration.

[0004] The filter 1 has a characteristic shown by a line 6 in FIG.

[0005]

In the above filter 1, if the equivalent parallel capacitance C _OB is increased as described later,
As shown by arrow 7, the degree of suppression can be increased. However, when the capacitance C _OB is increased, the pass bandwidth becomes narrow as shown by the arrow 8 and the loss increases as shown by the arrow 9, and the characteristic becomes as shown by the line 10.

If the suppression degree is to be set to 20 dB or more,
The pass bandwidth is less than 1% in fractional bandwidth,
The above-mentioned specification of the mobile phone cannot be satisfied. Therefore, an object of the present invention is to provide a surface acoustic wave filter capable of simultaneously achieving an increase in the pass band width and an improvement in the degree of suppression outside the pass band.

[0007]

Means for Solving the Problems The invention according to claim 1
Is a first one terminal pair elastic surface having a predetermined resonance frequency
Wave resonator in parallel arm and anti-resonance frequency of the first resonator
Second one-port surface acoustic wave having substantially coincident resonance frequencies
Ladder type surface acoustic wave filter with resonators arranged in series
In the above, the series arm (61) is provided with a second one-terminal pair elastic member.
Surface wave resonator (R _Two, R_Two) Are connected in series and arranged
And an inductance (L_S)
The surface acoustic wave filter is characterized in that
You.

According to a second aspect of the present invention, a first terminal-pair surface acoustic wave resonator having a predetermined resonance frequency is provided in a parallel arm with a resonance frequency substantially matching the anti-resonance frequency of the first resonator. In a ladder-type surface acoustic wave filter having a second one-port surface acoustic wave resonator having a series arrangement in a serial arm, a second one-port surface acoustic wave resonator ( R ₂ , R ₂ ) are connected and arranged in series and an inductance (L ₁ ) is added thereto in series, and the first surface acoustic wave resonator is connected to the central excitation electrode (131) And reflectors (132, 133) on both sides, and the distance between the reflectors and the center of the excitation electrode is d = (n + β).
When λ (where n is an integer, β is a real number equal to or less than 1 and λ is the period of the comb-shaped electrode corresponding to the resonance frequency), the arrangement is such that β is substantially 0.4. This is a characteristic surface acoustic wave filter.

[0009]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS First, the basic principle of the present invention will be described. FIG. 1 shows the principle configuration of a surface acoustic wave filter 20 according to the present invention. _{Reference numeral} 21 denotes a first one-terminal surface acoustic wave resonator having a predetermined resonance frequency f _rp and arranged on the parallel arm 22. 23 is a second one-terminal surface acoustic wave resonator;
It has a resonance frequency f _as which is substantially equal to the anti-resonance frequency f _rp of the first resonator 21, and is arranged on the series arm 24. Reference numeral 25 denotes an inductance, which is added in series to the first resonator 21 and disposed on the parallel arm 22.

The principle that a circuit having a one-port surface acoustic wave resonator in a series arm and a parallel arm has filter characteristics is as follows. This principle is necessary for explanation of the principle of the present patent, and will be described in detail here. In order to evaluate whether or not the resonance circuit exhibits the filter characteristics, a method using image parameters is easy to understand. This method is described in detail in "Theory and Design of Filters" by Yanagisawa et al. (Sanbo Publishing, Electronics Selection, published in 1974).

The principle will be described below based on this. FIG. 2 shows a basic ladder circuit showing the filter characteristics. In the figure, the hatched black box indicates the surface acoustic wave resonator 3
0,31. Now, for simplicity of description, it is assumed that the surface acoustic wave resonator is a reactance circuit having no resistance, and the impedance of the resonator 30 in the series arm is Z = jx,
The admittance of the resonator 31 of the parallel arm is Y = jb.

According to the image parameter method, if the input side voltage and current are V ₁ and I ₁ , and the output side is V ₂ and I ₂ (see FIG. 2),

[0013]

(Equation 1)

Image transmission amount γ (complex number) defined by
Has important implications. That is,

[0015]

(Equation 2)

In the equation, if the value represented by this equation is an imaginary number, the entire two-port circuit of FIG. 2 shows a pass characteristic, and if the value is a real number, it shows an attenuation characteristic. Here, the symbols A, B, C, and D are four-terminal constants when the entire circuit of FIG. 2 is represented by an F matrix, and when expressed by the above-described x and b, they are as follows. A = 1 B = xx C = jb D = 1−bx (3) Therefore, the expression (2) becomes the following expression.

[0017]

(Equation 3)

From equation (4), 0 <bx <1, that is, b and x
2 has the same sign and a small value, the entire circuit of FIG. 2 shows a pass characteristic. When bx <0 or bx> 1, that is, when b and x have different signs or a large value of the bx product, the circuit shows an attenuation characteristic. You can see that. Here, in order to further qualitatively know the frequency characteristics of b and x, the frequency characteristics of the impedance and admittance of the surface acoustic wave resonator are examined.

The one-port pair surface acoustic wave resonator comprises a comb-shaped electrode 40 as shown in FIG. 3A (see P.76-Nikkei Electronics November 29, 1976).
P. 98). 41 is an electrode pair, 42 is an opening length (intersection width), and 43 is a comb-shaped electrode period. This comb-shaped electrode is generally represented by an equivalent circuit 45 as shown in FIG. Here, C _O is the capacitance of the comb-shaped electrode, and C ₁ and L ₁ are equivalent constants.

This equivalent circuit 45 is hereinafter represented by a symbol 46 shown in FIG. FIGS. 4A and 4B qualitatively show the frequency dependence of impedance and admittance when each of the comb electrodes is represented by an equivalent circuit as shown in FIG. 3B. The characteristic shown in the figure is a double resonance characteristic having two resonance frequencies fr and fa, similarly to the resonator made of quartz. Where f
r is called a resonance frequency, and fa is called an anti-resonance frequency. When the resonators having such double resonance characteristics are arranged in the series arm and the parallel arm, respectively, and the anti-resonance frequency fap of the parallel arm is made substantially equal to the resonance frequency frs of the series arm, the band having the center frequency as the center frequency is obtained. A circuit exhibiting pass-type filter characteristics can be configured. The reason is that as shown in the diagram of the frequency characteristic of the immittance in FIG. 5A, near the center frequency where fap ≒ frs, 0 <bx <1 is satisfied and the passband is obtained from the above condition, This is because bx> 1 in a frequency region slightly distant from the frequency, and bx <0 in a region far away from the frequency, which is an attenuation region.

Therefore, the surface acoustic wave filter 1 having the structure shown in FIG. 1 qualitatively has a filter characteristic indicated by a middle line 47 in FIG. 5B. [Factors for Determining Pass Bandwidth] Next, factors for determining the bandwidth in such a resonator type surface acoustic wave filter will be considered.

As can be seen from FIG. 5, the bandwidth is mainly determined by the difference between the resonance frequency fr and the anti-resonance frequency fa of each resonator. If the difference is large, the bandwidth is wide and wide, and if the difference is small, the bandwidth is narrow. Here, fr and fa can be determined from the following equations using the equivalent circuit constants in FIG.

[0023]

(Equation 4)

The fractional bandwidth (Δf / f ₀ ) is mainly fr, f
Since it is determined from the difference of a, it is expressed by the following equation using the equations (6) and (7). Δf / f ₀ = 2 (fa−fr) / (fa + fr) ≒ 2 / (4γ + 1) (8) As is apparent from the above equation, γ (capacity ratio) is an important factor that determines the fractional bandwidth. However, this value is disclosed in JP-A-52-1.
As described in Japanese Patent Application Laid-Open No. 9044, it is almost determined by the type of the substrate material forming the comb-shaped electrode. For example, in an ST-cut quartz having a small electromechanical coupling coefficient of a material, γ is 1300 or more, whereas in a 36 ° Ycut-x propagating LiTaO ₃ having a large
γ takes the value of the 15th place. The fractional bandwidth is expressed by ST
0.04% in cut quartz, 36 ° Ycut-X propagation L
In iTaO ₃ , it is 3.3%. Therefore, if the substrate material is determined, the bandwidth is almost determined.

If the equivalent parallel capacitance C _OB is increased as disclosed in Japanese Patent Application Laid-Open No. 52-19044 in order to increase the degree of out-of-band suppression, the bandwidth becomes narrower. This will be described in detail with reference to FIG. As is clear from the above explanation of the principle, when the admittance value is increased while the fr and fa of the parallel resonator are fixed (in order to increase the admittance value, the opening length of the comb-shaped electrode or The capacitance C ₀ is increased by increasing the logarithm, and as shown in FIG. 6 (A), the bx product is negative and increases outside the band, so that the attenuation increases and the characteristics are improved. Bx> 1 because the bx product is positive and increases
As a result, the passband where 0 <bx <1 is narrowed, and a sufficient band cannot be obtained. Figure 6 shows this situation.
It is represented by the arrow in (B).

[Improvement of Pass Bandwidth] As one means for solving the above points, the difference between fr and fa of at least one of the series arm resonator and the parallel arm resonator is increased. It is necessary to satisfy two conditions of increasing the impedance value or the admittance value. The reason for increasing the impedance value and the admittance value is to increase the out-of-band attenuation. If this can be realized, it is possible to improve the out-of-band attenuation without widening or narrowing the pass band.

First, the conditions fr, fa of the resonator are satisfied.
As a method for widening the difference, a method of adding an inductance L in series with a one-port surface acoustic wave resonator is effective. FIGS. 7A and 7B show changes in impedance and admittance frequency when 8 nH is connected as L in series with a one-port surface acoustic wave resonator. Each constant of the equivalent circuit of the surface acoustic wave resonator used for the calculation is shown in FIG.

In FIG. 7A, a line 50 indicates an impedance characteristic before L is added. Line 51 shows the impedance characteristic after L is added. In FIG. 7B, a line 52 indicates admittance characteristics before L is added. Line 53 is
The admittance characteristic after adding L is shown. FIG. 7 (A)
It can be seen that the addition of L widens the interval between fr and fa. In this case, the frequency was increased by about 30 MHz. The reason for this is that, as is apparent from the frequency characteristic of the impedance in FIG. 3A, the impedance of only the original resonator is raised to the + side by ωL by adding L in series, so that fr becomes fr ′. Because it has changed to At this time, fa hardly moves. The admittance, which is the reciprocal of impedance, is the same for the same reason (A).
Changes as shown in FIG. Also in this case, it can be clearly seen that fr has changed to fr '.

Under the following conditions, the admittance value is increased by adding L, as is clear from FIG. 7B. However, the impedance values are
As shown in (A), it is smaller outside the band.
Therefore, when this method is applied to a resonance circuit of a series arm, a method of increasing the impedance value is further required. This can be solved by connecting a plurality of the same surface acoustic wave resonators in series.

In FIG. 8, a line 55 indicates the impedance characteristic of one resonator. A line 56 indicates the impedance characteristic of the resonance part when n resonators are connected in series.
As shown in FIG. 8, by connecting n resonators, the impedance value of the resonator section becomes n times. On the other hand, regarding the difference between fa and fr, the expansion of the resonance frequency when L is connected is fr ", which is slightly narrower than fr 'in the case of one resonator, but is fa and fr more than when L is not connected. The difference between fa and fr can be further increased by increasing the value of L if necessary.

As a second means for increasing the pass band width, as shown in FIG. 44, the anti-resonance frequency fa of the parallel arm resonator
Instead of making p and the series arm resonance frequency frs approximately match,
A method of setting frs> fap is conceivable. Where frs> fap
In this case, as shown in FIG.
x <0, and the passband condition is not satisfied,
There is a risk of increased losses and ripple.

However, by controlling the magnitude of Δf by setting frs−fap = Δf, it is possible to substantially increase the loss and increase the ripple while realizing the expansion of the pass band. Further, by increasing Δf, it is possible to simultaneously improve the out-of-band suppression degree. Details will be described later in an eleventh embodiment.

Hereinafter, the contents of the present invention will be described with reference to specific examples. Most of the examples were performed by simulation. Therefore, first, the simulation used in the present invention will be briefly described, and a comparison with an experiment will be shown in order to prove the validity of the simulation. FIG.
The equivalent circuit shown in (B) can simply simulate the characteristics of a one-port SAW resonator, but changes in the logarithm, aperture length, electrode thickness, etc. of the comb-shaped electrodes constituting the resonator, and the effect of the reflector Is difficult to simulate accurately. Therefore, based on the Smith equivalent circuit that the inventors have already developed, we use a method of expressing this as a transfer matrix,
(O.Ikata et al.:1990 ULTRASONIC S
See YMPOSIUM Proceedings, vol. 1, pp. 83-86, (1990). ).

FIG. 9A shows the result of a simulation when a one-port surface acoustic wave resonator is provided in a parallel arm. FIG. 9 (B) shows that the material of the parallel arms is Al-2% Cu,
The experimental results are shown in the case where a one-port surface acoustic wave resonator composed of comb electrodes having a film thickness of 1600 ° is provided, and a bonding wire (L = 1.5 nH) having a length of 3 mm is connected to the resonator. .

9 (A) and 9 (B), the movement of the resonance point (indicated by fr ₁ , fr ₂ , fr ₃ in the figure) due to the change of the aperture length and the amount of attenuation near the resonance point are as follows. It turns out that the experimental value and the calculated value agree well. FIG. 10 (A)
Shows simulation results when a resonator is provided in the series arm. Since the bonding pad used in the experiment described later was rather large, the simulation considered a 0.5 pF capacitor as its stray capacitance.

FIG. 10B shows the result of an experiment in which a resonator is connected to the series arm. Comparing FIGS. 10A and 10B, the fact that the anti-resonance frequencies fa ₁ , fa ₂ , and fa ₃ do not depend on the aperture length, the change in the attenuation near the anti-resonance frequency, and the like are well known from experiments. You can see that we are doing it. Therefore, it is clear that the filter characteristics when these are combined well match the experiments, and the following examples were simulated.

FIG. 11 shows a surface acoustic wave filter 60 according to a first embodiment of the present invention. At present, one example of the specifications of domestic automobiles and mobile phones is:
± 8.5 MHz with 933.5 MHz as center frequency
Is the transmission band of the mobile device, from which -55 MHz
± 8.5 with centered frequency at 878.5 MHz
There is a specification that a range of MHz is a reception band.

This embodiment is designed so as to be suitable for the transmitting filter of the mobile device. The same applies to other embodiments described later. One terminal pair surface acoustic wave resonators R ₂ and R ₄ are arranged on the series arm 61. One terminal pair surface acoustic wave resonators R ₁ , R ₃ , R ₅ are arranged on the parallel arms 62, 63, 64, respectively.

L ₁ , L ₂ , L ₃ are inductances, which are connected to the resonators R ₁ , R ₃ , R _{5, respectively,} to connect the parallel arms 6.
2, 63, 64. Each of the resonators R _{1 to} R ₅ has a comb-shaped electrode structure shown in FIG. The logarithm is 100,
The opening length is 80 μm. The material is Al-2% Cu, and the film thickness is 3,000 °.

Also, the period of the comb-shaped electrode is appropriately determined,
Each resonator R ₁ , R ₃ , R _{5 in the} parallel arms 62, 63, 64
Has a resonance frequency of 912 MHz and an anti-resonance frequency of 934 MHz.
MHz. Each resonator R ₂ , R ₄ in the series arm 61
Has a resonance frequency of 934 MHz and an anti-resonance frequency of 962 M
Hz.

The inductances L ₁ , L ₂ and L ₃ are all 4
nH. The surface acoustic wave filter 60 having the above configuration has a pass characteristic indicated by a line 65 in FIG. When the inductance L is 2 nH and 6 nH, the pass characteristics of the filter 60 in FIG. 11 are as shown by lines 66 and 67 in FIG. 12, respectively.

Based on FIG. 12, L with respect to the pass band width
Expressing the dependency is as shown by a line 70 in FIG. Here, the frequency width at the level of the amount of attenuation that is -3 dB lower than the minimum insertion loss was defined as the pass bandwidth. Similarly, based on FIG.
The dependence is represented by a line 71 in FIG.

As can be seen from FIG. 12, if L is too large, the suppression area on the 55 MHz lower frequency side from the center frequency cannot be sufficiently obtained. Therefore, L is set to 4 nH as described above. The value of L is appropriately selected according to the specifications of the filter. The pass characteristic of the filter 1 of the conventional configuration shown in FIG.
It becomes as shown by.

In FIG. 12, the pass characteristic (line 65) of the filter 60 of the present embodiment is changed to the pass characteristic (line 68) of the conventional filter.
In comparison with the filter 60 of the present embodiment, the pass band width is wider as shown by an arrow 75 and the degree of suppression outside the pass band is higher as shown by an arrow 76, as compared with the conventional filter.
In addition, the loss is low as indicated by the arrow 77.
14 and 15 show the surface acoustic wave filter 60 of FIG.
A surface acoustic wave filter device 80 that realizes the above is shown.

Numeral 81 is a ceramic package, numeral 82 is a filter chip, and numeral 83 is a lid functioning as a ground. The ceramic package 81 is made of alumina ceramic and has a size of 5.5 × 4 mm ² and a small height of 1.5 mm. The ceramic package 81 has Au electrode terminals 84 _{-1 to} 84 _-6 formed thereon.

The filter chip 82 is made of LiTaO ₃ , has a size of 2 × 1.5 mm ^{2 and} a thickness of 0.5 mm. The surface of this filter chip 82 has a logarithm of 10
0, resonators R _{1 to} R ₅ having a comb-shaped electrode structure with an aperture length of 80 μm, a material of Al-2% Cu, and a film thickness of 3,000 °
However, they are staggered so as not to share the propagation path of the surface acoustic wave with each other.

On the surface of the filter chip 82, two signal line terminals 85 _-1 and 85-1 as bonding terminals are provided.
_85-2 and three grounding terminals _85-3 , _85-4 , _85-5
Is formed. Reference numerals 86 _{-1 to} 86 _-5 denote bonding wires, made of Al or Au, having a diameter of 25 μmφ, and connected by bonding to the terminals 84 _{-1 to} 84 _-5 and the terminals 85 _{-1 to} 85 _-5 , respectively. I have.

Among them, the wires 86 _-1 and 86 _{-2 respectively} constitute a part 61a and 61b of the serial arm 61 in FIG. Wire 86 _-3 Yes connected between the ground electrode terminals 84 _-3 and 85 _-3, the wire 86 _-4 connected between the other ground electrode terminal 84 _-4 85 _-4 Yes, wire 86
_-5 is connected between the other ground electrode terminals _84-5 and _85-5 .

The length of each of the wires 86 _{-3 to} 86 _-5 is as long as 2.0 mm. As described above, a thin and long wire has an inductance component according to the theory of high frequency. According to the theoretical formula of the aerial ribbon inductor (Kuraishi: Science and Engineering Course, “Example Circumferential Microwave Circuit” described in P199 of Tokyo Denki University Press), the above-mentioned wires 86 _-3 , 86 _-4 , 86 _-5 The inductance is about 1 nH.

This is not enough to obtain an inductance of 4 nH, and a ceramic package and L on the filter chip as shown in FIGS. 40 and 41 described later were used. Thus, the inductances L ₁ , L ₂ , and L ₃ in FIG. 11 are configured. Embodiment 2 FIG. 16 shows a surface acoustic wave filter 90 according to a second embodiment of the present invention.

In the figure, parts corresponding to those shown in FIG. 11 are denoted by the same reference numerals. Aperture length A _S of the resonator R ₂ in the series arm 61 is 80 [mu] m. The parallel arm 62 has a resonator R
_1A and the inductance L ₁ is are disposed are connected in series. Resonator R _1A is the aperture length A _P is 120 [mu] m.

[0052] aperture length A _P is longer than the aperture length A _S, which is 1.5 times the aperture length A _S. The logarithms N _P and N _S of the resonators R ₂ and R _1A are both equal to 100. This filter 90 has a pass characteristic indicated by a line 91 in FIG. The filter 60 of FIG.
It can be seen that the degree of suppression outside the pass band is improved as shown by the arrow 92 without changing the pass band width, as compared with the pass characteristics of FIG.

FIG. 18 shows the aperture length dependence of the pass characteristic of the filter having the configuration of FIG. FIG. 17A shows the relationship between the opening length (A _S ) of each series arm resonator when L is added as shown in FIG. 16 and when L is not added as shown in FIG. Aperture length of parallel arm resonator (A
Indicating the ratio A _P / A _S of _P), the band suppression degree of relationship.

When the inductance L of 4 nH is added, the degree of out-of-band suppression is as shown by a line 92, and when the inductance L is not added, it is as shown by a line 93. FIG. 18B shows A _P /
Shows the relationship between the passband width A _S. Pass bandwidth is 4n
If an inductance L of H is added, line 9
5 and when the inductance L is not added, as shown by a line 96.

FIGS. 18A and 18B show the following. Series arm 61 a aperture length A _P of the resonator R _1A in the parallel arm 62
By making the opening length A _S of the resonator R ₂ in the inside longer,
The degree of out-of-band suppression increases. By adding the inductance L ₁ on the parallel arm 62, as compared with the case having no inductance, the resonator R
The effect of increasing the aperture length A _P of _1A becomes large and almost no degradation of bandwidth.

From the above, it can be seen that the filter 90 of the above embodiment has a pass characteristic in which the degree of suppression outside the pass band is increased without reducing the pass band width at all, as compared with the filter 60 of FIG. I understand. Embodiment 3 FIG. 19 shows a surface acoustic wave filter 100 according to a third embodiment of the present invention.

In the figure, parts corresponding to those shown in FIGS. 11 and 16 are denoted by the same reference numerals. The logarithm N _S of the resonator R ₂ of the series arm 61 is 100. A resonator R _1B and an inductance L ₁ are connected to the parallel arm 62 in series. The logarithm N _{P of the} resonator R _1B is 150, which is larger than the logarithm N _S of the resonator R ₂ , which is 1.5 times the logarithm N _P.

The opening lengths A _S , of the resonators R ₂ and R _1A are
A _P are both 80 [mu] m, equal. This filter 10
0 has a pass characteristic indicated by a line 101 in FIG. Comparing this pass characteristic with the pass characteristic of the filter 60 of FIG. 11 shown by the line 65, it can be seen that the degree of suppression outside the pass band is improved as shown by the arrow 102 without reducing the pass band width.

Also, as compared with the pass characteristic of the filter 90 shown in FIG. Therefore, the filter 100 according to the present embodiment increases the degree of suppression outside the pass band without reducing the pass band width as compared with the filter 11 of FIG.
It has a transmission characteristic with less loss deterioration compared to 0.

Fourth Embodiment FIG. 21 shows a surface acoustic wave filter 110 according to a fourth embodiment of the present invention. In this embodiment,
Anti-resonance frequency f _a and the resonance frequency f _r of the resonant circuit of the series arm
The transmission characteristics are improved by enlarging the difference from the above. In the figure, portions corresponding to the components shown in FIG. 11 are denoted by the same reference numerals.

Two identical resonators R ₂ are connected in series between the parallel arms 62 and 63 of the series arm 61, and an inductance L _S of 3 nH is added thereto in series. Similarly, two identical resonators R ₄ are connected in series between the parallel arms 63 and 64 of the series arm 61, and an inductance L _S of 3 nH is added to this in series.

The parallel arm 62 has only one resonator R position. Similarly, the parallel arm 63 has one resonator R
Only ₃ are arranged. Similarly, the parallel arms 64, only one resonator R ₄ is are arranged. This filter 110 has a pass characteristic indicated by a line 111 in FIG. Here, the effect of adding the inductance L _S and the resonators R ₂ and R ₄ will be described.

When the inductance L _S and one of the resonators R ₂ and R ₄ are deleted from the filter 110 of FIG.
This is the same as the conventional filter 1 shown in FIG. The pass characteristic in this state is as shown by a line 68 (see FIG. 12). With the addition of the inductance L _S , the pass bandwidth was increased as indicated by the arrow 112 and the degree of out-of-band suppression was increased as indicated by the arrow 113. In particular, with regard to the pass bandwidth, the expansion toward the high frequency side was particularly large, and the bandwidth of about 15 MHz was expanded toward the high frequency side. The passage characteristic is line 11
As shown in FIG.

In this state, the degree of out-of-band suppression is not sufficient. Therefore, one resonator R ₂ and R ₄ was added. By adding the one resonator R ₂ and R ₄ , the out-of-band suppression degree is reduced by about 5 as shown by the arrow 115 without narrowing the pass band width.
The dB was improved, and the transmission characteristics indicated by the line 111 were obtained.

Comparing line 111 with line 68, the arrow 11
As shown in FIG. 6, the loss is also improved as compared with the conventional case. The number of the resonators R ₂ and R ₄ of the series arm 61 may be three or more. Further, as shown by a two-dot chain line in FIG.
An inductance may be inserted between 2 and 64.

Embodiment 5 FIG. 21 shows a surface acoustic wave filter 120 according to a fifth embodiment of the present invention. In the figure, FIG.
The same reference numerals are given to the same components as those shown in FIG. Inductance value of the inductance L ₁ of the parallel arm 62 is 4 nH.

The inductance value of the inductance L ₂ of another parallel arm 63 is 5.5 nH. Still another parallel arm 6
Inductance value of the inductance L ₃ of 4 is 7nH. As described above, by making the inductance values of the inductances L _{1 to} L ₃ of the parallel arms 62 to 64 different, the filter 120 has a pass characteristic indicated by a line 121 in FIG.

Here, the inductances of the inductances L _{1 to} L ₃ are all equal to 4 nH.
Let's compare it with the zero pass characteristic. This filter 60 has a pass characteristic (see FIG. 12) indicated by a line 65 in FIG. In the pass characteristic of the filter 120 of the present embodiment, the degree of suppression outside the pass band is increased as shown by an arrow 122 without reducing the pass band width at all, as compared with the pass characteristic of the filter 60.

Looking at the lower frequency side than the pass band,
In the case of the filter 60 having only one attenuation pole 123 near 902 MHz, 875 MHz and 89
Attenuation poles 124 and 125 are generated at two places of 2 MHz. Thereby, the frequency band 126 between the two attenuation poles 124 and 125 becomes the stop band 127.

[Embodiment 6] FIG. 25 shows a surface acoustic wave filter 130 according to a sixth embodiment of the present invention. In this embodiment,
This is to reduce the loss. In the figure, portions corresponding to the components shown in FIG. 11 are denoted by the same reference numerals, and description thereof will be omitted.

The first surface acoustic wave resonator R _1B of the parallel arm 62
Has a configuration in which an excitation electrode 131 and reflectors 132 and 133 on both sides thereof are arranged as shown in FIG. Reflector 13
Reference numerals 2 and 133 denote the distance d between the center of the excitation electrode 131 and the reflectors 132 and 133 as follows: d = (n + β) · λ (where n is an appropriate integer, β is a real number of 1 or less, and λ is resonance (Which is the cycle of the comb-shaped electrode corresponding to the frequency).

The logarithm of the reflectors 132 and 133 is 50
It is. The resonator R _1B provided with a reflector is represented by a symbol added with “*” as shown in FIG. The other parallel arm 63,
The 64 resonators R _3B , R _5B are also provided with reflectors, similarly to the above-described resonator R _1B .

The filter 130 having the above configuration has a pass characteristic indicated by a line 134 in FIG. This passing characteristic is shown in FIG.
Compared with the pass characteristic of the first filter 60 (shown by the line 65), the insertion loss in the normal band is reduced as shown by the arrow 135. Here, as shown in FIG. 26, the ripple r _P is provided on both sides of the excitation electrode 131 of the parallel arm,
This is caused by arranging 133.

Here, the reason why the arrangement positions of the reflectors 132 and 133 are determined as described above will be described. In the above equation, the ripple r is changed by changing β from 0 to 0.5.
The effect of _{P on} the width is as shown by the line 140 in FIG. In the figure, a point 141 is a point where the ripple width is the minimum,
Β at this time is 0.4.

From this, β is set to 0.4.
FIG. 29 shows a surface acoustic wave filter device 150 that realizes the filter 130 of FIG. In the figure, portions corresponding to those shown in FIG. 14 are denoted by the same reference numerals, and description thereof will be omitted.

132, 133, 151, 152, 15
Reference numerals 3 and 154 denote reflectors. Next, a modification of the first one-port surface acoustic wave resonator will be described. FIG. 30 shows a modification. This resonator R ₁ Ba has the excitation electrode 1
In this configuration, the comb-shaped electrodes 160 and 161 whose electrical loads are short-circuited are arranged as reflectors on both sides of 31.

FIG. 31 shows another modification. The resonator R ₁ B _b is as a reflector on both sides of the excitation electrode 131, a configuration in which arranged strip array type electrodes 165 and 166. Seventh Embodiment FIG. 32 shows a surface acoustic wave filter 170 according to a seventh embodiment of the present invention. In this embodiment, the loss is reduced in the same manner as in the sixth embodiment. In the figure, the same reference numerals are given to the portions corresponding to the components shown in FIG. 21 and the description thereof is omitted.

The filter 170 is the filter 11 of FIG.
26, the first surface acoustic wave resonators R _1B , R _3B , and R _5B of each of the parallel arms 62, 63, 64 are respectively determined with β of 0.4 on both sides of the excitation electrode 131 as shown in FIG. This is a configuration in which reflectors 132 and 133 are arranged at positions. This filter 17
According to 0, a pass characteristic in which the loss in the pass band is smaller and the ripple is suppressed is obtained as compared with the characteristic indicated by the line 111 in FIG.

[Embodiment 8] This embodiment aims at removing the ripple r _P in FIG. First, a description will be given of means for effectively removing a ripple that appears when the reflector is added.

The inventors examined the relationship between the frequency position where ripples appeared and the electrode thickness by simulation. In the simulation, the effect of increasing the film thickness was replaced by increasing the ratio between the acoustic impedance under the electrode (Zm) and the acoustic impedance at the free surface (Zo). As described in the literature (1), the increase in the electrode film thickness means the increase in the mass, which is considered to be directly proportional to the increase in the discontinuity of the acoustic impedance. / Zm = Vo / Vm = 1 + k 2/2 + α (t) ... (9) (Vo, Vm: the speed of sound under the free surface and the electrode, k ^2: electromechanical coupling coefficient), and the thickness of the alpha (t) t This was varied as a parameter proportional to

With this arrangement, the center frequency fo of the filter becomes fo = 2fo '/ (1 + Q) (10). As the film thickness increases, the center frequency fo' becomes lower from the center frequency fo 'when there is no discontinuity in acoustic impedance. This is consistent with the well-known experimental fact of moving to the side.
As a result of the simulation, when α (t) is increased, that is, as the electrode film thickness is increased, the frequency position where the ripple r _P appears moves to the high frequency side of the pass band as shown by an arrow 180 in FIG. Eventually, it was found that it eventually fell into the attenuation pole on the high frequency side. This is schematically shown in FIG.

In FIG. 33, another ripple r _S is caused by the reflector of the series arm resonator. FIG.
Reference numeral 4 denotes a transmission characteristic when α (t) = 0.08, in which the ripple generated from the reflector of the resonator in the parallel arm falls just inside the attenuation pole on the high frequency side. Therefore, in the figure, the ripple disappears from the pass band, and the insertion loss is considerably reduced. In this figure, the center of the pass band is (1)
Since it has moved to the low frequency side according to the equation (0), in order to correct this, the resonance frequency of the resonators of the series arm and the parallel arm is shifted to the high frequency side by 15 MHz so that the center frequency becomes 932 MHz.

To see this in correspondence with the actual film thickness, a chip was prototyped and its passing characteristics were examined. FIG. 35 (A),
Lines 185, 186, and 187 in (B) and (C) show the corresponding transmission characteristics when the film thickness is 2000, 3000, and 4000, respectively. The center frequency changes by changing the film thickness. In the data shown in the figure, the period of the comb-shaped electrode is changed so as to correct the center frequency so that the center frequency does not change much.

As is apparent from FIG. 35, the ripple r _P of the resonator of the parallel arm and the ripple r _S of the serial arm out of the band, which appeared in the band at 2000 °, are 3000.
In the case of Å, it moves to the high frequency side and becomes r _P ', r _S ',
As a result of r _P ′ being buried in the attenuation pole on the high frequency side, good characteristics were obtained with no ripple in the band. This result agrees qualitatively with the simulation result.

However, loss degradation due to bulk waves that cannot be calculated by simulation when the film thickness is increased (Ebata et al .: “Surface acoustic wave resonator on LiTaO ₃ substrate and its application to VTR oscillator”, IEICE Transactions. Magazine, vol.
J66-C, No.1, pp23-30, 1988) and improvement in loss due to resistance loss, and the balance is also an important factor. Therefore, FIG.
The change of the minimum insertion loss when the film thickness was changed is plotted in (A).

In the figure, a line 190 indicates a loss due to a bulk wave,
Line 191 shows the loss due to ohmic loss. Line 192 is the experimental value
It is. As can be seen from the figure, the insertion loss is about 2500 °
And the effects of both are balanced, from about 3500Å to bulk
The increase in loss due to waves becomes dominant and begins to degrade. FIG.
The line 193 in FIG. 26B corresponds to the excitation electrode 131 in FIG.
R when the film thicknesses of the devices 132 and 133 are changed._P
The passband center frequency f at the frequency position ₀Changes to
Is shown.

FIG. 36A and FIG. 36B are comprehensively determined.
And the film thickness is in the range of 2600 to 4000
Is suitable because it does not produce ripples and has little loss deterioration.
That's right. This is almost determined from the center frequency of the filter.
Electrode period λ of the parallel arm resonator _P(4.4 at 932 MHz)
μm, see FIG. 26).
0.09.

This embodiment is based on the results of the above study. FIG. 37 shows a first one-port surface acoustic wave resonator 200 of a surface acoustic wave filter according to the present invention. In the figure, 20
1 Excitation electrodes 202 and 203 are reflectors, each of which is made of Al or made of Al mixed with different metals by several% by weight.
The film thickness t ₁ is 0.06 to 0.09 times the electrode period λp.

The pass characteristics of a surface acoustic wave filter in which this resonator 200 is applied to the resonators R _1B , R _3B , and R _5B in FIGS. 25 and 32 are as shown by a line 205 in FIG.
No ripple appears in the passband. In addition, when made of the above-mentioned Al alloy, the power handling characteristics are improved as compared with the case of made of Al. The dissimilar metals to be mixed are Cu, T
i.

FIG. 39 shows a resonator 210 which is a modification of the above resonator. 211 is an excitation electrode, and 212 and 213 are reflectors. These are made of Au. Since this phenomenon occurs due to the effect of the mass addition effect, the optimum film thickness value range becomes smaller than the above value by the ratio to the density of Al.

Al density / Au density = 2.7 / 18.
Since 9 = 0.143, the optimum thickness t ₂ is 0.14
The thickness is 0.0086 to 0.013 times as large as the electrode period λ _P by three times. FIG. 25 and FIG.
The pass characteristic of the surface acoustic wave filter applied to the resonators R _1B , R _3B , and R _5B in FIG. 2 also becomes as shown in FIG. 38, and no ripple appears in the pass band.

[Embodiment 9] This embodiment is another example of realizing the inductances L ₁ , L ₂ and L ₃ in FIG.
In FIG. 40, portions corresponding to the components shown in FIG. 14 are denoted by the same reference numerals, and description thereof will be omitted.

[0093] 220 and 221 is a zigzag microstrip lines, are formed on the ceramic package 81 extend from the respective terminals 84 _-3 and 84 _-5.
The tips of the microstrip lines 220 and 221 are connected to the ground. Each microstrip line 22
The pattern width of 0,221 is 100 μm, and the length between the microstrip line and the ground is 0.5 mm.

The relative permittivity of the ceramic package 81 is 9
Then, from the theoretical formula of the ribbon inductor, the inductance value of the microstrip lines 220 and 221 is 2 nH. Example 10 This example is yet another example of realizing the inductance L _1, L _2, L ₃ in FIG.

In FIG. 41, portions corresponding to the components shown in FIG. 14 are denoted by the same reference numerals, and description thereof will be omitted. 23
Reference numerals 0 and 231 denote zigzag microstrip lines which extend from the resonators R ₁ and R ₂ and are formed on the filter chip 82. The tip of the micro strip line 230 and 231, terminals 85 _-3, are then 85 _-5 formation.

Each microstrip line 230, 23
1 has a thickness of 3000 mm, a width of 60 μm, and a total length of about 2 m
m. Assuming that the relative permittivity of the filter chip (LiTaO ₃ ) 82 is 44, the microstrip line 23
The inductance value of 0,231 is 2.2 from the theoretical formula.
nH.

[0097] Incidentally, the inductance can also be formed by combining the bonding wire 86 _-3, the microstrip line 220 on the ceramic package 81, the microstrip line 230 on the filter chip 82 as appropriate. Embodiment 11 FIG. 42 shows a surface acoustic wave filter 240 according to an eleventh embodiment of the present invention. FIG. 43 shows a configuration embodying this.

For convenience of explanation, an outline of the present embodiment and a basic configuration of the present embodiment will be described first. In the present embodiment, the resonance frequency frs of the resonator in the series arm is appropriately set higher than the anti-resonance frequency fap of the resonator in the parallel arm to increase the pass bandwidth, and Δf≡ frs −fap
Is set within a range that does not significantly degrade the loss in the pass band.

Basic Configuration of this Embodiment In each of the above embodiments, fap = frs is an essential condition for forming a pass band of a filter. However, as long as this condition is maintained, there is an upper limit in the pass band. Therefore, in order to increase the passband, we considered fap <frs as shown in FIG.

As a result, as is apparent from FIG.
In the range of p <f <frs, bx <0, and there is a possibility that an attenuation range may be obtained from the above-described theory. However, in reality, if the magnitude of Δf (= frs−fap) is limited as described below, the value of the bx product remains at a very small value, so that no attenuation occurs, and there is virtually no problem as a pass band. It turned out there was no.

FIG. 45 shows the pass characteristic of the ladder filter when Δf = frs−fap is increased from zero. As an experimental condition, the piezoelectric substrate has an electromechanical coupling coefficient of 0.05.
Of LiTaO ₃ and an Al electrode for a comb-shaped electrode having a thickness of 300
A condition of 0 ° was used. The electrode configuration is based on a ladder-type configuration in which a parallel resonator and a series resonator are connected as shown in FIG. 42, and these are cascaded in two stages. The resonator is connected to the last stage. The reason why the ladder type circuit is multi-staged is to increase the degree of out-of-band suppression to a practical value.

However, since the number of stages increases the loss in the pass band, the number of stages is adjusted according to the specific filter specifications. This example is an example of a configuration for achieving a loss of 2 dB or less and an out-of-band suppression of 20 dB or more. The design conditions of the comb-shaped electrode are as follows. Both resonators of the series-parallel arm have an opening length of 180 μm and a logarithm of 50 pairs. Since the logarithm and the opening length conditions of the series-parallel resonator are the same, the ratio P = Cp / Cs of the respective capacitances is 1.

FIG. 45A shows the case where Δf = 0, that is, the case of the above embodiment. FIG. 6B shows Δf
= 10 MHz. Compared with FIG. 7A, the minimum insertion loss of the pass band is hardly degraded, and the pass band width (the band that guarantees a loss of 2.5 dB or less) is 22.
From 40 MHz to 40 MHz. The reason why the bandwidth is improved more than the increase in Δf is that loss recovery on the low frequency side of the pass band is observed.

Further, the degree of out-of-band suppression is also improved. FIG.
In (A) and (B), the degree of out-of-band suppression (shown in the figure) on the high frequency side is improved from 19 dB to 20 dB.
As described above, the expansion of Δf is a technique that not only has an effect on bandwidth expansion but also can improve out-of-band suppression at the same time. As described above, when Δf is increased, the characteristics are improved, but the amount of increase is limited.

FIG. 45 (C) is a pass characteristic diagram when Δf = 19 MHz. Loss degradation starts to be seen on the slightly higher frequency side in the pass band. In this case, it is about 2.5 dB.
This also causes an increase in in-band ripple. In this example, the ripple specification limit is about 1.0 dB. Any further increase in Δf resulted in loss degradation and increased in-band ripple. Therefore, Δf = 19 MHz is the limit when increasing Δf. Further, the degree of out-of-band suppression at this time is about 21 dB as shown in FIG.
An improvement of 2 dB is obtained as compared with 5 (A).

At this time, the value of the above-mentioned bx product was examined by taking the case of Δf = 19 MHz in FIG. 45C as an example. First, the surface acoustic wave resonator forming the parallel arm and the surface acoustic wave resonator forming the series arm shown in FIG. 42 are separately manufactured, and the circuit configuration shown in FIGS. The arm resonator measured admittance, and the series arm resonator measured impedance. Measurements performed using a network analyzer to measure the respective S _21. Then, the values were substituted into the equations shown in FIGS. 46A and 46B to obtain the impedance Z _p and the admittance Y _p .

As a result, a frequency characteristic as shown in FIG. 47 was obtained. This characteristic is the value of only the imaginary part of admittance and impedance, that is, the value of b or x. When the value of the bx product is calculated from these, the frequency characteristics are as shown in FIG. It can be seen from the figure that the bx product has a negative and small value in the range of fap <f <frs.

The maximum value | bx _max | of the absolute value of the bx product is

[0109]

(Equation 5)

At this time, it was 0.06 in this embodiment. That is, when the value of | bx _max | is equal to or less than this value, it is understood that the deterioration of the insertion loss and the in-band ripple can be suppressed to 1 dB or less. Δf> 19MH
Assuming that z, the | bx _max | value also increases, and both the loss deterioration and the in-band ripple become 1 dB or more, which is not practical.

Therefore, the value | bx _max | is an index of the upper limit of the characteristic deterioration, and determines the allowable value of Δf. The following is a more generalized and detailed description. FIG. 49 shows an equivalent circuit diagram when the surface acoustic wave resonator is approximated by an LC double resonance circuit as in FIG. 3 and assembled into a ladder type filter as shown in FIG.

Assuming that the impedance of the surface acoustic wave resonator of the series arm is Zs and the admittance of the surface acoustic wave resonator of the parallel arm is Yp,

[0113]

(Equation 6)

Is obtained. here,

[0115]

(Equation 7)

Is as follows. When the bx product is obtained from equations (11) and (12),

[0117]

(Equation 8)

Is obtained. The angular frequency ω giving an extreme value to bx in the equation (13) is obtained from ∂ (bx) / ∂ω = 0, and

[0119]

(Equation 9)

Is obtained. The value obtained by substituting this into the expression (13) becomes the maximum value of the absolute value of the bx product in the pass band. If you ask for this

[0121]

(Equation 10)

Is obtained. Here, Δω = ω _rs −ω _ap = 2π · Δf (16) When the equation (15) is plotted using P = C _op / C _os as a parameter as a relationship between bx _max and Δf / _frs , the result is as shown in FIG.

In the figure, if the condition of the allowable value of the product of bx previously obtained experimentally of 0.06 or less is shown, the area becomes a hatched area. Therefore, acceptable value α is determined for P = C _op / C _os by different Delta] f / f _rs, it (15)
Assuming that | bx _max | = 0.06 in the equation, the following equation is obtained.

[0124]

[Equation 11]

The capacity ratio γ depends on the substrate material, and
For example, 36 ° Y-cut propagation LiTaO _ThreeAbout 15
Was. Therefore, equation (17) becomes

[0126]

(Equation 12)

Is obtained. When P = 1, α = 0.02 and Δf in the case of the embodiment of FIG. 45 where f _rs = 948 MHz.
= 19 MHz, and it can be confirmed that equation (18) holds. The effect of increasing Δf is effective for a piezoelectric substrate material having a small capacitance ratio γ, that is, a substrate material having a large electromechanical coupling coefficient. Equation (17) was obtained for such a piezoelectric substrate material.

Since the γ value is proportional to the reciprocal of the electromechanical coupling coefficient k ² , another high electric power is obtained by using the γ value of 36 ° Y cut X propagation LiTaO _{3 and} the value of k ² = 0.05. Material with mechanical coupling coefficient 64 ° Y-cut X-propagation LiNbO ₃
(K ² = 0.11) and 41 ° Y-cut X-propagation LiNb
When the γ value of O ₃ (k ² = 0.17) is obtained, the former is 6.
8. The latter is 4.4. These k ² values are described in the literature (“Applications for Piezoelectric Leaky Surface W
ave ": K.Yamanouchi and M.Takeuchi, 1990 ULTRASONICS
SYMPOSIUM Proceedings, pp11-18, 1990).

FIG. 51 shows the relationship between the capacitance ratio γ and the electromechanical coupling coefficient k ² . Relationship shown in FIG uses the value of the 36 ° Y k ² and γ value of the cut propagation LiTaO _3,

[0130]

(Equation 13)

Is obtained. From the relationship of FIG. 51, the X propagation LiTa of the 64Y ° cut and the 41 ° Y cut
The γ value of O ₃ can be obtained, and γ = 6.8 and 4.4 as described above, respectively. Configuration of Embodiment 11 Here, the configuration of the embodiment shown in FIGS. 42 and 43 will be described.

Numeral 241 denotes a 36 ° Y-LiTaO ₃ piezoelectric substrate having a size of 1.5 × 2 × 0.5 mm. In order from the input side, a parallel arm resonator (Rp ₁ ), a series arm resonator (Rs ₁ ), a parallel arm resonator (Rp ₂ ), a series arm resonator (Rs ₂ ), and a parallel arm resonator (Rp ₃ ) They are arranged in order.

The individual resonators have reflectors 242 on both sides.
(Short-circuit type). Each resonator has an opening length of 180 μm, the number of pairs of electrode fingers is 50, and the number of reflectors is 50. Only the period of the comb-shaped electrode fingers is changed between the parallel arm resonator and the series arm resonator. The period of the parallel arm resonator is λ
p = 4.39 μm (Since the pattern width and the gap are 1: 1, the pattern width is λp / 4 ≒ 1.1 μm), and the series resonator period is λs = 4.16 μm (similarly, the pattern width is λs / 4 = 1.04 µm).

In each cycle, the resonance frequency (frp, frs) of each resonator is a predetermined value (frp = 893 MHz,
frs = 942MHz) and so as to λs = V _m / frs, and is obtained by determining from λp = V _m / frp. Here, V _m is 30
The sound velocity of a surface wave of a 36 ° Y-cut X-propagating LiTaO ₃ crystal at 00 °, which was experimentally determined to be V _m = 3920 _m / s.

The surface acoustic wave filter 240 having the above configuration is
It has a broadband and low-loss transmission characteristic shown in FIG. Note that Δf = 19 MHz. In FIG. 43, λp
Is changed to 4.35 μm, Δf is 10 MHz
And the characteristic shown in FIG. 45 (B) is obtained.

The electrode material is an Al—Cu alloy, the thickness is 3000 °, and the electrodes are arranged so that the surface wave propagates in the X-axis direction of the piezoelectric substrate 241. Next, an example in which another piezoelectric substrate is used will be described. In the case of 64 ° Y-cut X-propagation LiNbO ₃ , γ = 6.8 (17)
ceremony,

[0137]

[Equation 14]

Is obtained. 41 ° Y-cut X-propagation LiNbO
_In the case of ₃ , γ = 4.4,

[0139]

(Equation 15)

Is as follows. As the γ value decreases, that is, as the substrate has a larger electromechanical coupling coefficient, α increases and Δ
Even if f is greatly increased, characteristic deterioration hardly occurs. [Twelfth Embodiment] FIG. 52 is a circuit diagram of a surface acoustic wave filter 250 according to a twelfth embodiment of the present invention.

FIG. 53 shows a structure embodying the surface acoustic wave filter having the circuit configuration of FIG. 54 and 55
Shows the characteristics of the surface acoustic wave filter shown in FIGS. Outline of Embodiment For convenience of explanation, first, an outline of the present embodiment will be described.
The surface acoustic wave filter according to the present embodiment is a ladder type surface acoustic wave filter in which a plurality of surface acoustic wave resonators are connected in series and in parallel with each other. The configuration is such that the image impedance between them is matched to reduce the loss at each connection point.

Thus, the insertion loss in the pass band can be reduced. Thinking process until completion of the invention Next, a thought process until completion of the present invention will be described.
As shown in FIGS. 56 (A) and 56 (B), bandpass characteristics can be obtained by ladder-type connection of at least one series arm resonator and parallel arm resonator. The ladder-type connection of the series arm resonators and the parallel arm resonators one by one constitutes a unit section of the filter.

At this time, it is desirable that the resonance frequency of the series arm resonator and the anti-resonance frequency of the parallel resonator match, or that the former has a higher frequency than the latter in view of the increase of the passband. There are two types of unit sections in FIGS. 56 (A) and (B) depending on whether the input / output ends are serial arms, and those connected in multiple stages are shown in FIGS. 57 (A), (B) and They are classified into three types as shown in FIG.

FIG. 57 (A) shows the case where one of the input / output sides is a serial arm and the other is a parallel arm (asymmetric type), and FIG. 57 (B) shows the case where the input / output end is a parallel arm (symmetric type). FIG. (C) shows a case where both the input and output terminals are series arms (symmetric type). When the number of stages is increased in this way, both the insertion loss and the degree of out-of-band suppression are n times as large as the unit section, and although the insertion loss generally worsens, the degree of out-of-band suppression is improved. In particular, when the loss in the unit section is close to 0, this multi-stage is effective.

However, if the impedance matching in the pass band between the unit sections is not appropriate, the insertion loss will be worse than the theoretical n times. If the impedance matching is not appropriate, the boundary of the unit section (line 1-
This is because power reflection occurs at each boundary from 1 ′ to nn ′), resulting in an increase in loss.

Assuming that the power reflection between the unit sections is Г, the loss is also n10log (Г). Therefore, it is important to reduce the insertion loss as much as possible by measuring the impedance between the unit sections and suppressing the power reflection at the boundary. Next, a method for achieving impedance matching between unit sections will be described. As shown in FIG. 58, generally, two different four-terminal constants (four constants A,
In the case where circuits having B, C, and D) are connected with each other for impedance matching, it is only necessary that the image impedances when the respective circuit sides are viewed from the boundary bb ′ are equal to each other.

As shown in FIG. 58, the image impedance Z _i1 of the circuit 1 as viewed from the side of the circuit 1 is represented by four terminal constants A ₁ , B
_It is expressed as follows using ₁ , C ₁ and D ₁ .

[0148]

(Equation 16)

Similarly, the image impedance Z _i2 of the circuit 2 is expressed by the following equation.

[0150]

[Equation 17]

These image impedances are determined independently of the load resistance (pure resistance) R ₀ . If the equations (21) and (22) are assumed to be equal, the impedance matching condition as follows is obtained. D ₁ B ₁ / C ₁ A ₁ = A ₂ B ₂ / C ₂ D ₂ (23) FIG. 59 shows a case where the above-described impedance matching condition is applied to a unit section of a ladder-type circuit.

FIG. 59A shows a connection method with poor impedance matching, which does not satisfy the condition of equation (23). Boundary b
The reflection coefficient た as viewed from the right side from −b ′ is

[0153]

(Equation 18)

Is obtained. In actual elements, Z _s Y _p does not become completely zero even in the pass band, so that Г does not become zero.
On the other hand, in FIG. 59 (B) or FIG. 59 (C), since the condition of the expression (23) is satisfied at the boundary bb ′, the reflection becomes 0 and no loss occurs. For example, in the case of FIG. 59B, the image impedance as viewed from the left side from the boundary bb ′ is:
From equation (21),

[0155]

[Equation 19]

Is obtained. The image impedance Z _i2 as seen from the boundary bb ′ on the right side is also obtained from equation (22).
It turns out that it becomes equal to _i1 . Therefore, impedance matching is achieved, and the reflection coefficient at the boundary becomes zero. FIG. 59 (C)
It is also proved that impedance matching has been achieved. Next, a method of connecting the unit sections in multiple stages using the connection method as shown in FIGS.

FIG. 60A shows a circuit in which n (> 2) unit sections are connected by repeating the connection method shown in FIGS. 59B and 59C alternately. If such a connection method is used, power reflection in each unit section does not occur regardless of how many stages are connected for the above-described reason. In the configuration shown in FIG. 60A, adding resonators of parallel arms adjacent to each other or resonators of serial arms and adding them together is equivalent to FIG. 60B.

As a result, it can be seen that only the arm closest to the input / output end has an impedance or admittance value that is half that of the inner arm. When this principle is applied to the three types of multi-stage methods shown in FIG. 57, the methods shown in FIGS. 61 (A), (B) and (C) are obtained as connection methods for impedance matching.

FIG. 61A shows a matching connection method corresponding to FIG. 57A, in which one of the input / output terminals is a serial arm and the other is a parallel arm. In this case, the impedance value of the series arm resonator at the end is half the impedance value of the inner series arm resonator, and the admittance value of the parallel arm resonator at the other end is also equal to the inner parallel arm resonator. Half the admittance value of the vessel.

Similarly, FIG. 61 (B) shows the matching connection method of FIG. 57 (B), and FIG. 61 (C) shows the matching connection method of FIG. 57 (C).
In the case of FIG. 61B, both ends are parallel arms, and the admittance value of the parallel arm resonators at both ends is half of the admittance value of the inner parallel arm resonators.

FIG. 61 (C) shows a case where both ends are series arms, and the impedance value of the series arm resonators at both ends is half of the impedance value of the series arm resonator inside it. Next, a twelfth embodiment of the present invention based on the above concept will be described.

FIG. 52 shows the elasticity according to the twelfth embodiment of the present invention.
3 shows a basic configuration of a surface acoustic wave filter 250. This elastic table
When the surface wave filter is embodied, it becomes as shown in FIG.
You. Three series arm resonators (Rs₁, Rs_Two, Rs_Three)When
Three parallel arm resonators (Rp ₁, Rp_Two, Rp_Three) And from
Are connected as shown in the equivalent circuit of FIG.
ing.

These six resonators all have an aperture length (90
μm) and the same number of electrode finger pairs (100 pairs). Each resonator has a short-circuited reflector on both sides as shown in FIG. The logarithm of the reflector is about 100 pairs.

The series arm resonators (Rs _{1 to} Rs ₃ ) all have the same length of electrode finger period (λs), and λs = 4.19.
μm. The period of the parallel arm resonators (Rp _{1 to} Rp ₃ ) is set to a different period λp = 4.38 μm. FIG. 62 shows a conventional configuration for this configuration as a comparison target.

[0165] For both FIGS. 52 and 62, the design conditions of the one-terminal SAW resonator series arm represented by impedance Z _s is the aperture length 90 [mu] m, a logarithmic 100 pairs. One-terminal-pair surface acoustic wave resonator in the parallel arm represented by admittance Y _p is the same condition. The piezoelectric substrate crystal has 36
A Y-cut X-propagation LiTaO ₃ is used, and a 3000 ° Al alloy film comb pattern is formed thereon as a surface acoustic wave resonator.

In FIG. 54, the solid line 251 is the filter of FIG.
250 are shown. Dashed line 252 is the conventional filter of FIG.
Shows the characteristics of ruta. From both, the filter 250 of the present embodiment
It can be seen that the loss is reduced. Especially the pass band
The improvement at both ends of the region is significant. Next, FIG.
Admittance Y of unit section (3)_pRepresented by
Reduced the logarithm from 100 pairs to 80 pairs
Admittance Y _pPass characteristics when the value of
In FIG. 55, this is indicated by a line 253. As well as improved insertion loss
You can see that it is. Therefore, the admittance value at the end is
Reduce the inner admittance value without making it 1/2
Even if alone is not enough, it can be said that it is effective. Impi
The same applies to the dance value.

Although the embodiment for the basic form shown in FIG. 61A has been described above, the same effect can be obtained even if a large number of unit sections increase in the center. Embodiment 13 FIG. 63 shows a surface acoustic wave filter 260 according to a thirteenth embodiment of the present invention. This surface acoustic wave filter is based on the configuration shown in FIG.

This surface acoustic wave filter 260 has the structure shown in FIG.
A loss reduction effect similar to that shown by the line 251 is obtained. [Embodiment 14] FIG. 64 shows a surface acoustic wave filter 270 according to a fourteenth embodiment of the present invention. This surface acoustic wave filter is based on the configuration shown in FIG.

This surface acoustic wave filter 270 also has the configuration shown in FIG.
A loss reduction effect similar to that shown by the line 251 is obtained. Embodiment 15 Next, a surface acoustic wave filter according to a fifteenth embodiment of the present invention will be described with reference to FIGS.

Outline of Embodiment For the convenience of explanation, first, the outline of this embodiment will be described.
This embodiment focuses on the resistance and conductance of the comb-shaped electrode as the cause of determining the insertion loss in the pass band, and reduces the resistance for the series-arranged resonators.
For a parallel arm resonator, the insertion loss of the filter characteristics when these are assembled in a ladder form is reduced by reducing the conductance.

Next, the background of the present invention will be described. Background of the Invention FIG. 65 shows that the resonance frequency (frs,
4 shows a basic configuration of a ladder-type filter circuit in which two surface acoustic wave resonators having different frp) are arranged. Here, the admittance of the parallel arm resonator is represented by Y _p = g + j · b (26) g: conductance b: susceptance

The impedance of the series arm resonator is given by Z _s = r + j · x (27) where r is a resistance and x is a reactance.

Assuming the above, the frequency characteristics of g, b, r, and x are as shown in FIG. Susceptance component b admittance Y _p of the parallel arm resonator (dotted line in FIG. 69)
Takes the maximum value at the resonance frequency frp, where the sign changes from + to −, and becomes 0 (zero) at the antiresonance frequency fap, and fap
With the above, the sign becomes + again and gradually increases. On the other hand, the conductance g of Y _p (dashed line in FIG. 69)
Similarly takes the maximum value at fap, decreases rapidly beyond fap, and gradually approaches zero.

The conductance g has only a positive value. Code to - reactance component x of the impedance component Z _s of the series arm resonators (solid line in FIG. 69) is, admittance and becomes 0 in reverse at the resonance frequency frs takes the maximum value at the antiresonance frequency fas, further from + , And from fas it approaches 0 from one side.

The resistance r gradually increases from 0, reaches a maximum value at the anti-resonance frequency fas, and gradually decreases at and above that. r also takes only the value of + similarly to g.
Here, in order to create a filter characteristic, it is required that the anti-resonance frequency fap of the parallel resonator is substantially equal to the resonance frequency frs of the series resonator, or that the latter is slightly larger.

The lower part of FIG. 69 shows the pass characteristics as a filter circuit in accordance with the frequency characteristics of the upper impedance and admittance. Take the passband near fap ≒ frs,
Otherwise, it is an attenuation region. As can be seen from the figure, b and x become 0 especially in the vicinity of the center frequency of the pass band.

Therefore, the pass characteristic as a filter is S ₂₁
Is determined only by r and g,

[0178]

(Equation 20)

Is as follows. Here, r> 0 and g> 0
In equation (28), S increases as both r and g increase. _{twenty one}Is more than 1
Smaller, 20 log | S_{twenty one}Insertion loss represented by |
It increases. Therefore, as r and g are both closer to 0,
The loss will be small.

Next, a description will be given of what portions of the comb-shaped electrodes forming the surface acoustic wave resonator have r and g. Here, in FIG. 3B, r ₁ is also taken into consideration. r ₁ is the sum of the electrical resistance of the comb-shaped electrode and the energy loss that leaks as a bulk wave from each end of the comb-shaped electrode finger into the substrate as acoustic resistance.

Since the resistance due to the emission of the bulk wave hardly depends on the shape of the comb-shaped electrode, the electric resistance r _{1 of the} comb-shaped electrode is small.
Is proportional to Especially near the center frequency of x = 0, r = r ₁
Becomes The admittance conductance g of the parallel arm resonator is equal to the conductance 1 of the electric resistance of the comb-shaped electrode.
/ R ₁ .

Now, assuming that the resistivity of the electrode finger of the comb-shaped electrode is ρ _o , the width of the electrode finger is W, the film thickness is t, the opening length of the series arm resonator is l _s , and the logarithm is N _s , r = l _s · ρ _o / (N _s · W · t) (29) Further, the opening length of the parallel arm resonator is l _p , and the logarithm is N
_{If p} is the same substrate and the same metal film is used, ρ _o ,
W, t is from approximately equal, and _{g = N p · W · t} / (l p · ρ o) ... (30).

[0183] Thus, the increase in insertion loss in the equation (28) is, r + 50r · g + 2500g = l s · ρ o / (N s · W · t) +50 · (l s / l p) · (N p / N _{s) +2500 · N p · W} · t / (l p · ρ o) ... is (31).

From equation (31), the series arm resonator has an aperture length l
_s is short, as the logarithmic N _s is large, also, the parallel resonators aperture length l _p is long, as the logarithmic N _p is small, it can be seen that the effect on the loss reduction. In particular, l _s / l _p <1, N _p / N _s
In other words, as the value is <1, the opening length of the serial arm resonator is shorter than that of the parallel arm resonator, and the logarithm of the serial arm resonator is larger than that of the parallel arm resonator. .

Here, the reason will be described. The (31) In the _{equation, r = r s (r s} : series arm resonators electrical resistance) and g = 1 / r _p: from a (r _p parallel electrical resistance arm) r + 50r · g + 2500g = r s +50 (R _s /
r _p ) +2500 (1 / r _p ). _{Therefore, (r s / r p)} <1, ie r _s <r _p
Then, an increase in insertion loss can be suppressed.

In this case, if l _s is too narrow, loss due to surface wave diffraction appears. Conversely, if l _p is too long, the Q of the parallel resonator is reduced due to an increase in resistance, and out of band on the low frequency side. Since the degree of suppression is worse, its size is limited. Further towards the t _s of the film thickness series arm of the metal film forming the comb-shaped electrodes, towards the parallel arm When t _p (31)
The formula is as follows:

[0187] r + 50r · g + 2500g = l s · ρ o / (N s · W · t) +50 · (l s / l p) · (N p / N s) (t p / t s) +2500 · N p · W · t _p / (1 _p · ρ _o ) (32) Accordingly, the loss can be similarly reduced by setting t _p / t _s . In addition, a filter made by arranging resonators made of two kinds of metal films (ρ _os , ρ _op ) having different resistivities in a series arm and a parallel arm may be used to make ρ _os / ρ _op <1. Although it is possible, it is not practical when actually manufacturing an element in consideration of mass productivity and the like.

Configuration of Embodiment 15 Next, an embodiment employing the above concept will be described.
FIG. 65 shows a surface acoustic wave filter 2 according to a fifteenth embodiment of the present invention.
80 shows a circuit configuration. FIG. 66 shows a structure embodying the circuit configuration of FIG.

The piezoelectric substrate 241 used was 36 ° Y cut X
LiTaO ₃ , and the electrode material is a 3000 ° Al film. Conventionally, the series arm, in the parallel arm both aperture length l _s = l _p = 90μm interdigital, whereas a logarithmic N _p = N _s = 100 pairs, in the present embodiment, the series arm, l _s = 45μ
m, N _s = 200 pairs, the parallel arm is l _p = 180 μm, N _p
= 50 pairs. l _p > l _s and N _s > N _p . Further, l _s / l _p = 0.25 and N _p / N _s = 0.
25.

At this time, the capacitance C _O of the comb-shaped electrode, which is determined by the product of the logarithm and the opening length, was kept unchanged. The solid line 281 in FIG. 66 is the characteristic of this embodiment, and the broken line 282 is the characteristic of the conventional example. The loss was 2.5 dB in the prior art, but 2.0 dB according to the present embodiment, and the loss was improved by 0.5 dB or more according to the present embodiment. That is, the insertion loss of the filter was improved by 25% in dB.

Further, in the case of the present embodiment, by increasing the logarithm of the series arm resonator, the power durability was improved, and the maximum power that could be applied was improved by 20%. In the case of the above embodiment, l
_s = _30μm began to appear the diffraction loss in the following, l _p = 300μ
Since the out-of-band degradation on the low frequency side began to occur at m or more, these values were the limits.

As described above, it is apparent that the insertion loss in the pass band can be improved by lowering the electric resistance of the series arm and increasing the electric resistance of the parallel arm (reducing the conductance). Further, it is also possible to adopt a configuration in which the film thickness of the parallel arm resonator is smaller than the film thickness of the series arm resonator. With this configuration, as in the case of the above-described embodiment, the loss of the pass band can be reduced.

[0193]

As described above, according to the first aspect of the present invention, the degree of suppression outside the pass band can be increased and the pass band width can be increased as compared with the conventional one. Loss can be reduced. According to the second aspect of the invention, in addition to the above-described effects, it is possible to reduce the loss in the pass band and suppress the ripple.

[Brief description of the drawings]

FIG. 1 is a principle diagram of a surface acoustic wave filter according to the present invention.

FIG. 2 is a diagram showing a basic configuration of a filter circuit using a resonator.

FIG. 3 is a diagram showing the structure of a one-port surface acoustic wave resonator, its equivalent circuit, and its symbol.

FIG. 4 is a diagram showing frequency characteristics of impedance and admittance of a one-port surface acoustic wave resonator.

5 is a diagram showing the immittance characteristics of the surface acoustic wave resonator near the resonance frequency and the filter characteristics of the filter of FIG. 1 formed by connecting them.

6 is a diagram illustrating the conventional surface acoustic wave filter of FIG.

FIG. 7 is a diagram illustrating an effect when an inductance is added to a resonator in series.

FIG. 8 is a diagram showing an effect when n one-port surface acoustic wave resonators are connected in series.

FIG. 9 is a diagram showing the dependence of the pass characteristic of the parallel arm resonator on the aperture length.

FIG. 10 is a diagram showing the aperture length dependence of the transmission characteristic of the series arm resonator.

FIG. 11 is a circuit diagram of a first embodiment of a surface acoustic wave filter according to the present invention.

FIG. 12 is a diagram illustrating a pass characteristic of the filter of FIG. 11;

FIG. 13 is a diagram showing the effect of adding inductance to the parallel arm resonator.

FIG. 14 is a plan view showing the structure of the surface acoustic wave filter of FIG. 11 with its lid removed.

FIG. 15 is a sectional view taken along line XV-XV in FIG. 41;

FIG. 16 is a view showing a second embodiment of the surface acoustic wave filter according to the present invention.

FIG. 17 is a diagram illustrating a pass characteristic of the filter of FIG. 16;

18 is a diagram showing the effect of increasing the aperture length of the parallel arm resonator (A _P) and aperture length of the series arm resonator ratio _{(A S) (A P /} A S).

FIG. 19 is a diagram showing a third embodiment of the surface acoustic wave filter according to the present invention.

FIG. 20 is a diagram showing the pass characteristics of the filter of FIG. 19;

FIG. 21 is a diagram showing a fourth embodiment of the surface acoustic wave filter according to the present invention.

FIG. 22 is a diagram illustrating pass characteristics of the filter of FIG. 21;

FIG. 23 is a diagram showing a fifth embodiment of the surface acoustic wave filter according to the present invention.

FIG. 24 is a diagram showing the pass characteristics of the filter of FIG. 23;

FIG. 25 is a circuit diagram of a surface acoustic wave filter according to a sixth embodiment of the present invention.

FIG. 26 is a diagram showing a first one-port pair elastic surface resonator in FIG. 25;

FIG. 27 is a diagram showing the pass characteristics of the filter of FIG. 25;

FIG. 28 is a diagram illustrating the influence of β on the ripple width due to the reflector installation position d = (n + β) · λ.

FIG. 29 is a plan view showing the structure of the surface acoustic wave filter of FIG. 25 with its lid removed.

FIG. 30 is a diagram showing a modification of the first one-port surface acoustic wave resonator shown in FIG. 25;

FIG. 31 is a view showing another modification of the first one-port surface acoustic wave resonator shown in FIG. 25;

FIG. 32 is a view showing a seventh embodiment of the surface acoustic wave filter according to the present invention.

FIG. 33 is a diagram showing an effect of an electrode film thickness (t) on a ripple occurrence position.

FIG. 34 is a diagram illustrating a state where a ripple (r _P ) due to a reflector of a parallel arm resonator falls to a high frequency attenuation pole.

FIG. 35 is a diagram showing the film thickness dependence of the pass characteristic of the resonator type filter.

FIG. 36 is a diagram showing the results of an experiment on the dependency of the insertion loss and the position of occurrence of ripple on the film thickness.

FIG. 37 is a diagram showing a first one-port surface acoustic wave resonator according to an eighth embodiment of the surface acoustic wave filter of the present invention.

FIG. 38 is a diagram showing the pass characteristics of a surface acoustic wave filter to which the resonator shown in FIG. 37 is applied;

FIG. 39 is a view showing a modification of the first one-port surface acoustic wave resonator according to the eighth embodiment of the surface acoustic wave filter of the present invention.

FIG. 40 is a diagram showing another example for realizing the inductance of the surface acoustic wave filter of FIG. 11;

41 is a diagram showing still another example for realizing the inductance of the surface acoustic wave filter of FIG. 11;

FIG. 42 is a circuit diagram of an eleventh embodiment of a surface acoustic wave filter according to the present invention.

FIG. 43 is a diagram showing a configuration embodying the circuit of FIG. 42;

FIG. 44 is a diagram showing the immittance characteristics of a surface acoustic wave resonator when fap <frp.

FIG. 45 is a diagram illustrating a change in the pass characteristic of the ladder-type filter when Δf≡frs−fap is increased from zero.

FIG. 46 is a diagram illustrating a method for measuring characteristics of a surface acoustic wave resonator.

FIG. 47 is a diagram illustrating characteristics of admittance and impedance of each surface acoustic wave resonator of the parallel arm and the serial arm.

FIG. 48 is a diagram showing the frequency dependence of the bx product.

FIG. 49 is a diagram showing a part of the circuit of FIG. 42 as an equivalent circuit of L and C.

FIG. 50 is a diagram showing a relationship between | bx _max | and Δf / frs.

FIG. 51 is a diagram showing a relationship between k ² and γ.

FIG. 52 is a circuit diagram of a twelfth embodiment of a surface acoustic wave filter according to the present invention.

FIG. 53 is a diagram showing a configuration embodying the circuit of FIG. 52;

FIG. 54 is a diagram showing characteristics of the surface acoustic wave filter of FIG. 53.

In the filter of FIG. 55 FIG. 63 is a diagram showing a characteristic when reduced output Y _p.

FIG. 56 is a circuit diagram of a unit section in which one surface acoustic wave resonator is formed in a ladder shape.

FIG. 57 is a circuit diagram of a circuit in which the unit sections of FIG. 56 are connected in multiple stages (n stages).

FIG. 58 is a diagram showing connections between two four-terminal circuits and boundaries thereof.

FIG. 59 is a diagram showing joining of unit sections.

FIG. 60 is a diagram illustrating a method of connecting a unit section to n (> 2) stages.

FIG. 61 is a diagram illustrating a configuration method of the ladder-type circuit according to the present embodiment.

FIG. 62 is a circuit diagram of a conventional surface acoustic wave filter.

FIG. 63 is a circuit diagram of a thirteenth embodiment of a surface acoustic wave filter according to the present invention.

FIG. 64 is a circuit diagram of a fourteenth embodiment of a surface acoustic wave filter according to the present invention.

FIG. 65 is a circuit diagram of a surface acoustic wave filter according to a fifteenth embodiment of the present invention.

FIG. 66 is a diagram showing a configuration embodying the circuit of FIG. 65;

FIG. 67 is a view showing characteristics of the filter of FIG. 66.

FIG. 68 is a diagram showing a ladder-type filter circuit in which surface acoustic wave resonators having different resonance frequencies are arranged in a parallel arm and a series arm.

FIG. 69 is a diagram correspondingly showing a frequency characteristic of admittance (Y _p ) of a parallel arm resonator and a frequency characteristic of impedance (Z _s ) of a series arm resonator.

FIG. 70 is a diagram showing an example of a conventional surface acoustic wave filter.

FIG. 71 is a diagram showing the pass characteristics of the filter of FIG. 70;

[Explanation of symbols]

60, 90, 100, 110, 120, 130, 24
0, 250, 260, 270, 280 Surface acoustic wave filter 80, 150 Surface acoustic wave filter device 81 Ceramic package 82 Filter chip 83 Lid 84 _{-1 to} 85 _-6 Electrode terminal 85 _{-1 to} 85 _-5 Terminal 86 _-1 86 _-5 bonding wires 125 attenuation pole 127 stopband 131,201,211 excitation electrode 132,133,160,161,166,167,2
02, 203, 212, 213, 242 Reflector 220, 221, 230, 231 Microstrip line 241 36 ° Y-cut X-propagation LiTaO ₃ substrate (chip) Rs ₁ , Rs ₂ Series arm resonator Rp _{1 to} Rp ₃ Parallel arm Resonator

 ──────────────────────────────────────────────────続 き Continuing on the front page (72) Inventor Tsutomu Miyashita 1015 Uedanaka, Nakahara-ku, Kawasaki City, Kanagawa Prefecture Inside Fujitsu Limited (72) Inventor Takashi Matsuda 1015 Kamikodanaka, Nakahara-ku, Kawasaki City, Kanagawa Prefecture Fujitsu Limited ( 72) Inventor Mitsuo Takamatsu 1015 Uedanaka, Nakahara-ku, Kawasaki City, Kanagawa Prefecture Inside Fujitsu Limited

Claims (2)

[Claims] 1. A first terminal having a predetermined resonance frequency and a pair of surface acoustic wave resonators connected in parallel to a second terminal having a resonance frequency substantially equal to the anti-resonance frequency of the first resonator. In a ladder-type surface acoustic wave filter in which surface acoustic wave resonators are arranged in series, a plurality of second one-port surface acoustic wave resonators (R ₂ , R ₂ ) are provided in the series arm (61). SAW filter, characterized in that the made thereto and arranged to connect in series by adding inductance (L _S) in series configuration. 2. A first terminal having a predetermined resonance frequency paired with a surface acoustic wave resonator and a second terminal having a resonance frequency substantially equal to the anti-resonance frequency of the first resonator. In a ladder-type surface acoustic wave filter in which a pair of surface acoustic wave resonators are arranged in a series arm, a second one-port pair surface acoustic wave resonator (R ₂ , R ₂ ) is provided in the series arm (61). A plurality of the surface acoustic wave resonators are connected in series and added with an inductance (L ₁ ) in series, and the first surface acoustic wave resonator is connected to a central excitation electrode (131) and reflectors (132, 132) on both sides thereof. 133), and the distance between the center of the reflector and the excitation electrode is d = (n + β) · λ (where n is an integer, β is a real number of 1 or less, and λ corresponds to the resonance frequency. (Period of the comb-shaped electrode), β is substantially 0.4. Surface acoustic wave filter.