JP3255899B2 - Surface acoustic wave filter - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a surface acoustic wave filter, and more particularly to a ladder type surface acoustic wave filter applicable to an RF (high frequency section) filter of a small mobile radio device such as a mobile phone and a mobile phone. .

[0002] One of the specifications of current automobiles and mobile phones in Japan
Example is ± 8.5MHz centered on 933.5MHz
Is the transmission band. In terms of fractional bandwidth, it is about 2%.

The surface acoustic wave filter must have characteristics satisfying the above-mentioned specifications. Specifically, the pass band width is as wide as 2% or more as a fractional bandwidth, and the loss is 1.5. ~ 2dB or less, 20dB suppression
It is required to be as high as ３０30 dB or more.

In order to satisfy this requirement, a surface acoustic wave filter using a surface acoustic wave element as a resonator instead of the conventional transversal type and having a ladder-type resonator has been desired. .

[0005]

2. Description of the Related Art FIG. 70 shows an equivalent circuit of a surface acoustic wave filter 1 described in JP-A-52-19044. This filter 1 includes a surface acoustic wave resonator 3
And the surface acoustic wave resonator 5 is disposed on the parallel arm 4, and the equivalent parallel capacitance C _OB of the resonator 5 of the parallel arm 4 is larger than the equivalent parallel capacitance C _OA of the resonator 3 of the series arm 2. Configuration.
This filter 1 has the characteristic shown by line 6 in FIG.

[0006]

In the above filter 1, if the equivalent parallel capacitance C _OB is increased as described later,
As shown by arrow 7, the degree of suppression can be increased. However, when the capacitance C _OB is increased, the pass bandwidth becomes narrow as shown by the arrow 8 and the loss increases as shown by the arrow 9, and the characteristic becomes as shown by the line 10.

If the suppression degree is to be 20 dB or more,
The pass bandwidth is less than 1% in fractional bandwidth,
The above-mentioned specification of the mobile phone cannot be satisfied.

SUMMARY OF THE INVENTION It is an object of the present invention to provide a surface acoustic wave filter capable of simultaneously increasing the pass band width and improving the degree of suppression outside the pass band.

[0009]

According to a first aspect of the present invention, a plurality of parallel arms and a series arm constituted by a surface acoustic wave resonator are directly connected in a ladder form to form a single piezoelectric substrate. is formed, have a band-pass characteristic, the respective surface acoustic waves both
The vibrators each have a non-overlapping propagation path and have a predetermined resonance.
The first one-port surface acoustic wave having a frequency (frp)
A vibrator is provided on each of the parallel arms, and an anti-resonance frequency of the first resonator is provided.
(Fap) or substantially higher resonance frequency
The second one-port surface acoustic wave resonator having the number (frs)
Connected to each of the series arms, and the second terminal
The electric resistance (rs) of the comb-shaped electrode forming the vibrator is
The electric potential of a comb-shaped electrode forming one terminal pair surface acoustic wave resonator
The second end is configured to be smaller than the air resistance (rp).
Resistance of Comb-shaped Electrode Forming Element to SAW Resonator
(Rs) constitutes the first one-port surface acoustic wave resonator.
Of reducing the electric resistance (rp) of the comb-shaped electrode
The stage has a comb shape that constitutes the second one-terminal surface acoustic wave resonator.
The opening length (ls) of the electrode is equal to the first terminal-to-surface acoustic wave.
Shorter than the opening length (lp) of the comb-shaped electrode constituting the resonator
And forming the second one-port surface acoustic wave resonator.
The number of comb electrodes (Ns) is the first one terminal pair elasticity table.
More than the logarithm (Np) of the comb-shaped electrodes constituting the surface wave resonator
Surface acoustic wave filter characterized by a comb-like configuration
It is. With this configuration, the pass band in the case of multistage
Can be reduced.

[0010]

[0011]

[0012]

According to the second aspect of the present invention, the electric resistance (rs) of the comb-shaped electrode forming the second one-terminal-pair surface acoustic wave resonator according to the first aspect is determined by the first one-terminal pair elasticity. Means for reducing the electrical resistance (rp) of the comb-shaped electrode constituting the surface acoustic wave resonator is determined by changing the thickness of the comb-shaped electrode made of a metal thin film constituting the first one-port surface acoustic wave resonator. The thickness of the second terminal-pair surface acoustic wave resonator is made thinner than that of the same metal. With this configuration, it is possible to reduce the insertion loss in the pass band in the case where the number of stages is increased.

[0014]

DESCRIPTION OF THE PREFERRED EMBODIMENTS First, one of the basic principles of the present invention will be described. FIG. 1 shows the principle configuration of a surface acoustic wave filter 20 that can be used in the present invention. Reference numeral 21 denotes a first one-terminal surface acoustic wave resonator having a predetermined resonance frequency frp.
And arranged on the parallel arm 22. Reference numeral 23 denotes a second one-terminal surface acoustic wave resonator having a resonance frequency frp substantially equal to the anti-resonance frequency fap of the first resonator 21.
It is arranged at 4. Reference numeral 25 denotes an inductance, which is added in series to the first resonator 21 and disposed on the parallel arm 22.

The principle that a circuit having a one-port pair surface acoustic wave resonator in a series arm and a parallel arm has filter characteristics is as follows. This principle is necessary for the explanation of the principle of the present invention, and will be described in detail here.

In order to evaluate whether or not the resonance circuit exhibits a filter characteristic, a method using image parameters is easy to understand. This method is based on the theory and design of filters by Yanagisawa et al.
(Sanpo Publishing: Electronics Selection, published in 1974)
Is described in detail.

The principle will be described below based on this.

FIG. 2 shows a basic ladder circuit showing the filter characteristics. In FIG. 3, hatched black boxes are surface acoustic wave resonators 30 and 31.

For the sake of simplicity, the surface acoustic wave resonator is assumed to be a reactance circuit having no resistance, the impedance of the series arm resonator 30 is Z = jx, and the impedance of the parallel arm resonator 31 is Z = jx. Let admittance be Y = jb.

According to the image parameter method, if the input side voltage and current are V ₁ and I ₁ and the output side is V ₂ and I ₂ (see FIG. 2),

[0021]

(Equation 1) The image transmission amount γ (complex number) defined by is important. That is,

[0022]

(Equation 2) If the value represented by this equation is an imaginary number in FIG.
The whole two-port pair circuit shows a pass characteristic, and a real number shows an attenuation characteristic. Here, the symbols A, B, C, and D are four-terminal constants when the entire circuit of FIG. 2 is represented by an F matrix, and when expressed by the above-described x and b, they are as follows.

A = 1 B = xx C = jb D = 1−bx (3) Therefore, the equation (2) becomes the following equation.

[0024]

(Equation 3) From equation (4), when 0 <bx <1, that is, when b and x have the same sign and small values, the entire circuit of FIG. 2 shows a pass characteristic, and bx <
0 or bx> 1, that is, b and x are different signs or bx
It can be seen that when the product is a large value, it exhibits an attenuation characteristic.

Here, in order to further qualitatively know the frequency characteristics of b and x, the frequency characteristics of the impedance and admittance of the surface acoustic wave resonator are examined.

The one-port pair surface acoustic wave resonator comprises a comb-shaped electrode 40 as shown in FIG. 3A (see P.76-Nikkei Electronics November 29, 1976).
P. 98).

Reference numeral 41 denotes an electrode pair, reference numeral 42 denotes an opening length (crossing width), and reference numeral 43 denotes a comb-shaped electrode period. This comb-shaped electrode is generally represented by an equivalent circuit 45 as shown in FIG. Here, C _O is the capacitance of the comb-shaped electrode, and C ₁ and L ₁ are equivalent constants.

This equivalent circuit 45 is hereinafter represented by a symbol 46 shown in FIG.

FIGS. 4A and 4B show the comb-shaped electrodes, respectively, in FIG.
(B) Qualitatively shows the frequency dependence of impedance and admittance when represented by an equivalent circuit.

The characteristics shown in FIG.
A double resonance characteristic having two resonance frequencies fr and fa is obtained.
Here, fr is called a resonance frequency, and fa is called an anti-resonance frequency.
Resonators having such a double resonance characteristic are arranged in a series arm and a parallel arm, respectively, and the anti-resonance frequency fa of the parallel arm is further increased.
When p is made substantially equal to the resonance frequency frs of the series arm, a circuit exhibiting a band-pass filter characteristic having the center frequency as the center frequency can be configured. The reason is that fap ≒ fr as shown in the diagram of the frequency characteristic of the immittance in FIG.
In the vicinity of the center frequency which is s, 0 <bx <1 is satisfied, and the passband is obtained from the above-mentioned condition. In a frequency region slightly away from the center frequency, bx> 1, and in a region far away from the center frequency, bx> 1
This is because it becomes <0 and both fall in the attenuation range.

Therefore, the surface acoustic wave filter 1 having the structure shown in FIG. 1 qualitatively has a filter characteristic indicated by a middle line 47 in FIG. 5B.

[Passing Band Determining Factors] Next, the bandwidth determining factors in such a resonator type surface acoustic wave filter will be considered.

As can be seen from FIG. 5, the bandwidth is mainly determined by the difference between the resonance frequency fr and the anti-resonance frequency fa of each resonator. If the difference is large, the bandwidth is wide and wide, and if the difference is small, the bandwidth is narrow. Here, fr and fa can be determined from the following equations using the equivalent circuit constants in FIG.

[0034]

(Equation 4) Since the fractional bandwidth (Δf / f ₀ ) is mainly determined from the difference between fr and fa, it is expressed by the following equation using equations (6) and (7).

Δf / f ₀ = 2 (fa−fr) / (fa + fr) ≒ 2 / (4γ + 1) (8) As is apparent from the above equation, γ (capacity ratio) is an important factor that determines the relative bandwidth. Become. However, this value is disclosed in JP-A-52-1.
As described in Japanese Patent Application Laid-Open No. 9044, it is almost determined by the type of the substrate material forming the comb-shaped electrode. For example, in an ST-cut quartz having a small electromechanical coupling coefficient of a material, γ is 1300 or more, whereas in a 36 ° Ycut-x propagating LiTaO ₃ having a large electromechanical coupling coefficient,
γ takes the value of the 15th place. The fractional bandwidth is expressed by ST
0.04% in cut quartz, 36 ° Ycut-X propagation L
In iTaO ₃ , it is 3.3%. Therefore, if the substrate material is determined, the bandwidth is almost determined.

If the equivalent parallel capacitance C _OB is increased as disclosed in Japanese Patent Application Laid-Open No. 52-19044 in order to increase the degree of out-of-band suppression, the bandwidth becomes narrower.

This will be described in detail with reference to FIG. As is clear from the above explanation of the principle, fr and f of the parallel resonator
When the admittance value is increased while a is fixed (in order to increase the admittance value, the capacitance C ₀ is increased by increasing the opening length or logarithm of the comb-shaped electrode while keeping γ constant). As shown in FIG. 6A, b
Since the x product increases negatively, the amount of attenuation increases and the characteristics are improved, but the bx product increases positively near the center frequency, so that b
The region where x> 1 is expanded, and as a result, the passband where 0 <bx <1 is narrowed, and a sufficient band cannot be obtained. This situation is represented by an arrow in FIG.

[Improvement of Pass Bandwidth] As one means for solving the above points, the difference between fr and fa of at least one of the series arm resonator and the parallel arm resonator is increased. It is necessary to satisfy two conditions of increasing the impedance value or the admittance value. The reason for increasing the impedance value and the admittance value is to increase the out-of-band attenuation. If this can be realized, it is possible to improve the out-of-band attenuation without widening or narrowing the pass band.

First, the conditions fr and fa of the resonator are satisfied.
As a method for widening the difference, a method of adding an inductance L in series with a one-port surface acoustic wave resonator is effective. FIGS. 7A and 7B show changes in impedance and admittance frequency when 8 nH is connected as L in series with a one-port surface acoustic wave resonator. Each constant of the equivalent circuit of the surface acoustic wave resonator used for the calculation is shown in FIG.

In FIG. 7A, a line 50 indicates the impedance characteristic before L is added. Line 51 shows the impedance characteristic after L is added.

In FIG. 7B, a line 52 indicates the admittance characteristics before L is added. The line 53 shows the admittance characteristic after adding L.

From FIG. 7A, it can be seen that the addition of L widens the interval between fr and fa. In this case, the frequency was increased by about 30 MHz. The reason for this is that, as is clear from the frequency characteristic of the impedance shown in FIG. 7A, the impedance of only the original resonator is raised to the + side by ωL by adding L in series, and as a result, f
This is because r has changed to fr '. At this time, fa hardly moves. The admittance, which is the reciprocal of the impedance, also changes for the same reason as shown in FIG. Also in this case, it can be clearly seen that fr has changed to fr '.

Under the following conditions, the admittance value is increased by adding L, as is clear from FIG. 7B. However, the impedance values are
As shown in (A), it is smaller outside the band.
Therefore, when this method is applied to a resonance circuit of a series arm, a method of increasing the impedance value is further required. This can be solved by connecting a plurality of the same surface acoustic wave resonators in series.

In FIG. 8, a line 55 indicates the impedance characteristic of one resonator. A line 56 indicates the impedance characteristic of the resonance part when n resonators are connected in series.

As shown in FIG. 8, by connecting n resonators, the impedance value of the resonator section becomes n times. On the other hand, regarding the difference between fa and fr, the spread of the resonance frequency when L is connected is fr ″,
Although slightly narrower than r ', f is higher than when L is not connected.
The difference between a and fr can be large. If necessary, the difference between fa and fr is further increased by increasing the value of L.

As a second means for expanding the pass band width, as shown in FIG. 44, the anti-resonance frequency f
Instead of making the ap and the series arm resonance frequency frs substantially coincide with each other, a method of setting frs> fap can be considered.

However, when frs> fap, FIG.
As shown in FIG. 7, bx <0 near the center frequency, the passband condition described above is not satisfied, and there is a danger that loss and ripple increase.

However, assuming that frs-fap = Δf, Δf
By controlling the magnitude of, it is possible to substantially increase the loss and increase the ripple and realize the expansion of the pass band. Further, by increasing Δf, it is possible to simultaneously improve the out-of-band suppression degree. See Example 11 for details.
It will be described later.

Hereinafter, the contents of the present invention will be described with reference to specific examples. Most of the examples were performed by simulation. Therefore, first, the simulation used in the present invention will be briefly described, and a comparison with an experiment will be shown in order to prove the validity of the simulation.

The equivalent circuit shown in FIG. 3B can simply simulate the characteristics of a one-port surface acoustic wave resonator. However, the equivalent circuit shown in FIG. In addition, it is difficult to accurately simulate the effect of the reflector. Therefore, based on the Smith equivalent circuit that the inventors have already developed, a method of expressing this in a transfer matrix was used and applied to a resonator (O. Ikata et al .: 1).
990 ULTRASONIC SYMPOSIUM Proceedings, vol.1, pp83-8
6, (1990), which is referred to as reference (1). ).

FIG. 9A shows the result of a simulation when a one-port surface acoustic wave resonator is provided in a parallel arm.

FIG. 9B shows that the material of the parallel arms is Al-2.
%, A one-port surface acoustic wave resonator comprising a comb-shaped electrode having a thickness of 1600 ° is provided, and the resonator has a length of 3 mm.
Shows the results of experiments when the bonding wire (L = 1.5 nH) is connected.

9 (A) and 9 (B), the movement of the resonance point (indicated by fr ₁ , fr ₂ , fr ₃ in the figure) due to the change of the aperture length and the attenuation near the resonance point are shown. It turns out that the experimental value and the calculated value agree well.

FIG. 10A shows the result of a simulation when a resonator is provided in the series arm. Since the bonding pad used in the experiment described later was rather large, in the simulation, the stray capacitance was 0.5 pF
Is considered.

FIG. 10B shows the result of an experiment when a resonator is connected to the series arm.

Comparison between FIGS. 10A and 10B shows that the anti-resonance frequencies fa ₁ , fa ₂ , and fa ₃ do not depend on the aperture length and that the attenuation changes near the anti-resonance frequency. It can be seen that they match well.

Therefore, it is clear that the filter characteristics when these are combined well agree with the experiment, and the following examples were simulated.

FIG. 11 shows a surface acoustic wave filter 60 according to a first embodiment of the present invention.

At present, one example of the specifications of automobiles and mobile phones in Japan is that, with a center frequency of 933.5 MHz, the range of ± 8.5 MHz is the transmission band of mobile equipment, and is -55 MHz away therefrom. There is a specification that a range of ± 8.5 MHz with a center frequency of 878.5 MHz is a reception band.

This embodiment is designed so as to be suitable for the transmitting filter of the mobile device. The same applies to other embodiments described later.

The series arm 61 has a one-terminal-pair surface acoustic wave resonator R
₂ and R ₄ are arranged.

One-port pair surface acoustic wave resonators R ₁ , R ₃ and R ₅ are arranged on the parallel arms 62, 63 and 64, respectively. L ₁ ,
L ₂ and L ₃ are inductances, and are resonators R ₁ and L ₁ , respectively.
R ₃ and R ₅ are connected to the parallel arms 62, 63 and 64. Each of the resonators R _{1 to} R ₅ has a comb-shaped electrode structure shown in FIG. The logarithm is 100 and the opening length is 80 μm. The material is Al-2% Cu, and the film thickness is 3,000 angstroms.

Further, the period of the comb-shaped electrode is appropriately determined,
The resonance frequency of each resonator R ₁ , R ₃ , R _{5 in} the parallel arms 62, 63, 64 is 912 MHz, and the anti-resonance frequency is 934 M
Hz. The resonance frequency of each of the resonators R ₂ and R _{4 in} the series arm 61 is 934 MHz, and the anti-resonance frequency is 962 MHz.
There is. The inductances L ₁ , L ₂ and L ₃ are all 4n
H.

The surface acoustic wave filter 60 having the above configuration has a pass characteristic indicated by a line 65 in FIG. When the inductance L is 2 nH and 6 nH, the pass characteristics of the filter 60 in FIG. 11 are as shown by lines 66 and 67 in FIG. 12, respectively.

Based on FIG. 12, L with respect to the pass band width
Expressing the dependency is as shown by a line 70 in FIG. Here, the frequency width at the level of the amount of attenuation that is -3 dB lower than the minimum insertion loss was defined as the pass bandwidth.

Similarly, based on FIG. 12, the L dependence on the degree of normal out-of-band suppression is represented by the line 7 in FIG.
As shown in FIG.

As can be seen from FIG. 12, if L is too large, a sufficient suppression region on the 55 MHz lower frequency side from the center frequency cannot be obtained. Therefore, L is set to 4 nH as described above. The value of L is appropriately selected according to the specifications of the filter.

The pass characteristic of the filter 1 having the conventional configuration shown in FIG. 70 is as shown by a line 68 in FIG.

In FIG. 12, the pass characteristic (line 65) of the filter 60 of this embodiment is changed to the pass characteristic (line 68) of the conventional filter.
In comparison with the filter 60 of the present embodiment, the pass band width is wider as shown by an arrow 75 and the degree of suppression outside the pass band is higher as shown by an arrow 76, as compared with the conventional filter.
In addition, the loss is low as indicated by the arrow 77.

FIGS. 14 and 15 show a surface acoustic wave filter device 80 which realizes the surface acoustic wave filter 60 of FIG.

Reference numeral 81 denotes a ceramic package, 82 denotes a filter chip, and 83 denotes a lid that functions as a ground. The ceramic package 81 is made of alumina ceramic and has a size of 5.5 × 4 mm ² and a small height of 1.5 mm. The ceramic package 81 has Au electrode terminals 84 _{-1 to} 84 _-6 formed thereon. Filter chip 8
2 is made of LiTaO ₃ and has a size of 2 × 1.5 mm
^{2 has} a thickness of 0.5 mm.

On the surface of this filter chip 82, resonators R ₁ to R having a comb-shaped electrode structure having a logarithm of 100, an opening length of 80 μm, a material of Al-2% Cu, and a film thickness of 3,000 ° are provided.
R ₅ are staggered so as not to share the surface acoustic wave propagation path with each other.

On the surface of the filter chip 82, two signal line terminals 85 _-1 and 85-1 as bonding terminals are provided.
_85-2 and three grounding terminals _85-3 , _85-4 , _85-5
Is formed. Reference numerals 86 _{-1 to} 86 _-5 denote bonding wires, made of Al or Au, having a diameter of 25 μmφ, and connected by bonding to the terminals 84 _{-1 to} 84 _-5 and the terminals 85 _{-1 to} 85 _-5 , respectively. I have. Of these, wire 8
6 _-1 and 86 _-2 are a part 61a of the serial arm 61 in FIG.
And 61b. Wire 86 _-3 Yes connected between the ground electrode terminals 84 _-3 and 85 _-3, the wire 86 _-4
Is Yes connected between the other ground electrode terminal 84 _-4 and 85 _-4, the wire 86 _-5 is coupled between the other ground electrode terminals 84 _-5 85 _-5. This wire 86 _{-3 to} 86
_-5 is as long as 2.0 mm.

As described above, a thin and long wire has an inductance according to the theory of high frequency.

According to the theoretical formula of the aerial ribbon inductor (Kuraishi: Science and Engineering Course, “Example Circular Microwave Circuit”, described in P199 of Tokyo Denki University Press), the wire 8
The inductance of 6 _-3 , 86 _-4 , and 86 _-5 is about 1 nH.

This is insufficient to obtain an inductance of 4 nH, and a ceramic package and L on the filter chip as shown in FIGS. 40 and 41 described later were used.

Thus, the inductances L ₁ , L ₂ and L ₃ in FIG. 11 are formed.

Embodiment 2 FIG. 16 shows a surface acoustic wave filter 90 according to a second embodiment of the present invention.

In the figure, parts corresponding to those shown in FIG. 11 are denoted by the same reference numerals.

The opening length A _S of the resonator R ₂ in the series arm 61 is
80 μm.

[0081] The parallel arm 62, are disposed a resonator R1A and inductance L ₁ is connected in series.

[0082] resonator R1A is an aperture length A _P is 120 [mu] m. Aperture length A _P is longer than the aperture length A _S, which is 1.5 times the aperture length A _S. Note that the logarithm N _{P of} the resonators R ₂ and R 1A,
N _S is 100 both equal.

This filter 90 has a pass characteristic indicated by line 91 in FIG.

Comparing this pass characteristic with the pass characteristic of the filter 60 of FIG. 11 shown by the line 65, it can be seen that, as shown by the arrow 92, the degree of suppression outside the pass band is improved without changing the pass band width. I understand.

FIG. 18 shows the aperture length dependence of the pass characteristic of the filter having the configuration of FIG.

FIG. 22A shows the opening length (A) of each series arm resonator when L is added as shown in FIG. 16 and when L is not added as shown in FIG.
The ratio A _P / A _S of the aperture length of the parallel arm resonator for _S) (A _P)
And the relationship between the degree of suppression and the out-of-band suppression degree. Out-of-band suppression is 4n
If an inductance L of H is added, line 9
2 and when the inductance L is not added, it becomes as shown by a line 93.

FIG. 18B shows the relationship between A _P / A _S and the pass bandwidth. The pass bandwidth is as shown by the line 95 when the inductance L of 4 nH is added, and as shown by the line 96 when the inductance L is not added.

The following can be understood from FIGS. 18 (A) and 18 (B).

The opening length A _P of the resonator R1A in the parallel arm 62
The by longer than the aperture length A _S of the resonator R ₂ in the series arm 61, out-of-band suppression degree is increased.

[0090] By adding the inductance L ₁ on the parallel arm 62, as compared with the case having no inductance, the effect of increasing the aperture length A _P of the resonator R1A is large and almost no degradation of bandwidth.

From the above, it can be seen that the filter 90 of the above embodiment has a pass characteristic in which the degree of suppression outside the pass band is increased without reducing the pass band width at all, as compared with the filter 60 of FIG. I understand.

[Embodiment 3] FIG. 19 shows a surface acoustic wave filter 100 according to a third embodiment of the present invention.

In the figure, parts corresponding to those shown in FIGS. 11 and 16 are denoted by the same reference numerals.

The logarithm N _S of the resonator R ₂ of the series arm 61 is 100
It is. The parallel arm 62, are disposed a resonator R1B and the inductance L ₁ is connected in series. The resonator R1B is
The logarithm N _P is 150, which is more than the logarithm N _{S of} the above-described resonator R ₂ , and is 1.5 times the logarithm N _S. The aperture lengths A _S and A _P of the resonators R ₂ and R 1A are both 80 μm, which are equal.

This filter 100 corresponds to the line 10 in FIG.
It has a pass characteristic indicated by 1. Comparing this pass characteristic with the pass characteristic of the filter 60 of FIG. 11 shown by line 65, without narrowing the pass bandwidth, as shown by the arrow 102,
It can be seen that the degree of suppression outside the passband is improved.

In addition, as compared with the pass characteristic of the filter 90 shown in FIG.

Therefore, the filter 100 according to the present embodiment increases the degree of suppression outside the pass band without reducing the pass band width as compared with the filter 11 of FIG.
It has a transmission characteristic with less loss deterioration compared to 0.

Embodiment 4 FIG. 21 shows a surface acoustic wave filter 110 according to a fourth embodiment of the present invention. In this embodiment,
It is obtained by improving the bandpass characteristic by enlarging the difference between the anti-resonance frequency f _a of the resonance circuit of the series arm and the resonance frequency f _r.

In the figure, parts corresponding to those shown in FIG. 11 are denoted by the same reference numerals. Of the serial arms 61, the parallel arms 6
The same resonator R ₂ in the portion between 2,63 are connected to the two series, it is further added inductance L _S of 3nH in series thereto. Similarly, of the serial arms 61, the parallel arms 6
3, 64, the same resonator R ₄ is connected in series, and an inductance L of 3 nH is connected in series with the same resonator R _4.
S is added. The parallel arm 62 has only one resonator R position. Also, the parallel arms 63, only one resonator R ₃ is are arranged. Similarly, the parallel arm 64 has
Only one resonator R ₄ is are arranged.

This filter 110 corresponds to line 11 in FIG.
It has a pass characteristic indicated by 1.

Here, the effect of adding the inductance L _S and the resonators R ₂ and R ₄ will be described.

The pass characteristic when the inductance L _S and the resonators R ₂ and R ₄ are eliminated from the filter 110 of FIG. 21 is as shown by a line 68 (see FIG. 12). With the addition of the inductance L _S , the pass bandwidth was increased as indicated by the arrow 112 and the degree of out-of-band suppression was increased as indicated by the arrow 113. In particular, with regard to the pass bandwidth, the expansion toward the high frequency side was particularly large, and the bandwidth of about 15 MHz was expanded toward the high frequency side. The pass characteristic is represented by the line 114
It became as shown by. In this state, the degree of out-of-band suppression is not sufficient. Therefore, one resonator R ₂ and R ₄ was added.

By adding this one resonator R ₂ , R ₄ , as shown by the arrow 115, without narrowing the pass band width,
The degree of out-of-band suppression was improved by about 5 dB, and the transmission characteristic indicated by line 111 was obtained. Comparing line 111 with line 68, the arrow 1
As shown at 16, the loss is also improved compared to line 68.

The number of resonators R ₂ and R _{4 in the} series arm 61 may be three or more.

Further, as shown by a two-dot chain line in FIG. 21, an inductance may be inserted into the parallel arms 62 to 64.

[Fifth Embodiment] FIG. 21 shows a surface acoustic wave filter 120 according to a fifth embodiment of the present invention.

In the figure, parts that are the same as the parts shown in FIG. 11 are given the same reference numerals, and descriptions thereof will be omitted.

The inductance value of the inductance L ₁ of the parallel arm 62 is 4 nH. Inductance value of the inductance L ₂ of another parallel arm 63 is 5.5 nH. Furthermore inductance value of the inductance L ₃ of another parallel arm 64 is 7 nH.

As described above, by making the inductance values of the inductances L _{1 to} L ₃ of the parallel arms 62 to 64 different, the filter 120 has a pass characteristic indicated by a line 121 in FIG.

Here, the filter 60 of FIG. 11 in which the inductance values of the inductances L _{1 to} L ₃ are all equal to 4 nH.
Let's compare with the passing characteristics of

This filter 60 has a pass characteristic (see FIG. 12) indicated by a line 65 in FIG.

The pass characteristic of the filter 120 of this embodiment is as follows.
Compared with the pass characteristic of the filter 60, the degree of suppression outside the pass band is increased as shown by the arrow 122 without reducing the pass band width at all. Looking at the low-frequency side of the pass band, the filter 60 has only one attenuation pole 123 near 902 MHz, whereas the filter 60 has 875 MHz.
Attenuation poles 124 and 125 are generated at two places of z and 892 MHz. As a result, the two attenuation poles 124, 125
Is a stop band 127.

[Embodiment 6] FIG. 25 shows a surface acoustic wave filter 130 according to a sixth embodiment of the present invention. In this embodiment,
This is to reduce the loss.

In the figure, parts corresponding to those shown in FIG. 11 are denoted by the same reference numerals, and description thereof is omitted.

The first surface acoustic wave resonator R1 of the parallel arm 62
B has a configuration in which an excitation electrode 131 and reflectors 132 and 133 are arranged on both sides of the excitation electrode 131 as shown in FIG. Reflector 1
32 and 133 are excitation electrodes 131 and reflectors 132 and 133
(Where n is an appropriate integer, β is a real number equal to or less than 1, and λ is the period of the comb-shaped electrode corresponding to the resonance frequency). At this time, it is arranged at the position where β = 0.4.

The logarithm of the reflectors 132 and 133 is 50
It is. The resonator R1B provided with a reflector is represented by a symbol added with "*" as shown in FIG. The other parallel arm 63,
The 64 resonators R3B and R5B are also provided with a reflector similarly to the above-described resonator R1B.

The filter 130 having the above configuration has a pass characteristic indicated by a line 134 in FIG. This passing characteristic is shown in FIG.
Compared with the pass characteristic of the first filter 60 (shown by the line 65), the insertion loss in the normal band is reduced as shown by the arrow 135.

Here, as shown in FIG. 26, the ripple r _P is applied to the reflectors 132 and 1 on both sides of the excitation electrode 131 of the parallel arm.
This is caused by disposing 33.

Here, the reason why the arrangement positions of the reflectors 132 and 133 are determined as described above will be described.

In the above equation, when β is changed from 0 to 0.5, the influence on the width of the ripple r _P is shown in FIG.
As shown at 40. In the figure, a point 141 is a point where the ripple width is minimum, and β at this time is 0.4. For this reason, β is set to 0.4.

FIG. 29 shows a surface acoustic wave filter device 150 which realizes the filter 130 of FIG. In the figure, FIG.
The same reference numerals are given to the portions corresponding to the components shown in FIG.
The description is omitted. 132,133,151,15
2, 153 and 154 are reflectors, respectively.

Next, a modification of the first one-port surface acoustic wave resonator will be described.

FIG. 30 shows a modification. This resonator R
₁ Ba is on both sides of the excitation electrode 131, as a reflector, a configuration in which the electrical load is arranged comb-shaped electrodes 160, 161 of the short-circuited.

FIG. 31 shows another modification. The resonator R ₁ B _b is as a reflector on both sides of the excitation electrode 131, a configuration in which arranged strip array type electrodes 165 and 166.

[Embodiment 7] FIG. 32 shows a surface acoustic wave filter 170 according to a seventh embodiment of the present invention. In this embodiment,
As in the sixth embodiment, the loss is reduced. In the figure, parts corresponding to those shown in FIG. 21 are denoted by the same reference numerals, and description thereof is omitted.

The filter 170 is the filter 11 shown in FIG.
26, the first surface acoustic wave resonators R1B, R3B, R5B of the respective parallel arms 62, 63, 64 are reflected at positions where β on both sides of the excitation electrode 131 is determined to be 0.4 as shown in FIG. In this configuration, devices 132 and 133 are provided. This filter 17
According to 0, a pass characteristic in which the loss in the pass band is smaller and the ripple is suppressed is obtained as compared with the characteristic indicated by the line 111 in FIG.

[Embodiment 8] This embodiment aims at eliminating the ripple r _P in FIG.

First, a description will be given of a means for effectively removing a ripple appearing when the reflector is added.

The inventors examined the relationship between the frequency position where ripples appeared and the electrode thickness by simulation. In the simulation, the effect of increasing the film thickness was replaced by increasing the ratio between the acoustic impedance under the electrode (Zm) and the acoustic impedance at the free surface (Zo). As described in the literature (1), the increase in the electrode film thickness means the increase in the mass, which is considered to be directly proportional to the increase in the discontinuity of the acoustic impedance. / Zm = Vo / Vm = 1 + k 2/2 + α (t) ... (9) (Vo, Vm: the speed of sound under the free surface and the electrode, k ^2: electromechanical coupling coefficient), and the thickness of the alpha (t) t This was varied as a parameter proportional to

With this arrangement, the center frequency fo of the filter becomes fo = 2fo '/ (1 + Q) (10). As the film thickness increases, the center frequency fo' becomes lower from the center frequency fo 'when there is no discontinuity in acoustic impedance. This is consistent with the well-known experimental fact of moving to the side.
As a result of the simulation, when α (t) is increased, that is, as the electrode film thickness is increased, the frequency position where the ripple r _P appears moves to the high frequency side of the pass band as shown by an arrow 180 in FIG. Eventually, it was found that it eventually fell into the attenuation pole on the high frequency side. This is schematically shown in FIG.

In FIG. 33, another ripple r _S is generated due to the reflector of the series arm resonator.

FIG. 34 shows the transmission characteristic when α (t) = 0.08 and the ripple generated from the reflector of the resonator of the parallel arm falls just inside the attenuation pole on the high frequency side. Therefore, in the figure, the ripple disappears from the pass band, and the insertion loss is considerably reduced. In this figure, since the center of the pass band has shifted to the lower frequency side according to the equation (10), the resonance frequency of the resonators of the series arm and the parallel arm is adjusted so that the center frequency becomes 932 MHz in order to correct this. 1
The frequency is shifted to the high frequency side by 5 MHz.

To see this in correspondence with the actual film thickness, a chip was prototyped and its passing characteristics were examined.

Line 18 in FIGS. 35 (A), (B) and (C)
5,186,187 have film thicknesses of 2,000 and 300, respectively.
The passing characteristics at 0 ° and 4000 ° are shown correspondingly.

Although the center frequency changes by changing the film thickness, the data shown in the figure are adjusted to correct this by changing the period of the comb-shaped electrode so that the center frequency does not change much.

As is clear from FIG. 35, the ripple r _P of the resonator of the parallel arm and the ripple r _S of the serial arm out of the band that appeared in the band at 2000 ° are 3000 °.
In the case of, it moves to the high frequency side and becomes r _P ', r _S ',
As a result of r _P ′ being buried in the attenuation pole on the high frequency side, good characteristics were obtained with no ripple in the band. This result agrees qualitatively with the simulation result.

However, loss degradation due to bulk waves that cannot be calculated by simulation when the film thickness is increased (Ebata et al .: “Surface acoustic wave resonator on LiTaO ₃ substrate and its application to VTR oscillator”, IEICE Transactions, Magazine, vol.
J66-C, No.1, pp23-30, 1988) and improvement in loss due to resistance loss, and the balance is also an important factor.

FIG. 36A plots the change in the minimum insertion loss when the film thickness is changed.

In the figure, a line 190 indicates a loss due to a bulk wave,
Line 191 shows the loss due to ohmic loss. Line 192 is the experimental value. As can be seen from the figure, the insertion loss is approximately 2500 ° and the effects of both are balanced. From about 3500 °, the loss increase due to the bulk wave becomes dominant and starts to deteriorate.

The line 193 in FIG. 36 (B) shows the frequency position of the ripple r _P with respect to the pass band center frequency f ₀ when the film thickness of the excitation electrode 131 and the reflectors 132 and 133 in FIG. 26 is changed. Indicates a change.

36A and 36B are comprehensively determined.
And the film thickness is in the range of 2600 to 4000
Is suitable because it does not produce ripples and has little loss deterioration.
That's right. This is almost determined from the center frequency of the filter.
Electrode period λ of the parallel arm resonator _P(4.4 at 932 MHz)
μm, see FIG. 26).
0.09.

This embodiment is based on the results of the above study.

FIG. 37 shows a first one-port surface acoustic wave resonator 200 of a surface acoustic wave filter according to the present invention.

[0144] In the figure, 201 excitation electrodes 202 and 203 are reflectors are made of Al mixed mixed with several percent dissimilar metals each made of Al or weight ratio, film thickness t ₁ is 0 electrode period λp The thickness is 0.06 to 0.09 times.

The pass characteristic of a surface acoustic wave filter in which this resonator 200 is applied to the resonators R1B, R3B, R5B in FIGS. 25 and 32 is as shown by a line 205 in FIG.
No ripple appears in the passband.

In the case where the above-mentioned Al alloy is used,
The power handling characteristics are improved as compared with the case of using Al. Dissimilar metals to be mixed are Cu, Ti and the like.

FIG. 39 shows a resonator 210 which is a modification of the above resonator. 211 is an excitation electrode, and 212 and 213 are reflectors. These are made of Au.

Since this phenomenon is caused by the effect of the mass addition effect, the optimum film thickness value range becomes smaller than the above value by the ratio with the density of Al. 1. Al density / Au density = 2.
Since 7 / 18.9 = 0.143, the optimum film thickness t
_2, and 0.143 times, 0.0086～ the electrode period lambda _P
The thickness is 0.013 times as large.

The pass characteristic of a surface acoustic wave filter in which this resonator 210 is applied to the resonators R1B, R3B, R5B in FIGS. 25 and 32 also becomes as shown in FIG. 38, and no ripple appears in the pass band.

[Embodiment 9] This embodiment is another example of realizing the inductances L ₁ , L ₂ and L ₃ in FIG.

In FIG. 40, portions corresponding to the components shown in FIG. 14 are denoted by the same reference numerals, and description thereof will be omitted.

[0152] 220 and 221 is a zigzag microstrip lines, are formed on the ceramic package 81 extend from the respective terminals 84 _-3 and 84 _-5.
The tips of the microstrip lines 220 and 221 are connected to the ground. Each microstrip line 22
The pattern width of 0,221 is 100 μm, and the length between the microstrip line and the ground is 0.5 mm. Assuming that the relative permittivity of the ceramic package 81 is 9, the inductance value of the microstrip lines 220 and 221 is 2 nH from the theoretical formula of the ribbon inductor.

[Embodiment 10] This embodiment is another example of realizing the inductances L ₁ , L ₂ and L ₃ in FIG.

In FIG. 41, portions corresponding to the components shown in FIG. 14 are denoted by the same reference numerals, and description thereof will be omitted.

Reference numerals 230 and 231 denote zigzag microstrip lines which extend from the resonators R ₁ and R ₂ , respectively, and are formed on the filter chip 82. A terminal 8 is provided at the tip of each microstrip line 230,231.
5 _-3 and 85 _-5 are formed. Each of the microstrip lines 230 and 231 has a thickness of 3000 mm and a width of 60 μm.
m, the total length is about 2 mm. Filter chip (LiTa
Assuming that the relative permittivity of O ₃ ) 82 is 44, the inductance value of the microstrip lines 230 and 231 is 2.2 nH according to the theoretical formula.

The inductance can be formed by appropriately combining the bonding wire 86 _-3 , the microstrip line 220 on the ceramic package 81, and the microstrip line 230 on the filter chip 82.

[Embodiment 11] FIG. 42 shows a surface acoustic wave filter 240 according to an eleventh embodiment of the present invention. FIG. 43 shows a configuration embodying this.

For convenience of explanation, the outline of this embodiment and the basic configuration of this embodiment will be described first.

Overview of the present embodiment In the present embodiment, the resonance frequency frs of the series arm resonator is appropriately set higher than the anti-resonance frequency fap of the parallel arm resonator.
It is intended to increase the pass band width, and to increase Δf≡frs
This is a configuration in which -fap is set within a range that does not significantly deteriorate the loss in the pass band.

Basic Configuration of this Embodiment In each of the above embodiments, fap = frs is an essential condition for forming a pass band of a filter.
However, as long as this condition is maintained, there is an upper limit in the pass band. In order to increase the pass bandwidth, FIG.
It was considered that fap <frs as shown in FIG.

Thus, as is apparent from FIG.
In the range of ap <f <frs, bx <0, and there is a possibility of becoming an attenuation region according to the above-mentioned theory. However, in reality, if the magnitude of Δf (= frs−fap) is limited as described below, the value of the bx product remains very small, so that attenuation does not occur, and there is substantially no problem as a pass band. It turned out there was no.

FIG. 45 shows the pass characteristics of the ladder filter when Δf = frs-fap is increased from zero.

As the experimental conditions, the piezoelectric substrate was made of LiTaO ₃ having an electromechanical coupling coefficient of 0.05, and Al electrode for a comb-shaped electrode.
The electrode was used under the condition of a film thickness of 3000 °.

The electrode configuration is basically a ladder-type configuration in which a parallel resonator and a series resonator are connected as shown in FIG. 42, which are cascaded in two stages, and the input side and output side are symmetrical. Are connected in the last stage for parallel operation. The reason why the ladder type circuit is multi-staged is to increase the degree of out-of-band suppression to a practical value.

However, since the number of stages increases the loss in the passband, the number of stages is adjusted according to the specific filter specifications. This example is an example of a configuration for achieving a loss of 2 dB or less and an out-of-band suppression of 20 dB or more. The design conditions of the comb-shaped electrode are as follows. Both resonators of the series-parallel arm have an opening length of 180 μm and a logarithm of 50 pairs. Since the logarithm and the opening length conditions of the series-parallel resonator are the same, the ratio P = Cp / Cs of the respective capacitances is 1.

FIG. 45A shows the case where Δf = 0, that is, the case of the above-described embodiment.

FIG. 17B shows the case where Δf = 10 MHz. Compared with FIG. 11A, the minimum insertion loss of the passband is hardly deteriorated, and the passband width (a bandwidth that guarantees a loss of 2.5 dB or less) is improved from 22 MHz to 40 MHz. The reason why the bandwidth is improved more than the increase in Δf is that loss recovery on the low frequency side of the pass band is observed.

Also, the degree of out-of-band suppression is improved. FIG.
In (A) and (B), the degree of out-of-band suppression (shown in the figure) on the high frequency side is improved from 19 dB to 20 dB.
As described above, the expansion of Δf is a technique that not only has an effect on bandwidth expansion but also can improve out-of-band suppression at the same time.

As described above, when Δf is increased, the characteristics are improved, but the amount of increase is limited.

FIG. 45C is a graph showing the transmission characteristics when Δf = 19 MHz. Loss degradation starts to be seen on the slightly higher frequency side in the pass band. In this case, it is about 2.5 dB.
This also causes an increase in in-band ripple. In this example, the ripple specification limit is about 1.0 dB. Any further increase in Δf resulted in loss degradation and increased in-band ripple. Therefore, Δf = 19 MHz is the limit when increasing Δf. Further, the out-of-band suppression degree at this time is about 21 dB as shown in FIG.
An improvement of 2 dB is obtained as compared with 5 (A).

At this time, the value of the above-mentioned bx product was examined by taking the case of Δf = 19 MHz in FIG. 45C as an example.

First, the surface acoustic wave resonator forming the parallel arm and the surface acoustic wave resonator forming the series arm shown in FIG. 42 are separately manufactured, and the circuit configuration shown in FIGS. 46 (A) and (B) is obtained. The parallel arm resonator measured admittance, and the serial arm resonator measured impedance. The measurement was performed using a network analyzer, and each S21 was measured. Then, the values were substituted into the equations shown in FIGS. 46A and 46B to obtain the impedance Zp and the admittance Yp.

As a result, a frequency characteristic as shown in FIG. 47 was obtained. This characteristic is the value of only the imaginary part of admittance and impedance, that is, the value of b or x.

When the value of the bx product is calculated from these, the frequency characteristics are as shown in FIG.

From the figure, it is assumed that b is within the range of fap <f <frs.
It can be seen that the x product has a small negative value.

The maximum value | bx _max | of the absolute value of the bx product is

[0177]

(Equation 5) And in the present example, it was 0.06. That is, when the value of | bx _max | is equal to or less than this value, it is understood that the deterioration of the insertion loss and the in-band ripple can be suppressed to 1 dB or less.

If Δf> 19 MHz, | bx _max |
The value also increases, and loss degradation and in-band ripple both become 1 dB or more, which is not practical.

Therefore, the value | bx _max | is an index of the upper limit of the characteristic deterioration, and determines the allowable value of Δf.

The following is a more generalized and detailed description.

FIG. 49 shows an equivalent circuit diagram when the surface acoustic wave resonator is approximated by an LC double resonance circuit as in FIG. 3 and assembled into a ladder type filter as shown in FIG.

If the impedance of the surface arm of the serial arm is Zs and the admittance of the surface arm of the parallel arm is Yp,

[0183]

(Equation 6) Becomes here,

[0184]

(Equation 7) It is.

When the bx product is obtained from the equations (11) and (12),

[0186]

(Equation 8) Becomes

The angular frequency ω giving the extreme value to bx in the equation (13) is obtained from ∂ (bx) / ∂ω = 0, and

[0188]

(Equation 9) Becomes

The value obtained by substituting this into the expression (13) becomes the maximum value of the absolute value of the bx product in the pass band. If you ask for this

[0190]

(Equation 10) Becomes

Here, Δω = ωrs−ωap = 2π · Δf (16)

When the equation (15) is plotted as a relation between bxmax and Δf / frs with P = Cop / Cos as a parameter, the result is as shown in FIG.

In FIG. 19, when the condition that the allowable value of the product of bx previously obtained is experimentally 0.06 or less is shown in the figure, the area becomes a hatched area.

Therefore, Δ which differs depending on P = Cop / Cos
The allowable value α of f / frs can be determined, and assuming that | bx _max | = 0.06 in equation (15), the following equation is obtained.

[0195]

[Equation 11]The capacity ratio γ is determined by the material of the substrate.
Cut propagation LiTaO _ThreeWas about 15.

Therefore, equation (17) is

[0197]

(Equation 12) Becomes

When P = 1, α = 0.02, and frs
Δf = 19M in the case of the embodiment of FIG.
Hz, and it can be confirmed that equation (18) holds.

The effect of increasing Δf is effective for a piezoelectric substrate material having a small capacitance ratio γ, that is, a substrate material having a large electromechanical coupling coefficient.
7) Formula was obtained.

Since the γ value is proportional to the reciprocal of the electromechanical coupling coefficient k ² , the other high electric power is obtained by using the γ value of 36 ° Y-cut X propagation LiTaO _{3 and} the value of k ² = 0.05. Material with mechanical coupling coefficient 64 ° Y-cut X-propagation LiNbO ₃
(K ² = 0.11) and 41 ° Y-cut X-propagation LiNb
When the γ value of O ₃ (k ² = 0.17) is obtained, the former is 6.
8. The latter is 4.4. These k ² values are described in the literature (“Applications for Piezoelectric Leaky Surface W
ave ": K.Yamanouchi and M.Takeuchi, 1990 ULTRASONICS
SYMPOSIUM Proceedings, pp11-18, 1990).

FIG. 51 shows the relationship between the capacitance ratio γ and the electromechanical coupling coefficient k ² .

The relationship shown in FIG.
Using the value of k ² and the γ value of O ₃ ,

[0203]

(Equation 13) It was obtained as.

From the relationship shown in FIG. 51, 64Y ° cut and 41
The γ value of the X-propagation LiTaO ₃ of the ° Y cut can be obtained, and γ = 6.8 and 4.4 as described above, respectively.

Structure of Embodiment 11 Here, the structure of the embodiment shown in FIGS. 42 and 43 will be described.

Reference numeral 241 denotes a 36 ° Y-LiTaO ₃ piezoelectric substrate having a size of 1.5 × 2 × 0.5 mm.

The parallel arm resonators (R
p ₁ ), series arm resonator (Rs ₁ ), parallel arm resonator (Rs ₁ )
p ₂ ), series arm resonator (Rs ₂ ), parallel arm resonator (Rs ₂ )
p ₃ ). Each resonator has a structure having reflectors 242 (short-circuit type) on both sides. Each resonator has an opening length of 180 μm and a pair of electrode fingers of 50 μm.
There are 50 pairs and reflectors.

Only the period of the comb-shaped electrode fingers is changed between the parallel arm resonator and the series arm resonator. The period of the parallel arm resonator is λp =
4.39 μm (the pattern width is λp / 4 ≒ 1.1 μm because the pattern width and the gap are 1: 1), and the period of the series resonator is λs = 4.16 μm (similarly, the pattern width is λs
/4=1.04 μm).

In each cycle, the resonance frequency (frp, frs) of each resonator is a predetermined value (frp = 893).
MHz, is obtained by determining from frs = 942MHz) and so as to λs = V _m / frs, and λp = V _m / frp. Here, V _m is 30
The sound velocity of a surface wave of a 36 ° Y-cut X-propagating LiTaO ₃ crystal at 00 °, which was experimentally determined to be V _m = 3920 _m / s.

[0210] The surface acoustic wave filter 240 having the above-described configuration includes:
It has a broadband and low-loss transmission characteristic shown in FIG. Note that Δf = 19 MHz.

In FIG. 43, when only λp is changed to 4.35 μm
Then, Δf becomes 10 MHz, and the characteristic of FIG. 45B is obtained. The electrode material is an Al-Cu alloy,
The film has a thickness of 3000 ° and is arranged so that the surface wave propagates in the X-axis direction of the piezoelectric substrate 241.

Next, an example in which another piezoelectric substrate is used will be described.

In the case of 64 ° Y-cut X-propagation LiNbO ₃ , γ = 6.8, and equation (17) gives:

[0214]

[Equation 14] Becomes

In the case of 41 ° Y-cut X-propagation LiNbO ₃ , γ = 4.4,

[0216]

(Equation 15) Becomes

As the γ value becomes smaller, that is, as the substrate has a larger electromechanical coupling coefficient, α becomes larger, and even if Δf is made larger, the characteristic hardly deteriorates.

[Twelfth Embodiment] FIG. 52 is a circuit diagram of a surface acoustic wave filter 250 according to a twelfth embodiment of the present invention.

FIG. 53 shows a structure embodying the surface acoustic wave filter having the circuit configuration of FIG.

FIGS. 54 and 55 show the characteristics of the surface acoustic wave filter shown in FIGS.

Overview of Embodiment For the convenience of explanation, an overview of this embodiment will be described first.

The surface acoustic wave filter of this embodiment is a ladder type surface acoustic wave filter in which a plurality of surface acoustic wave resonators are connected in series and parallel to each other.
In this configuration, the image impedance is matched between the unit sections each consisting of one unit, and the loss at each connection point is reduced.

As a result, it is possible to reduce the insertion loss in the pass band.

Thinking process until the invention is completed Next, the thinking process until the present invention is completed will be described.

As shown in FIGS. 56A and 56B, a band-pass characteristic can be obtained by connecting at least one series arm resonator and a parallel arm resonator in a ladder form. In addition,
The ladder-type connection of each series arm resonator and parallel arm resonator constitutes a unit section of the filter.

In this case, it is desirable that the resonance frequency of the series arm resonator and the anti-resonance frequency of the parallel resonator match, or that the former has a higher frequency than the latter in view of the expansion of the passband. There are two types of unit sections in FIGS. 56 (A) and (B) depending on whether the input / output ends are serial arms, and those connected in multiple stages are shown in FIGS. 57 (A), (B) and They are classified into three types as shown in FIG.

FIG. 57 (A) shows the case where one of the input / output sides is a serial arm and the other is a parallel arm (asymmetric type), and FIG. 57 (B) shows the case where the input / output ends are both parallel arms (symmetric type). FIG. (C) shows a case where both the input and output terminals are series arms (symmetric type).

When the number of stages is increased in this way, both the insertion loss and the out-of-band suppression are n times as large as the unit section, and although the insertion loss generally worsens, the out-of-band suppression is improved. In particular, when the loss in the unit section is close to 0, this multi-stage is effective.

However, if the impedance matching in the pass band between the unit sections is not appropriate, the insertion loss will be worse than the theoretical n times.

If the impedance matching is not appropriate, power reflection occurs at the boundary of the unit section (each boundary from line 1-1 ′ to nn ′ in FIG. 57), resulting in an increase in loss.

Assuming that the power reflection between the unit sections is Г, the loss is also n10log (Г). Therefore, it is important to reduce the insertion loss as much as possible by measuring the impedance between the unit sections and suppressing the power reflection at the boundary.

Next, a method for achieving impedance matching between unit sections will be described.

As shown in FIG. 58, when circuits having two different four-terminal constants (four constants A, B, C, and D of the F matrix) are generally connected with each other for impedance matching, a boundary b− It suffices that the image impedances when viewing the respective circuit sides from b ′ are equal to each other.

As shown in FIG. 58, the image impedance Zi1 seen from the circuit 1 side is the four-terminal constant A ₁ of the circuit ₁ ,
It is expressed as follows using B ₁ , C ₁ , and D ₁ .

[0235]

(Equation 16) Similarly, the image impedance Zi2 looking at the circuit 2 side is
It is expressed as the following equation.

[0236]

[Equation 17] These image impedances are load resistance (pure resistance)
It is determined independently of R ₀ .

Assuming that equations (21) and (22) are equal, an impedance matching condition as follows is obtained.

D ₁ B ₁ / C ₁ A ₁ = A ₂ B ₂ / C ₂ D ₂ (23) FIG. 59 shows a case where the above-described impedance matching condition is applied to a unit section of a ladder-type circuit.

FIG. 59A shows a connection method with poor impedance matching, which does not satisfy the condition of equation (23).

The reflection coefficient た as seen from the boundary bb ′ on the right side
Is

[0241]

(Equation 18) Becomes Z _s Y _p is completely zero even in the pass band in an actual element.
Does not become zero because it does not become zero.

In comparison with FIG. 59B or FIG.
In (C), since the boundary bb 'satisfies the condition of the expression (23), the reflection becomes 0 and no loss occurs.

For example, in the case of FIG.
The image impedance seen from the left side from b ′ is (21)
From the formula,

[0244]

[Equation 19] Becomes From the equation (22), it can be seen that the image impedance Zi2 on the right side of the boundary bb 'is also equal to Zi1.

Therefore, impedance matching is achieved, and the reflection coefficient at the boundary becomes zero.

FIG. 59 (C) also proves that impedance matching has been achieved.

Next, a method of connecting the unit sections in multiple stages using the connection method shown in FIGS. 59 (B) and 59 (C) will be considered.

FIG. 60A shows a circuit in which n (> 2) unit sections are connected by repeating the connection method shown in FIGS. 59B and 59C alternately. If such a connection method is used, power reflection in each unit section does not occur regardless of how many stages are connected for the above-described reason.

In the configuration of FIG. 60 (A), if the resonators of the parallel arm or the resonators of the serial arm which are adjacent to each other are added and put together, it becomes equivalent to FIG. 60 (B).

As a result, it can be seen that only the arm closest to the input / output end has an impedance or admittance value that is half that of the inner arm.

When this principle is applied to the three types of multi-stage methods shown in FIG. 57, as connection methods for achieving impedance matching, FIGS. 61 (A), (B), and
The method of (C) is obtained.

FIG. 61A shows a matching connection method corresponding to FIG. 57A, in which one of the input and output ends is a serial arm and the other is a parallel arm. In this case, the impedance value of the series arm resonator at the end is half the impedance value of the inner series arm resonator, and the admittance value of the parallel arm resonator at the other end is also equal to the inner parallel arm resonator. Half the admittance value of the vessel.

Similarly, FIG. 61 (B) shows the matching connection method of FIG. 57 (B), and FIG. 61 (C) shows the matching connection method of FIG. 57 (C).

In the case of FIG. 61B, both ends are parallel arms, and the admittance value of the parallel arm resonators at both ends is half of the admittance value of the parallel arm resonators inside.

In the case of FIG. 61 (C), both ends are series arms, and the impedance value of the series arm resonators at both ends is half of the impedance value of the series arm resonator inside it.

Next, a description will be given of a twelfth embodiment of the present invention based on the above concept.

FIG. 52 shows a basic structure of a surface acoustic wave filter 250 according to a twelfth embodiment of the present invention.

When this surface acoustic wave filter is embodied,
As shown in FIG.

The three series arm resonators (Rs ₁ , Rs ₂ , Rs
₃ ) and three parallel arm resonators (Rp ₁ , Rp ₂ , Rp ₃ ), which are connected as shown in the equivalent circuit of FIG. These six resonators all have an aperture length (90 μm).
m) are the same, and the number of electrode finger pairs (100 pairs) is also the same. Each resonator has a short-circuited reflector on both sides as shown in FIG. The log of the reflector is 1
It is about 00 pairs.

The series arm resonators (Rs _{1 to} Rs ₃ ) have the same length of the electrode finger period (λs), and λs = 4.19 μm.
m. The period of the parallel arm resonators (Rp _{1 to} Rp ₃ ) is set to a different period λp = 4.38 μm.

FIG. 62 shows a comparative example of this configuration as a comparison target.

[0262] For both FIGS. 52 and 62, the design conditions of the one-terminal SAW resonator series arm represented by impedance Z _s is the aperture length 90 [mu] m, a logarithmic 100 pairs. One-terminal-pair surface acoustic wave resonator in the parallel arm represented by admittance Y _p is the same condition.

The piezoelectric substrate crystal has a 36 ° Y cut X propagation L
iTaO ₃ was used and a surface acoustic wave resonator 3
A comb-shaped pattern of an Al alloy film of 000 ° is formed.

In FIG. 54, the solid line 251 indicates the characteristic of the filter 250 shown in FIG. A broken line 252 indicates the characteristic of the filter of the comparative example of FIG. From both, the filter 25 of the present embodiment
It can be seen that 0 is lower loss. In particular, the improvement at both ends of the pass band is large.

Next, in the filter of the comparative example of FIG.
Parallel resonator represented by admittance Y _p of unit sections (3) only, in FIG. 55 the passage characteristic when a smaller value of admittance Y _p to reduce the logarithm to 80 to 100 pairs, indicated by the line 253. Similarly, it can be seen that the insertion loss is improved. Therefore, it can be said that even if the admittance value at the end is not reduced to 1/2, simply reducing it from the admittance value at the inner side is not sufficient, but is effective. The same applies to the impedance value.

Although the embodiment for the basic form shown in FIG. 61A has been described above, the same effect can be obtained even if a large number of unit sections increase in the center.

Embodiment 13 FIG. 63 shows a thirteenth embodiment of the present invention.
9 shows a surface acoustic wave filter 260 according to an embodiment.

This surface acoustic wave filter is shown in FIG.
This is based on the configuration method shown in FIG. This surface acoustic wave filter 260 has a loss reduction effect similar to that shown by the line 251 in FIG.

[Embodiment 14] FIG. 64 shows a fourteenth embodiment of the present invention.
This is a surface acoustic wave filter 270 according to an embodiment.

This surface acoustic wave filter is shown in FIG.
This is based on the configuration method shown in FIG. This surface acoustic wave filter 270 also has a loss reduction effect similar to that shown by the line 251 in FIG.

[Embodiment 15] Next, a surface acoustic wave filter according to a fifteenth embodiment of the present invention will be described with reference to FIGS.

Overview of Embodiment For convenience of explanation, an overview of this embodiment will be described first.

This embodiment focuses on the resistance and the conductance of the comb-shaped electrode as the cause of determining the insertion loss in the pass band. For the resonators described above, by reducing the conductance, the insertion loss of the filter characteristics when these are assembled in a ladder form is reduced.

Next, the background and the like of the present invention will be described.

FIG. 65 shows the resonance frequency (fr) applied to each of the series arm and the parallel arm.
2 shows a basic configuration of a ladder-type filter circuit in which two surface acoustic wave resonators having different (s, frp) are arranged.

Here, the admittance of the parallel arm resonator is represented by Y _p = g + j · b (26) g: conductance b: susceptance

The impedance of the series arm resonator is given by Z _s = r + j · x (27) where r is a resistance and x is a reactance.

With this assumption, the frequency characteristics of g, b, r, and x are as shown in FIG.

[0279] susceptance component b admittance Y _p of the parallel arm resonator (dotted line in FIG. 69) takes the maximum value at the resonance frequency frp, where codes from + - changing to 0 at the anti-resonance frequency fap (zero ), The sign becomes + again at fap or more, and gradually increases.

On the other hand, the conductance g of Y _p (FIG. 69)
Similarly, the dashed line in the drawing) takes the maximum value at fap, and
When it exceeds p, it decreases rapidly and gradually approaches zero.

It should be noted that the conductance g has only a positive value.

The reactance x (solid line in FIG. 69) of the impedance Z _s of the series arm resonator is opposite to the admittance and becomes 0 at the resonance frequency frs, and the anti-resonance frequency fa
take the maximum value at s, change the sign from + to-, and
Above s, it approaches 0 from one side.

Further, the resistance r gradually increases from 0, reaches a maximum value at the anti-resonance frequency fas, and gradually decreases above that. r also takes only the value of + similarly to g.

Here, in order to produce filter characteristics, it is required that the anti-resonance frequency fap of the parallel resonator and the resonance frequency frs of the series resonator be substantially the same or that the latter is slightly larger.

The lower part of FIG. 69 shows the pass characteristic as a filter circuit in accordance with the upper impedance and admittance frequency characteristics. A pass band is set near fap ≒ frs, and the rest is an attenuation region. As can be seen from the figure, b and x become 0 especially in the vicinity of the center frequency of the pass band.

Therefore, the pass characteristic as a filter is S21
Is determined only by r and g,

[0287]

(Equation 20) Becomes

Here, since r> 0 and g> 0, (2
In equation (8), as both r and g increase, S21 becomes smaller than 1, and the insertion loss represented by 20log | S21 | also increases.

Therefore, as r and g are both closer to 0, the insertion loss is smaller.

Next, a description will be given of what portions of the comb-shaped electrodes forming the surface acoustic wave resonator have r and g.

Here, in FIG. 3B, r ₁ is also taken into consideration.

R ₁ is the sum of the electric resistance of the comb-shaped electrode and the energy loss that leaks from each end of the comb-shaped electrode finger into the substrate as a bulk wave as acoustic resistance.

Since the resistance due to the emission of the bulk wave hardly depends on the shape of the comb-shaped electrode, the electric resistance r _{1 of the} comb-shaped electrode is small.
Is proportional to Especially near the center frequency of x = 0, r = r ₁
Becomes The admittance conductance g of the parallel arm resonator is equal to the conductance 1 of the electric resistance of the comb-shaped electrode.
/ R ₁ .

Now, assuming that the resistivity of the electrode finger of the comb-shaped electrode is ρ _o , the width of the electrode finger is W, the film thickness is t, the opening length of the series arm resonator is l _s , and the logarithm is N _s , r = l _s · ρ _o / (N _s · W · t) (29)

If the opening length of the parallel arm resonator is l _p and the logarithm is N _p , when the same substrate and the same metal film are used, ρo
, W, t are substantially equal, so that g = N _p · W · t / (l _p · ρ _o ) (30)

[0296] Thus, the increase in insertion loss in the equation (28) is, r + 50r · g + 2500g = l s · ρ o / (N s · W · t) +50 · (l s / l p) · (N p / N _{s) +2500 · N p · W} · t / (l p · ρ o) ... is (31).

From equation (31), the series arm resonator has an aperture length l
_s is short, as the logarithmic N _s is large, also, the parallel resonators aperture length l _p is long, as the logarithmic N _p is small, it can be seen that the effect on the loss reduction. In particular, as l _s / l _p <1, N _p / N _s <1, in other words, the opening length of the series arm resonator is shorter than that of the parallel arm resonator, and the logarithm is the series arm resonator. Is more effective than the parallel arm resonator.

Now, the reason will be described.

In the above equation (31), r = r _s (r _s :
Series arm resonators electrical resistance) and _{g = 1 / r p (r} p: Since a parallel electrical resistance arm) r + 50r · g + 2500g = r s +50 (r s / r p)
+2500 to become (1 / r _p). Therefore, increase of the insertion loss if _{(r s / r p) <} 1, i.e. r _s <r _p can be suppressed.

In this case, if l _s is too small, loss due to surface wave diffraction appears. Conversely, if l _p is too long, the Q of the parallel resonator is reduced due to an increase in resistance, and out of band on the low frequency side. Since the degree of suppression is worse, its size is limited.

[0301] Further towards the t _s of the film thickness series arm of the metal film forming the comb-shaped electrodes, towards the parallel arm When t _p (3
The expression 1) is as follows.

[0302] r + 50r · g + 2500g = l s · ρ o / (N s · W · t) +50 · (l s / l p) · (N p / N s) (t p / t s) +2500 · N p · W · t _p / (1 _p · ρ _o ) (32) Accordingly, the loss can be similarly reduced by setting t _p / t _s .

[0303] In addition, the resistivity is different (ρos, ρop)
It is possible to make a filter by arranging resonators composed of two kinds of metal films in a series arm and a parallel arm to make ρos / ρop <1, but in the case of actually making an element, consider mass productivity etc. Then it is not practical.

Configuration of Embodiment 15 Next, an embodiment employing the above concept will be described.

FIG. 65 shows a circuit configuration of a surface acoustic wave filter 280 according to a fifteenth embodiment of the present invention.

FIG. 66 shows a structure embodying the circuit configuration of FIG. 65. The used piezoelectric substrate 241 is 36 ° Y-cut XL
iTaO ₃ , and the electrode material is a 3000 ° Al film.

In the comparative example, both the serial arm and the parallel arm have the comb-shaped electrode opening length l _s = l _p = 90 μm and the logarithm N _p = N _s = 100 pairs. , L _s = 45
μm, N _s = 200 pairs, the parallel arm is l _p = 180 μm, N _p
= 50 pairs. l _p > l _s and N _s > N _p . Also, a _{= 0.25 l s / l p =} 0.25 and N _p / N _s.

At this time, the capacitance C _O of the comb-shaped electrode, which is determined by the product of the logarithm and the opening length, was kept unchanged.

In FIG. 66, the solid line 281 is the characteristic of this embodiment, and the broken line 282 is the characteristic of the comparative example. Conventionally, the loss was 2.5
What was dB was changed to 2.0 dB according to the present embodiment,
According to the present embodiment, it is improved by 0.5 dB or more. That is, the insertion loss of the filter was improved by 25% in dB.

In the case of the present embodiment, the power durability was also improved by increasing the number of series arm resonators, and the maximum power that could be applied was improved by 20%.

In the case of the above embodiment, diffraction loss began to appear at l _s = 30 μm or less, and out-of-band degradation on the low frequency side began to occur at l _p = 300 μm or more. .

As described above, it is apparent that the insertion loss in the pass band can be improved by reducing the electric resistance of the series arm and increasing the electric resistance of the parallel arm (reducing the conductance).

It is also possible to adopt a configuration in which the thickness of the parallel arm resonator is thinner than the thickness of the series arm resonator. With this configuration, as in the case of the above-described embodiment, the loss of the pass band can be reduced.

As described above, according to the present invention, since a plurality of parallel arms and a series arm are formed on one piezoelectric substrate, the resonance frequency of the surface acoustic wave resonator can be accurately controlled.
When the frequency control of the filter for the broadband can be achieved
Both reduce insertion loss in the passband when multi-stages are used.
can do.

[Brief description of the drawings]

FIG. 1 is a principle diagram of a surface acoustic wave filter according to the present invention.

FIG. 2 is a diagram showing a basic configuration of a filter circuit using a resonator.

FIG. 3 is a diagram showing the structure of a one-port surface acoustic wave resonator, its equivalent circuit, and its symbol.

FIG. 4 is a diagram showing frequency characteristics of impedance and admittance of a one-port surface acoustic wave resonator.

5 is a diagram showing the immittance characteristics of the surface acoustic wave resonator near the resonance frequency and the filter characteristics of the filter of FIG. 1 formed by connecting them.

FIG. 6 is a diagram illustrating a conventional surface acoustic wave filter.

FIG. 7 is a diagram illustrating an effect when an inductance is added to a resonator in series.

FIG. 8 is a diagram showing an effect when n one-port surface acoustic wave resonators are connected in series.

FIG. 9 is a diagram showing the dependence of the pass characteristic of the parallel arm resonator on the aperture length.

FIG. 10 is a diagram showing the aperture length dependence of the transmission characteristic of the series arm resonator.

FIG. 11 is a circuit diagram of a first embodiment of a surface acoustic wave filter according to the present invention.

FIG. 12 is a diagram illustrating a pass characteristic of the filter of FIG. 11;

FIG. 13 is a diagram showing the effect of adding inductance to the parallel arm resonator.

FIG. 14 is a plan view showing the structure of the surface acoustic wave filter of FIG. 11 with its lid removed.

FIG. 15 is a sectional view taken along line XV-XV in FIG. 41;

FIG. 16 is a view showing a second embodiment of the surface acoustic wave filter according to the present invention.

FIG. 17 is a diagram illustrating a pass characteristic of the filter of FIG. 16;

18 is a diagram showing the effect of increasing the aperture length of the parallel arm resonator (A _P) and aperture length of the series arm resonator ratio _{(A S) (A P /} A S).

FIG. 19 is a diagram showing a third embodiment of the surface acoustic wave filter according to the present invention.

FIG. 20 is a diagram showing the pass characteristics of the filter of FIG. 19;

FIG. 21 is a diagram showing a fourth embodiment of the surface acoustic wave filter according to the present invention.

FIG. 22 is a diagram illustrating pass characteristics of the filter of FIG. 21;

FIG. 23 is a diagram showing a fifth embodiment of the surface acoustic wave filter according to the present invention.

FIG. 24 is a diagram showing the pass characteristics of the filter of FIG. 23;

FIG. 25 is a circuit diagram of a surface acoustic wave filter according to a sixth embodiment of the present invention.

FIG. 26 is a diagram showing a first one-port pair elastic surface resonator in FIG. 25;

FIG. 27 is a diagram showing the pass characteristics of the filter of FIG. 25;

FIG. 28 is a diagram illustrating the influence of β on the ripple width due to the reflector installation position d = (n + β) · λ.

FIG. 29 is a plan view showing the structure of the surface acoustic wave filter of FIG. 25 with its lid removed.

FIG. 30 is a diagram showing a modification of the first one-port surface acoustic wave resonator shown in FIG. 25;

FIG. 31 is a view showing another modification of the first one-port surface acoustic wave resonator shown in FIG. 25;

FIG. 32 is a view showing a seventh embodiment of the surface acoustic wave filter according to the present invention.

FIG. 33 is a diagram showing an effect of an electrode film thickness (t) on a ripple occurrence position.

FIG. 34 is a diagram illustrating a state where a ripple (r _P ) due to a reflector of a parallel arm resonator falls to a high frequency attenuation pole.

FIG. 35 is a diagram showing the film thickness dependence of the pass characteristic of the resonator type filter.

FIG. 36 is a diagram showing the results of an experiment on the dependency of the insertion loss and the position of occurrence of ripple on the film thickness.

FIG. 37 is a diagram showing a first one-port surface acoustic wave resonator according to an eighth embodiment of the surface acoustic wave filter of the present invention.

FIG. 38 is a diagram showing the pass characteristics of a surface acoustic wave filter to which the resonator shown in FIG. 37 is applied;

FIG. 39 is a view showing a modification of the first one-port surface acoustic wave resonator according to the eighth embodiment of the surface acoustic wave filter of the present invention.

FIG. 40 is a diagram showing another example for realizing the inductance of the surface acoustic wave filter of FIG. 11;

41 is a diagram showing still another example for realizing the inductance of the surface acoustic wave filter of FIG. 11;

FIG. 42 is a circuit diagram of an eleventh embodiment of a surface acoustic wave filter according to the present invention.

FIG. 43 is a diagram showing a configuration embodying the circuit of FIG. 42;

FIG. 44 is a diagram showing the immittance characteristics of a surface acoustic wave resonator when fap <frp.

FIG. 45 is a diagram illustrating a change in the pass characteristic of the ladder-type filter when Δf≡frs-fap is increased from zero.

FIG. 46 is a diagram illustrating a method for measuring characteristics of a surface acoustic wave resonator.

FIG. 47 is a diagram illustrating characteristics of admittance and impedance of each surface acoustic wave resonator of the parallel arm and the serial arm.

FIG. 48 is a diagram showing the frequency dependence of the bx product.

FIG. 49 is a diagram showing a part of the circuit of FIG. 42 as an equivalent circuit of L and C.

FIG. 50 is a diagram showing a relationship between | bx _max | and Δf / frs.

FIG. 51 is a diagram showing a relationship between k ² and γ.

FIG. 52 is a circuit diagram of a twelfth embodiment of a surface acoustic wave filter according to the present invention.

FIG. 53 is a diagram showing a configuration embodying the circuit of FIG. 52;

FIG. 54 is a diagram showing characteristics of the surface acoustic wave filter of FIG. 53.

In the filter of FIG. 55 FIG. 63 is a diagram showing a characteristic when reduced output Y _p.

FIG. 56 is a circuit diagram of a unit section in which one surface acoustic wave resonator is formed in a ladder shape.

FIG. 57 is a circuit diagram of a circuit in which the unit sections of FIG. 56 are connected in multiple stages (n stages).

FIG. 58 is a diagram showing connections between two four-terminal circuits and boundaries thereof.

FIG. 59 is a diagram showing joining of unit sections.

FIG. 60 is a diagram illustrating a method of connecting a unit section to n (> 2) stages.

FIG. 61 is a diagram illustrating a configuration method of the ladder-type circuit according to the present embodiment.

FIG. 62 is a circuit diagram of a conventional surface acoustic wave filter.

FIG. 63 is a circuit diagram of a thirteenth embodiment of a surface acoustic wave filter according to the present invention.

FIG. 64 is a circuit diagram of a fourteenth embodiment of a surface acoustic wave filter according to the present invention.

FIG. 65 is a circuit diagram of a surface acoustic wave filter according to a fifteenth embodiment of the present invention.

FIG. 66 is a diagram showing a configuration embodying the circuit of FIG. 65;

FIG. 67 is a view showing characteristics of the filter of FIG. 66.

FIG. 68 is a diagram showing a ladder-type filter circuit in which surface acoustic wave resonators having different resonance frequencies are arranged in a parallel arm and a series arm.

FIG. 69 is a diagram correspondingly showing a frequency characteristic of admittance (Y _p ) of a parallel arm resonator and a frequency characteristic of impedance (Z _s ) of a series arm resonator.

FIG. 70 is a diagram showing an example of a conventional surface acoustic wave filter.

FIG. 71 is a diagram showing the pass characteristics of the filter of FIG. 70;

[Explanation of symbols]

60, 90, 100, 110, 120, 130, 24
0, 250, 260, 270, 280 Surface acoustic wave filter 80, 150 Surface acoustic wave filter device 81 Ceramic package 82 Filter chip 83 Lid 84 _{-1 to} 85 _-6 Electrode terminal 85 _{-1 to} 85 _-5 Terminal 86 _-1 86 _-5 bonding wires 125 attenuation pole 127 stopband 131,201,211 excitation electrode 132,133,160,161,166,167,2
02, 203, 212, 213, 242 Reflector 220, 221, 230, 231 Microstrip line 241 36 ° Y-cut X-propagation LiTaO ₃ substrate (chip) Rs ₁ , Rs ₂ Series arm resonator Rp _{1 to} Rp ₃ Parallel arm Resonator

Continued on the front page (72) Inventor Takashi Matsuda 1015 Uedanaka, Nakahara-ku, Kawasaki City, Kanagawa Prefecture Inside Fujitsu Limited (72) Inventor Mitsuo Takamatsu 1015 Uedanaka, Nakahara-ku, Nakazaki-ku Kawasaki City, Kanagawa Prefecture Fujitsu Limited (56) Reference Document JP-A-63-132515 (JP, A) JP-A-53-123051 (JP, A) JP-A-58-171120 (JP, A) JP-A-52-19044 (JP, A) JP-A 51-190 36091 (JP, A) JP-A-64-81403 (JP, A) JP-A-3-205908 (JP, A) JP-A-1-260911 (JP, A) JP-A-57-178812 (JP, A) JP-A-57-2105 (JP, A) (58) Fields investigated (Int. Cl. ⁷ , DB name) H03H 9/64 H03H 9/145

Claims (2)

(57) [Claims] A plurality of surface acoustic wave resonators;
Parallel arm and series arm are directly connected in a ladder
The surface acoustic wave resonators are formed on a circuit board and have band-pass characteristics.
And a first one-port bullet having a predetermined resonance frequency (frp)
A surface acoustic wave resonator is attached to each of the parallel arms,
Approximately or larger than resonance frequency (fap)
Terminal-to-elastic surface with high resonance frequency (frs)
Comb-shaped electrode connecting a wave resonator to each of the series arms to form the second one- port surface acoustic wave resonator
Is equivalent to the first terminal-to-surface acoustic wave.
Than the electric resistance (rp) of the comb-shaped electrode forming the resonator
A comb-shaped electrode having a small configuration and forming the second one-port pair surface acoustic wave resonator
Of the first terminal-pair surface acoustic wave
Than the electrical resistance (rp) of the comb-shaped electrode constituting the resonator
Means for reducing the size of the second one-terminal surface acoustic wave resonator
The opening length (ls) of the comb-shaped electrode to be formed is the first terminal.
The opening length of the comb-shaped electrode constituting the surface acoustic wave resonator (l
p), and the second one-port pair surface acoustic wave
The logarithm (Ns) of the comb-shaped electrodes constituting the vibrator is the first one.
The logarithm (N) of the terminal-to-comb electrode forming the surface acoustic wave resonator
A surface acoustic wave filter having a configuration larger than p) . 2. A second one-port surface acoustic wave resonator is formed.
The electric resistance (rs) of the comb-shaped electrode is changed by the first terminal.
The electric resistance of the comb-shaped electrode that constitutes the surface acoustic wave resonator
The means for making it smaller than (rp) is the first one-port bullet.
Of Comb-shaped Electrode Made of Metallic Thin Film Constituting Conductive Surface Wave Resonator
A film of the same metal as the second one-port surface acoustic wave resonator.
2. The structure according to claim 1, wherein the thickness is smaller than the thickness.
3. The surface acoustic wave filter according to 1.