JPH1080145A - Resonance-type power conversion apparatus - Google Patents

Abstract

PROBLEM TO BE SOLVED: To guarantee stable operation which is synchronized with a resonance current and to prevent a leading phase causing a switching loss so as to be operated always at a lagging phase in a resonance-type power conversion apparatus which takes into consideration an operating frequency at several MHz. SOLUTION: A resonance-type power conversion apparatus is provided with an inverter provided with a plurality of voltage drive-type semiconductor elements Q1, Q2 having a function which does not prevent a reverse current and with a resonance means which contains a reactor Lr and a capacitor Cr, and it supplies an AC current. In the resonance-type power conversion apparatus, charging voltage means C1, C2 which charge respective voltage drive-type semiconductor elements according to currents flowing in the semiconductor elements are installed, drive means 5, 6 which apply control voltages to respective control terminals of the voltage drive-type semiconductor elements at respectively arbitrary timings are provided, a comparison means 9 which compares a charging voltage with a reference voltage is provided, the control voltages are applied to the voltage drive-type semiconductor elements on the basis of the output of the comparison means, and the respective elements are turned on and off.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a resonance type power conversion device, and more particularly to a resonance type power conversion device employed for a lighting device having a high resonance frequency or a resonance type active filter for suppressing harmonics.

[0002]

2. Description of the Related Art Generally, a fluorescent lamp has a negative resistance, so that a ballast is required to stabilize an alternating current flowing through the lamp. 2. Description of the Related Art In recent years, a resonant inverter has been used as an electronic ballast for increasing the frequency of a lamp current in order to prevent flickering of the lamp and improve luminous efficiency. In this resonance type inverter, two power semiconductor switching elements having a function of preventing a reverse current are connected in series, and a resonance reactor (referred to as a choke coil), a capacitor and a lamp are connected to an output terminal corresponding to the connection point. Are generally connected in series.
The current flowing through the resonance reactor is controlled by changing the operating frequency of the inverter. Assuming that a switching frequency at which two power semiconductor elements are turned on and off alternately is f, and a resonance frequency determined by the reactor and the capacitor is fo, if f is not constant with respect to fo, the lamp current also changes. In order to stabilize the driving frequency of the switching element, there is a control circuit as disclosed in JP-A-8-9655. The control circuit detects a voltage between both ends of a freewheel diode (having a function of preventing a reverse current) provided in the power semiconductor switching element, turns on the power semiconductor switching element, and uses a integrator to reduce the current of the power semiconductor switching element. It is characterized in that the switching is performed in synchronization with the resonance current flowing through the lamp by a control method of integrating and turning off the element when this value is higher than a reference value.

[0003]

The switching frequency of a general inverter for lighting a fluorescent lamp is about 50 kHz.
A high-frequency AC voltage is applied between electrodes provided at both ends of the lamp, and plasma is maintained by a high electric field generated in the lamp tube. In recent years, there has been reported an electrodeless lamp in which the frequency is increased to several MHz, a high-frequency magnetic field is generated by a high-frequency AC current, and the magnetic field maintains the plasma in the lamp tube. Alternatively, even in a general resonance type power supply, since the volume of the resonance reactor and capacitor can be reduced in accordance with the frequency, it has been studied to increase the switching frequency to MHz. In the case of such a resonance type inverter of several MHz, there is a concern about operation delay in a method of turning off the element by comparing the integrated value of the current with the reference value as in the above-mentioned known example. In a known example, a part of the current flowing through the power semiconductor switching element is charged in a capacitor in an integrator, and the charged voltage is compared with a reference value.
The charged voltage is discharged by the integrator itself by the next switching time. Here, the time during which the integrator discharges the charging voltage is determined by the resistance value of the element inside the integrator, but the discharge time changes due to variation in element characteristics or a change in resistance value due to temperature rise. The variation in the discharge time is, for example, 0.1
μs, the operation delay for the ON period of the element is 50
For a frequency of kHz it is only 0.5% of one wavelength, but for 2 MHz it can be as much as 20% of one wavelength. As described above, when the high-frequency resonance type inverter of several MHz is controlled by the conventional method, there is a problem that the variation of the switching frequency due to the operation delay becomes considerably large.

In the case of a lamp lighting device, there is a relationship as shown in FIG. 8A between the switching frequency f of the inverter and the resonance frequency fo determined by the reactor and the capacitor, and the operation at the resonance point (f = fo) is determined. At the boundary, f> fo is called a lag phase, and f <fo is called a lead phase. Regardless of whether the phase is lagging or leading, the current IL of the resonance circuit decreases as compared with the value at the resonance point. Therefore, it is desired to operate the inverter near the resonance point. However, in the case of the advanced phase, a problem occurs in which a current flowing through the inverter flows. Hereinafter, the problem in the case of the advanced phase will be specifically described. As shown in FIG. 8B, the power semiconductor switching elements are Q1 and Q2, and the circulating diodes incorporated therein are QD1 and QD2, respectively.
Assuming that the output voltage taken out from the connection position of Q1 and Q2 is Vo, the operation in the case of the advanced phase is as shown in FIG. In FIG. 8 (c), the output voltage V of the inverter
The waveform of the resonance current IL leads the waveform of o by the phase ψ. In the case of this leading phase, as a one-cycle operation, the resonance current IL switches from positive to negative during the ON period of Q1, and the current flows through QD1. Next, when the control circuit turns off Q1 and turns on Q2, on the contrary, a reverse voltage is suddenly applied to QD1 which had flowed a forward current before that.
As a result, the electrons and holes accumulated inside QD1 (hereinafter, referred to as QD1)
It is called a residual carrier. ) Is discharged, and a reverse current (hereinafter referred to as a reverse recovery current) flows through the QD1 from the cathode to the anode. This current flows through Q2 and becomes a through current for the inverter. The time during which the residual carriers are discharged is described in the specification of the device as the reverse recovery time of the diode, and is 0.05 to 0.1 μs even in a device having a short reverse recovery time called a high-speed diode. Several MHz
When operating the resonance type inverter near the resonance point, there is a high possibility that a leading phase is generated due to a variation in switching frequency, and since the frequency is high, a loss due to a through current becomes a factor which determines a thermal operation limit of the inverter device. .

SUMMARY OF THE INVENTION An object of the present invention is to provide a resonance type power converter that takes into account an operating frequency of several MHz, guarantees a stable operation synchronized with a resonance current, and prevents a leading phase that causes switching loss. .

[0006]

The object of the present invention is to provide, in a resonance type power converter, charging voltage means for charging in accordance with a current flowing through each voltage-driven semiconductor element having a function of not blocking a reverse current. Driving means for applying a control voltage to each control terminal of the driving type semiconductor element at an arbitrary timing, and comparing means for comparing the charging voltage with a reference voltage, wherein the control voltage is controlled based on an output of the comparing means. The problem is solved by applying the voltage to the voltage-driven semiconductor device and turning the device on and off. Further, in the resonance type power converter, charging voltage means for charging according to a current flowing through each voltage driven semiconductor element having a function of not blocking a reverse current is provided, and each control terminal of the voltage driven type semiconductor element is provided at each control terminal. A drive unit for applying a control voltage at an arbitrary timing is provided, and the charge voltage is superimposed on the control voltage, and the voltage-driven semiconductor element is turned on and off based on the magnitude of the charge voltage, Will be resolved.

According to the present invention, the voltage-driven semiconductor element of the inverter is turned on in synchronization with the resonance current by the charging voltage means.
Turn off. In addition, both the high-potential side element and the low-potential side element are turned on when a current flows through the provided freewheeling diode, and conversely, turned off when a current flows through the element. This guarantees that there is no current flowing through the high-potential-side freewheeling diode immediately before the low-potential-side element turns on.It does not operate in the leading phase, but always operates in the lagging phase. Will do. As a result, the reverse recovery current of the diode does not flow, the switching loss can be reduced, and the stable operation of the inverter synchronized with the resonance current can be guaranteed. In addition, by applying the present invention to the lighting devices of the electrodeless lamp and the fluorescent lamp, a dimming function can be newly added, and the short-circuit current when the arm is short-circuited is automatically narrowed down.
In addition, as a protection function for avoiding abnormalities at the end of the lamp life, the operation of the resonance type power converter is stopped, so that the device can be protected. In addition, by applying the present invention to a high-efficiency converter for suppressing harmonics, it is possible to suppress harmonics of a current input from an AC power supply, and furthermore, to protect an arm when short-circuited.

[0008]

Embodiments of the present invention will be described below with reference to the drawings. FIG. 1 is a configuration diagram showing one embodiment of a resonance type power converter according to the present invention. In FIG.
Bridge-connected Q1 and Q2 are power MOSFETs
A drain terminal for inputting a current, a source terminal for outputting a current, and a gate terminal to which a control voltage is applied or removed, and a current flowing between the drain and the source by applying or removing a control voltage to or from the gate terminal. Flow through or shut off. The MOSFET incorporates a diode in the direction from the source terminal to the drain terminal. Hereinafter, the diode incorporated in Q1 is referred to as QD1, and the diode incorporated in Q2 is referred to as QD2. The drain terminal of Q1 is connected to the positive electrode of the power supply 15, the capacitor C1 is connected between the source terminal of Q1 and the drain terminal of Q2, and the connection between C1 and Q2 is the output terminal O of the inverter. A capacitor C2 is connected between the source terminal of Q2 and the negative electrode of the power supply 15, and a connection point between the capacitor C2 and the negative electrode of the power supply 15 is set to a common potential, which is called N. A resonance reactor Lr and a resonance capacitor Cr are connected in series between the output terminal O and the common potential N, and Cr is provided with a load resistor R in parallel.

Next, the inverter control circuit 16 will be described. First, the driving circuit of Q1 provides a driving power supply 13 with the output point O as a reference potential, and the elements 1 and 2 having their control terminals connected in common between the positive and negative electrodes of the driving power supply 13.
The output is taken out from the connection point of the elements 1 and 2 and connected to the gate of Q1. The complementary switch means is a CMOS inverter or similar means. When the element 1 is turned on (at this time, 2 is off), a current for applying a control voltage is applied to the gate terminal of Q1, and when the element 2 is turned on, (Time 1 is off), and a current for discharging the charged electric charge is applied to the gate terminal of Q1.
A signal is supplied from the NAND circuit 7 to the control terminal of the complementary switch means composed of the elements 1 and 2. The NAND circuit 7
A signal obtained by comparing the voltage of C1 with the reference voltage 11 by the comparator 9 and a signal of the level shift means 5 are input. The comparator 9 preferably has a characteristic having a hysteresis. The reference voltage for changing the output of the comparator 9 from Low to High is V _LH , and the reference voltage for changing the output from High to Low is V _HL . Here, the voltage of C1 is the reference voltage 11
When the voltage is lower than (V _LH ), the output of the comparator 9 is set to be High, and the relation of V _HL > V _LH is set. The level shift means 5 is a means for converting a signal based on the common potential N of the lower arm into a signal based on the output O of the upper arm, and activates the voltage-driven semiconductor element of the inverter at a predetermined frequency during startup. Generates a signal to turn on and off.
Next, the driving circuit of Q2 provides a driving power supply 14 with the common N as a reference potential, and a complementary switch composed of elements 3 and 4 whose control terminals are commonly connected between the positive and negative electrodes of the driving power supply 14. A means is provided to take out the output from the connection point of the elements 3 and 4 and connect it to the gate of Q2. A signal is supplied from the NAND circuit 8 to the control terminal of the complementary switch means constituted by the elements 3 and 4. The NAND circuit 8 includes a comparator 10
A signal obtained by comparing the voltage of C2 with the reference voltage 12,
A signal from the start / stop means 6 is input. The comparator 10 has hysteresis similarly to the above-described comparator 9, and when the voltage of C2 is lower than the reference voltage 12 (V _LH ), the output of the comparator 10 becomes High in the same manner as the comparator 9. _Yes , V _HL
The same applies to the relationship> V _LH . The activation stopping means 6 has a function of issuing a trigger signal when the activation is stopped.

The operation of the start / stop means 6 and the operation of the entire resonance circuit will be described with reference to FIGS. (1) Operation at Start-Up When the signal S is input, the start-stop unit 6 outputs a signal of a predetermined frequency fs, as shown in FIG. here,
The frequency fs is the resonance frequency f determined by Lr and Cr in FIG.
Choose higher than o. Vo represents the voltage at the output terminal O, and IL represents the resonance current. The start-stop unit 6 supplies the level shift unit 5 with signals L and H obtained by inverting High and Low of the signal of the frequency fs at the same time. With these signals, the inverter including Q1 and Q2 starts to start with a delay phase 遅 れ with respect to the resonance frequency fo. That is, C
The voltages charged in the capacitors 1 and C2 are represented by VC, respectively.
1, VC2, first, a signal is applied to the gate of Q1 by the H signal of the level shift means 5, Q1 is turned on, C1 is charged by the current flowing through Q1, and VC1 is charged.
Increases. If the frequency fs at the time of startup is set higher than the frequency f at the time of steady operation, C
The voltage charged to 1 does not reach the reference voltage V _HL of the comparator 9,
The ON period of Q1 is determined by the drive signal transmitted by the level shift means 5. Next, L of the level shift means 5
When the signal Q1 is turned off by the signal and the control signal is applied to the gate Q2 by the H signal of the start / stop means 6,
The resonance current IL is the feedback diode QD2 built in Q2.
And the voltage VC charged to C2 in the previous cycle
Decrease 2 When the polarity of the resonance current IL changes, the current flows through Q2, C2 charges this current, and VC2 increases. However, as described above, the frequency fs is set to be higher than the frequency f at the time of steady operation. because you are, VC2 does not reach the reference voltage V _HL of the comparator 10 during Q2 of the on period. Q2
Is determined by the drive signal transmitted by the start / stop means 6, after which Q2 is turned off and a control voltage is applied to Q1 via the level shift means 5. Resonant current I
L flows through the freewheeling diode QD1 built into Q1, and reduces the voltage VC1 charged to C1 in the previous cycle. When the polarity of the resonance current IL changes, the current flows through Q1. The above is the operation of one cycle, and Q1 and Q2 switch according to the frequency fs supplied by the start / stop means 6. (2) Transition from startup to steady operation Transition to steady operation is performed by gradually lowering the frequency fs. The following describes an operation assuming that the frequency fs of the signal output from the start / stop unit 6 is slightly lower than the frequency f during the steady operation. FIG. 3 is an explanatory diagram illustrating the following operation modes (a) to (d). Mode (a) : When Q1 is on, the voltage VC1 charged to C1 by the current flowing through Q1 is equal to the reference voltage V of the comparator 9.
_{When the} signal reaches _HL , the output of the comparator 9 changes to Low, the output of the NAND circuit 7 becomes High regardless of the output of the level shift means 5, and the output of the complementary switch composed of the elements 1 and 2 becomes Low. And the gate voltage of Q1 is
And Q1 turns off. Mode (b) : By turning off Q1, the resonance current IL becomes Q
2 to reduce the voltage VC2 charged to C2 in the previous cycle. The period during which the current flows through QD2 is longer than at the time of startup, and VC2 continues to decrease. When the current drops below V _LH , the output of comparator 10 becomes High.
And if the signal of the start / stop means 6 is H, NAN
The output of the D circuit 8 becomes High, and Q2 turns on. Note that as long as the current continues to flow through QD2, VC2 decreases,
Although the battery is charged to the opposite polarity, the output of the comparator 10 is still High. Mode (c) : When the polarity of the resonance current IL changes, the current flows through Q2, and when the voltage VC2 of C2 reaches the reference voltage _VHL of the comparator 10, the output of the comparator 10 changes to Low,
Regardless of the output of the start / stop means 6, the output of the NAND circuit 8 becomes High and Q2 is turned off. Mode (d) : By turning off Q2, the resonance current IL becomes Q
1 through the freewheeling diode QD1 to reduce the voltage VC1 charged to C1 in the previous cycle. The period during which a current flows through QD1 is long, and VC1 continues to decrease. When VC1 falls below V _LH , the output of comparator 9 changes to High. If the signal of level shift means 5 is H, the output of NAND circuit 7 is High, Q1 turns on. As long as the current continues to flow through QD1, Vc1 decreases and charges to the opposite polarity. Here, if the capacitances of the capacitors C1 and C2 are several tens times or more larger than the resonance capacitor Cr, C
Since the combined value of 1, C2 is almost equal to Cr, there is almost no influence on the resonance current. Also, C1 and C
2 is 8 times or more larger than the gate-source capacitance Cgs of Q1 and Q2, the voltage of the driving power supplies 13 and 14 is divided by C1 and Cgs or C2 and Cgs even if it is divided.
There is no problem because a sufficient control voltage is applied to gs.

As described above, in this embodiment, C1, C2
The charging voltage of C1 and C2 is reduced by the current flowing through the freewheel diode provided in each of the elements Q1 and Q2. When the charging voltage becomes lower than the reference value, Q1 and Q2
Q2 is turned on, and conversely, a current flows through the element, and the charging voltage of C1 and C2 increases. When the charging voltage becomes higher than the reference value, Q1 and Q2 are turned off. As a result, the voltage-driven semiconductor devices Q1 and Q2 of the inverter are turned on and off in synchronization with the resonance current IL. This is because Q1 (or Q2) is turned on immediately before Q2 (or Q1) is turned on. ) Guarantees that there is no current flowing through the freewheeling diode, and does not operate in the leading phase, which means that it always becomes the lagging phase ψ. Thereby, in the present embodiment, the reverse recovery current of the diode does not flow,
Switching loss can be reduced.

FIG. 4 shows another embodiment of the present invention.
(A) is a configuration diagram of the resonance type power converter according to the present embodiment, and (b) is an operation diagram thereof. This embodiment eliminates the comparator for comparing the voltages of C1 and C2 shown in the embodiment of FIG. 1 with the reference value _VLH or _VHL, and differs from the control unit 6 in that the start-stop unit 6 of the embodiment of FIG. In that The control means 6 has a function of transmitting a random control command to Q1 and Q2 in accordance with the signal S without being limited to the sequence at the time of startup described in FIG. Other configurations are shown in FIG.
This is the same as the embodiment. Since voltage-driven devices such as MOSFETs have been miniaturized year by year and the on-resistance of the devices has been reduced, when the gate-source voltage slightly exceeds the threshold voltage, the devices become saturated. Not
A sufficient current can be passed. In the embodiment of FIG. 4A, the voltage values of the driving power supplies 13 and 14 are V
cc, the gate-source voltage of Q1 is (Vc
c-VC1) gate voltage (control voltage) is applied.
As described above, VC1 increases in accordance with the current flowing through Q1, and Q1 turns on and off depending on whether (Vcc-VC1) is higher or lower than the gate threshold voltage Vth of Q1. . The embodiment of FIG.
The feature is that the inverter operation synchronized with the resonance frequency is continued without using a special comparator by utilizing the fact that (Vcc-VC1) increases or decreases according to the resonance current IL.
The same applies to turning on and off of Q2.

The operation will be described with reference to FIG. VC
1, VC2 changes to positive or negative according to the resonance current IL,
1, applied between the gate and source of Q2 with opposite polarity. As a result, (Vcc-VC1) becomes the threshold voltage Vt of Q1.
When it becomes h or less, Q1 turns off and current flows through QD2. FIG. 1 shows that VC2 decreases due to the current of QD2.
When (Vcc-VC2) becomes equal to or higher than the threshold voltage Vth of Q2, Q2 can be turned on. The above operation is repeated in one cycle. That is, when the voltage-driven semiconductor element Q1 on the high potential side of the inverter is turned on and the current flowing into the resonance circuits Lr and Cr is set to the positive direction and the current flows in the positive direction through the element Q1, C1 is charged with this current. And the gate-source voltage of Q1 is (V
cc-VC1) is applied. As a result, when a current flows, the gate voltage of Q1 decreases and the on-resistance of Q1 increases.
Work to turn off. Next, when Q1 is turned off, the resonance current IL flows through the freewheel diode QD2 of the voltage-driven semiconductor element Q2 on the low potential side. This current is a reverse current to Q2, and the charge voltage that C2 was charging in the previous cycle is discharged by this reverse current,
Further, a voltage of the opposite polarity is charged. This voltage superimposes a positive voltage on the gate (control) voltage of Q2, and when this voltage exceeds the gate threshold of Q2, Q
2 goes to the ON state. However, the freewheeling diode QD2
No current flows in Q2 as long as the current flows through Q2. Next, when the polarity of the resonance current changes, the current becomes Q2
This current is charged in C2, and a negative voltage is superimposed on the gate (control) voltage of Q2, thereby reducing the gate (control) voltage of Q2. When a current further flows and the gate (control) voltage of Q2 falls below the gate threshold of Q2, Q2 turns off. As a result, the resonance current I
L flows through the freewheeling diode QD1 of Q1. The reverse current reduces the voltage charged in the previous cycle of C1. Thereafter, the charging voltage changes to the opposite polarity, superimposing a positive voltage on the gate (control) voltage of Q1, and when this voltage exceeds the gate threshold of Q1, Q1 turns on. The above is a one-cycle operation, and the voltage-driven semiconductor elements Q1 and Q2 of the inverter are turned on and off in synchronization with the resonance current IL by C1 and C2. Also, Q1,
Any element of Q2 is provided with a freewheeling diode QD
1. The transistor turns on when a current flows through QD2, and turns off when a current flows through Q1 and Q2. This guarantees that there is no state in which current flows through the high-potential-side freewheeling diode QD1 immediately before the low-potential-side element Q2 is turned on. become. A major difference from the embodiment of FIG. 1 is that the present embodiment keeps elements 1 and 3 of the complementary switch means of the upper and lower arms on, respectively,
Q1 and Q2 are turned on only by the voltage of VC1 and VC2,
The feature is that it is turned off. According to this operation, since Q1 and Q2 are switched without passing through the comparator and the NAND circuit, the time delay is small and the inverter is suitable for a high-frequency resonant inverter of several MHz or more.

As described above, in this embodiment, C1, C2
The charging voltage of C1 and C2 is reduced by the current flowing through the freewheeling diode provided in each of the elements Q1 and Q2, and the gate-source voltage (gate (control) voltage) of Q1 and Q2 based on this charging voltage Is lower than the respective gate threshold voltages, Q1 and Q2 are turned on. Conversely, current flows through the elements, and the charging voltages of C1 and C2 increase, and the gate voltages of Q1 and Q2 based on the charging voltages are reduced. When the voltage becomes higher than each gate threshold voltage, Q1, Q
Turn 2 off. As a result, also in the present embodiment, FIG.
Similarly to the embodiment, the voltage-driven semiconductor elements Q1 and Q2 of the inverter are turned on and off in synchronization with the resonance current IL, and always operate with the delay phase 遅 れ. Therefore, the reverse recovery current of the diode does not flow, and the switching loss can be reduced.

FIG. 5 shows an embodiment in which the present invention is applied to a lighting device for an electrodeless lamp. In FIG. 5, R is an electrodeless lamp. The electrodeless lamp winds a coil in the vicinity of a discharge bulb filled with mercury or amalgam and an inert gas, and flows a high-frequency current to generate a high-frequency magnetic field in the bulb. Generate and turn on the lamp. The feature is that there is no filament as compared with a normal fluorescent lamp.
The inverter for driving this electrodeless lamp has capacitors C1 and C2 on the source side, respectively, as in the configuration shown in FIG.
The control means 16 for driving Q1 and Q2 has the same configuration as that of FIG. Note that Q1 and Q2
May be configured the same as in FIG. 4A. A reactor Lr for resonance and a capacitor Cr are connected in series between the output O of the inverter and the common N,
An electrodeless lamp R and an auxiliary capacitor C4 are connected in series to both ends of Cr. Further, the inverter rectifies an AC current received from the AC power supply 17 through a rectifier circuit including a low-pass filter 18 and diodes D1 to D4, charges the smoothing capacitor CE, and supplies a current from the CE to the inverter.

To efficiently turn on the electrodeless lamp R, it is important to apply an alternating voltage of several MHz or more from the inverter to the resonance circuits Lr and Cr. According to the configuration of FIG. 5, as described in the operation description of FIG.
In this embodiment, the inverter can be driven in synchronization with the inverter, and the lag phase can be guaranteed even with a high-frequency alternating voltage of several MHz or more. Can be turned on.

Although there is no example of a current electrodeless lamp lighting device having a lamp dimming function, like the ordinary fluorescent lamp with a filament, according to the configuration of FIG.
By changing the switching frequency of the inverter,
Dimming is possible. That is, as shown in FIG. 8A, as the switching frequency f of the inverter is increased with respect to the resonance frequency fo, the resonance current decreases with the delay phase. However, for electrodeless lamps,
Since the frequency of the resonance point is several MHz, it is not easy to drive the inverter at a switching frequency that is many times the switching frequency. In particular, in a high-frequency inverter, an arm short-circuit in which Q1 and Q2 are simultaneously turned on is expected, and a countermeasure has been a problem. Here, it is assumed that the control circuit 16 shown in FIG. 1 shown in FIG. 5 is changed to the control circuit 16 shown in FIG. 4A, and the inverter is driven by the control means 6 and the level shift means 5 at a frequency several times higher than that in the normal lighting. The protection processing in the case where an arm short circuit occurs will be described. Q1 and Q2
Are turned on at the same time, Q
1 and Q2 flow, and this current is Q1 or Q2
To the value determined by the saturation current of On the other hand, this short-circuit current always passes through C1 and C2, and C1 and C2 are charged with a larger current than at the time of lighting, and the respective voltage increases faster. Although the saturation current of the voltage-driven element is lower as the gate voltage is smaller, the gate voltages of Q1 and Q2 are represented by (Vcc-VC1) and (Vcc-VC2), respectively, as described above. Conversely, as VC1 and VC2 increase, the gate voltage decreases and the saturation current (ie, short-circuit current) decreases. This phenomenon is a negative feedback effect, and the short-circuit current is automatically reduced by the negative feedback effect. As described above, in the resonance type inverter of the present invention, it is guaranteed that the device is not destroyed due to the short circuit of the arm, and as a result, it can be said that the inverter is suitable also from the viewpoint of protection of the device in the case of the short circuit of the arm.

FIG. 6 shows an embodiment in which the present invention is applied to a lighting device for a fluorescent lamp. In FIG. 6, R is a fluorescent lamp with a filament. In the case of a fluorescent lamp with a filament, the configuration of a resonance circuit including a lamp connected between the output O of the inverter and the common N is slightly different from the configuration of FIG. Other configurations are the same as those of FIG. 5, and have the same effects as the high-frequency electrodeless lamp lighting device of FIG. The control means 16 for driving Q1 and Q2 may have the same configuration as in FIG. In the case of a fluorescent lamp with a filament, emission from one or both filaments may end at the end of life. For example, in FIG. 6, when the emission of both filaments is eliminated, the lamp R becomes equal to the high resistance, and the configuration is such that Lr, Cr and the auxiliary capacitor C4 are connected in series between the output O and the common N. Since the combined frequency of Cr and C4 increases the resonance frequency fo, the conventional lighting device reaches a leading phase state of f <fo. According to the present embodiment, even in such a state at the end of the lamp life, it is possible to suppress the occurrence of loss due to the advanced phase or the overcurrent due to the change of the resonance point. That is, when the resonance frequency fo increases and the resonance current increases due to the combined capacitance of Cr and C4, the time for charging C1 and C2 by the current flowing through Q1 or Q2 also decreases, and the current flows through these elements. The operation of detecting the voltages of C1 and C2 during the flowing period and turning off Q1 or Q2 is maintained. That is, this follows the change in the resonance frequency and maintains the lag phase state. By the way, if it is assumed that the leading phase has been reached, the reverse recovery current flowing through the freewheeling diode on one side flows through C1 and C2, so that VC1, V2
C2 both increase. In the case of the embodiment of FIG. 1, as the reverse recovery current increases, the voltages of VC1 and VC2 exceed the reference value _VHL , and the comparators 9 and 10 operate to turn off Q1 and Q2. In the case of the embodiment of FIG. 4, (Vcc−
(VC1) and (Vcc-VC2) become equal to or lower than the gate threshold voltage, and Q1 and Q2 are turned off. As described above, the inverter of this embodiment stops operating without synchronizing with the cycle of resonance, but this stopping operation is effective as a protection function for avoiding abnormalities at the end of lamp life.

FIG. 7 shows an embodiment in which the present invention is applied to a high-efficiency converter for suppressing harmonics. FIG. 7A shows the configuration. In recent years, as regulations on power supply harmonics are enforced, many high-efficiency converters for suppressing harmonics have been reported. One of them is a converter called a current interrupting type, which rectifies an AC current received from an AC power supply 17 through a rectifying circuit using a low-pass filter 18 and diodes D1 to D4, and supplies the rectified current to a choke coil Lr. In a normal current interrupt type converter, the current flowing through Lr is chopped to a high frequency by a power semiconductor switch element, and the energy stored in Lr is supplied to a power supply capacitor CE. L
Since the amplitude of the current flowing through r changes according to the AC voltage, the waveform of the high-frequency current passed through the low-pass filter 18 has a shape close to a sine wave. In this embodiment, a diode D is connected between the high potential side of the rectifier circuit and the output O of the inverter.
5, a choke coil Lr and a resonance capacitor Cr are connected in series, and a diode D6 is provided in such a direction that the connection point of D5 and Lr is on the anode side and the high potential side of the power supply capacitor CE is on the cathode side. Efficiency converter. The configuration of the inverter and the control means 16 for driving Q1 and Q2 are the same as those in FIG. R provided at both ends of the CE is a load. The control means 16 for driving Q1 and Q2 may have the same configuration as in FIG.

FIGS. 7B and 7C show operation modes of the present embodiment. In FIG. 7B, when Q2 is turned on, the current is supplied from the rectifier circuit to D5, Lr, Cr, Q2, C2.
Flows through the route that returns to the AC side. This current is a resonance current whose resonance frequency is determined by Lr and Cr, has a sinusoidal waveform, and its amplitude is proportional to the amplitude of the AC power supply 17. This current causes the voltage VC2 of C2 to increase, and its value becomes V2
Q2 turns off when _HL is exceeded. The current flowing through Lr returns to the AC side through the freewheeling diodes QD1 and CE of Q1, as shown by the broken line in (b), and charges CE.
This current causes the voltage VC1 of C1 charged in the previous cycle to decrease, and when the value falls below V _LH , Q1 can be turned on. Next, in FIG. 7C, the polarity of the resonance current changes, and the voltage charged in Cr in FIG. 7B is discharged, so that current flows through the paths of Cr, Lr, D6, Q1, and C1, This current increases the voltage on C1, and when its value exceeds _VHL , Q1 turns off. And
As shown by the broken line in (c), the current is Cr, Lr, D6,
The path changes through CE, C2, and QD2.
By reducing C2, Q2 can be turned on. This behavior is
As in the embodiment of FIG. 1, the difference is that the amplitude of the resonance current is proportional to the amplitude of the AC power supply 17. This effect appears during the ON periods of Q1 and Q2. That is, when the amplitude of the AC power supply 17 is low, the current is small, so that VC1 and VC2
There a longer time to reach V _HL, the ON period of Q1 and Q2 is increased. Conversely, larger current when the amplitude of the AC power supply 17 is high, VC1 and VC2 is less time to reach V _HL, the ON period of Q1 and Q2 is reduced. in this case,
Q1 and Q2 are automatically pulse width modulated according to the sine wave of the AC power supply 17, and the current input from the AC power supply 17 has a sine wave compared to the current that has been chipped by the normal current interrupt type converter. To reduce harmonics. As described above, in the present embodiment, by using the resonance type inverter provided with C1 and C2, harmonics of the current input from the AC power supply 17 can be suppressed.
As described in the embodiment of FIG.
1. Since the reverse recovery of QD2 does not occur, the switching loss is reduced and, as described in the embodiment of FIG.

In this embodiment, the diode D5,
Although the choke coil Lr and the resonance capacitor Cr are connected in series between the high potential side of the rectifier circuit and the output O of the inverter, they are connected in series between the low potential side of the rectifier circuit and the output O of the inverter. You may.

[0022]

As described above, according to the present invention,
In a high-frequency resonance type power converter, since the lead phase is prevented and the phase is always operated in the lag phase, the reverse recovery current of the diode does not flow, reducing the switching loss and stabilizing the inverter synchronized with the resonance current. Operation can be guaranteed. In addition, in a lighting device for an electrodeless lamp operating at several MHz or more, a new dimming function can be added, and switching loss is reduced.
The electrodeless lamp R can be efficiently turned on, and the short-circuit current at the time of arm short-circuit can be automatically narrowed to protect the device. Further, in a normal fluorescent lamp lighting device, even if the resonance frequency changes at the end of the lamp life, the delay phase can be maintained by following the resonance frequency, and protection for avoiding abnormalities at the end of the lamp life. As a function, the operation of the resonance type power converter can be stopped to realize protection. Moreover, in the high-efficiency converter for suppressing harmonics, it is possible to suppress the harmonics of the current input from the AC power supply, to reduce the switching loss, and to provide protection in the event of an arm short-circuit.

[Brief description of the drawings]

FIG. 1 is a configuration diagram of a resonance type power converter according to an embodiment of the present invention.

FIG. 2 is an operation sequence of the embodiment of FIG. 1;

FIG. 3 is an operation mode of the embodiment of FIG. 1;

FIG. 4 is a configuration diagram of a resonance type power conversion device according to another embodiment of the present invention.

FIG. 5 is a configuration diagram of an electrodeless lamp lighting device using the present invention.

FIG. 6 is a configuration diagram of a fluorescent lamp lighting device using the present invention.

FIG. 7 is a configuration diagram of a high-efficiency converter for suppressing harmonics using the present invention.

FIG. 8 is an explanatory diagram of a resonance type power converter.

[Explanation of symbols]

Q1, Q2 Power MOSFETs D1 to D6 Diodes C1 to C4, Cr, CE Capacitor Lr Resonance reactor R Load, electrodeless lamp or fluorescent lamp 1-4 Semiconductor switch element 5 ... Level shift means 6... Start-stop means (control means) 7 and 8... NAND circuits 9 and 10... Voltage comparators 11 and 12. Control means 17: AC power supply 18, Low-pass filter

──────────────────────────────────────────────────続 き Continued on the front page (51) Int.Cl. ⁶ Identification code Agency reference number FI Technical display location H05B 41/24 H05B 41/24 M

Claims (9)

[Claims] 1. An inverter having a plurality of voltage-driven semiconductor elements having a function of not blocking a reverse current, and a resonance type including an inductor and a capacitor on an output side of the inverter, and supplying an alternating current. In the power conversion device, charging voltage means for charging according to a current flowing through each of the voltage-driven semiconductor elements is provided, and a drive voltage for applying a control voltage to each control terminal of the voltage-driven semiconductor element at an arbitrary timing. Means for comparing the charging voltage with a reference voltage, applying the control voltage to the voltage-driven semiconductor element based on the output of the comparing means, and turning the element on and off. Resonance type power converter. 2. The voltage-driven semiconductor device according to claim 1, wherein the voltage-driven semiconductor device is turned on when the charging voltage according to a current flowing in the reverse direction through the voltage-driven semiconductor device becomes equal to or less than a first reference value. A resonance-type power converter, wherein the voltage-driven semiconductor device is turned off when the charging voltage corresponding to a current flowing in the voltage-driven semiconductor device in a forward direction becomes equal to or higher than a second reference value. 3. A resonant type for supplying an AC current, comprising: an inverter having a plurality of voltage-driven semiconductor elements having a function of not blocking a reverse current; and a resonance means including a reactor and a capacitor on the output side of the inverter. In the power conversion device, charging voltage means for charging according to a current flowing through each of the voltage-driven semiconductor elements is provided, and a drive voltage for applying a control voltage to each control terminal of the voltage-driven semiconductor element at an arbitrary timing. Means for superimposing the charging voltage on the control voltage, and turning on and off the voltage-driven semiconductor element based on the magnitude of the charging voltage. 4. The voltage-driven semiconductor device according to claim 3, wherein the voltage-driven semiconductor device is turned on when the charging voltage corresponding to a current flowing through the voltage-driven semiconductor device in a reverse direction is equal to or lower than a predetermined value. A resonance-type power converter, wherein the voltage-driven semiconductor element is turned off when the charging voltage according to a current flowing through the element in a forward direction is equal to or higher than a predetermined value. 5. The voltage-driven semiconductor device according to claim 1, wherein the drive unit applies a control voltage to turn on and off the inverter at a preset frequency at the time of startup to each of the voltage-driven semiconductor elements. A resonance-type power converter, wherein a constant control voltage is applied to each of the voltage-driven semiconductor elements in a steady state when the voltage of the charging voltage means reaches a predetermined voltage. 6. The method according to claim 1, wherein when a current passing through the inverter is generated, at least one of the voltages is based on a voltage of the charging voltage means that increases according to the current. A resonance type power converter, wherein a drive type semiconductor element is turned off. 7. The resonance type power converter according to claim 1, wherein an electrodeless lamp or a fluorescent lamp is connected to the resonance means in series or in parallel with one of the resonance means. 8. A converter for rectifying an AC voltage, an inverter including a voltage-driven semiconductor element having a function of preventing a reverse current, and an output between the inverter and one of a high-voltage terminal and a low-voltage terminal of the converter. Resonance means including at least the reactor and the capacitor provided, and a smoothing capacitor for smoothing the current of the resonance means,
Driving means for applying a control voltage to each control terminal of the voltage-driven semiconductor element at an arbitrary timing, charging voltage means for charging in accordance with a current flowing through each of the voltage-driven semiconductor elements, And a comparison unit for comparing the reference voltage with the reference voltage, and applying the control voltage to the voltage-driven semiconductor device based on an output of the comparison unit to turn on and off the device. apparatus. 9. A converter for rectifying an AC voltage, an inverter including a voltage-driven semiconductor element having a function of preventing a reverse current, and an output between the inverter and one of a high-voltage terminal and a low-voltage terminal of the converter. Resonance means including at least the reactor and the capacitor provided, and a smoothing capacitor for smoothing the current of the resonance means,
A driving unit that applies a control voltage to each control terminal of the voltage-driven semiconductor element at an arbitrary timing, and a charging voltage corresponding to a current flowing through each of the voltage-driven semiconductor elements to the control voltage of the element. A resonance-type power converter, comprising: charging voltage means for superimposing, and turning on and off the voltage-driven semiconductor element based on the charging voltage.