JPH10336536A - Highly integrated television tuner on single microcircuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a relatively inexpensive television tuner which requires small space in order to be used on a printed circuit board by selecting an architecture, which raises and converts an RF input signal into a higher internal frequency and minimizing filtering at the time of input to a receiver. SOLUTION: A first IF signal of 1200 MHz is mixed with a fixed 1180 MHz reference output of a second local oscillator (LO2) 412 and generates a second IF at a mixer 2(MIX2). A first IF filter(FIFF) 509 between a mixer 1(MIX1) 408 of an RF converter 110 and the MIX2 510 filters and deletes all the frequency components of an image which becomes 1160 MHz and the MIX2 510 is made to be a basic mixer similar to the MIX1 408 rather than an image exclusion mixer. The second IF signal is filtered by a second IF filter(SIFF) 420 for the final video band width.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to television tuner circuits, and more particularly, to highly integrated television tuners assembled in a single microcircuit device.

[0002]

BACKGROUND OF THE INVENTION One of the most important costs in television manufacture is the cost of a tuner. The typical cost of a television (TV) tuner is about $ 15,000,
And that is significant for the overall cost of the television receiver. One solution to reduce tuner costs is to reduce the number of tuners.

[0003] Traditionally, tuners are composed of two basic elements. The first element is to convert high frequency to intermediate frequency (RF to IF). Then the second element is
Convert IF to baseband. TV tuners were originally designed for receiving television broadcasts in television receivers, which are essentially stand-alone units containing cathode ray tubes. As such, TV tuners were originally integrated components embedded in single purpose devices.

[0004] Currently, however, state-of-the-art consumer electronic devices use TV tuners that are not an integral part of television receivers. The tuner is a separation element connected at some point to the cathode ray tube, but the tuner is not an integral part of the monitor. For example, a TV tuner can be assembled on a circuit board and then attached to a personal computer (PC) system, thereby making the PC function as a television receiver. These tuners convert radio frequency television signals into baseband (or low frequency) video signals, which can then be transmitted to other elements in the PC for video processing applications.

[0005] Circuit elements that perform RF-to-IF conversion typically consist of one or two integrated circuits and a number of discrete elements—inductors, capacitors, and / or transistors. IF-to-baseband conversion typically involves separate integrated circuits, several filter elements such as ceramic filters and SAW filters, a series of tuning and control elements such as resistors and potentiometers, variable inductors and / or Or include a capacitor, and some additional external components. Thus, the complexity of the tuner is quite high, and there are typically 100 to 200 elements on the circuit board. Furthermore, state-of-the-art TV tuners still require that each tuner be adjusted by manual tuning before leaving the factory. This manual tuning is one of the most expensive costs associated with the manufacturing process and is an important factor in tuner cost.

[0006] Past broadcast television tuners have evolved for over 60 years. The earliest tuners utilized tube technology and required the use of a minimum number of tubes due to their cost, power consumption and size. Therefore, passive components such as resistors, capacitors, inductors and transistors were used as much as possible in most designs. This design style is T
V tuner elements, especially vacuum tubes, are bipolar and MO
It continued until around 1960 when it was replaced by S transistors. However, the total number of active elements has continued to limit the cost and size of TV tuners and minimize the total number of active elements.

In the early 1970's, integrated circuits became practical as a component in television tuners, and their design techniques changed dramatically. Many features of tuners that use only one tube or transistor are
Replaced with 20 individual transistors, which would be able to perform the same function with good accuracy, small space, low power, low heat generation and low cost. The introduction of integrated circuits is gentle,
Initially there were only low frequency components, and then extended to high frequency active components. Nevertheless, many active components outside the integrated circuit remained in the TV tuner design.

[0008] With one advance, the SAW (surface acoustic wave) filter, there is considerable variation, some manually tuned inductors and capacitors have been removed from the tuner, and receive filter performance has been reduced in much smaller spaces and at lower cost. Could be reduced and improved.
However, SAW filters assembled on ceramic substrates cannot be integrated on silicon wafers with the rest of the active circuitry, and therefore must remain discrete components in the final design. 19
The trend in the eighties was to reduce all of the passive components and simplify their associated manual tuning at the factory. Recently, TV tuners have been reduced from a very large case of about 2 "x 5" x 1 "
Reduced size to a much smaller case of 2 "x 2" x 3/8 ". TV tuners are used in increasingly smaller computers, television receivers, and VCRs, so smaller sizes are highly respected The size of the TV tuner must also decrease as the equipment in which the tuner is used becomes smaller, and as the size of the tuner decreases and as the tuner is used in a variety of devices, the cost becomes more significant. And should not be a significant part of the end product cost.
When a tuner is used in a television receiver, the size of the tuner is less important because the television receiver has an inherently large volume. However, when the tuner is used in other electronic devices, the space becomes larger and the physical space of the tuner becomes significant.

Before describing the monolithic television tuner of the present invention, it will be useful to describe a state-of-the-art television tuner found in the prior art.

Although there has been a theoretical proposal to integrate a TV tuner in a single microcircuit, no realization has been known. Next, the best known prior art, although very compact, is not a fully integrated tuner as shown in FIG.

FIG. 3 shows a functional electric block diagram of a current state-of-the-art TV tuner configuration. The television tuner 300 is configured in a single metal shield assembly that includes a printed circuit board, and on which all of the associated tuner elements are mounted,
And are electrically connected. TV tuner 300
Is designed to be a module that attaches to another printed circuit board to connect input and output signals directly to the appropriate terminations in a television receiving system.
The metal shield is used to transmit unwanted external signals to the TV tuner 3.
00 and to prevent TV tuner 3
00 is used to prevent emission of signals that interfere with the operation of the external device.

The prior art TV tuner 300 has three
It consists of two integrated circuits, a preamplifier and mixer 305, an IF and baseband signal processor 310, and a frequency synthesizer and inter integrated circuit (IIC or I ² C) bus interface 315. The television tuner 300 also includes a bandpass and image reject notch filter 304, a bandpass and image reject notch filter 312, a surface acoustic wave (SAW) filter 316,
It comprises a plurality of individual elements including a video carrier filter 324 and an audio carrier phase shifter 360. The television tuner 300 receives a standard television RF signal from an antenna 302 or cable system connection (not shown) through a bandpass and image reject notch filter 304. The bandpass and image reject notch filter 304 controls the TV tuner 300 so that a minimum number of unwanted signals are present in the TV tuner 300.
Restrict the signal entering. Therefore, the filter 304 limits the image response generated by the first mixer, as described below. Filter 304 also attenuates signals outside the fairly narrow (100 MHz) range around the desired signal. Finally, known interfering signals, such as FM broadcasts, shortwave service signals, signals in the intermediate frequency band, and civil band radio signals are specifically rejected by filter 304.

The preamplifier 306 of the preamplifier and mixer 305 receives the output of the bandpass and image rejection notch filter 304 and minimizes the noise level increase (typically 8-10 dB), and Increase the level (10 dB). The gain of the preamplifier 306 is controlled by an automatic gain control (AGC) 338 so that when a very strong signal enters the TV tuner 300, the overall gain is reduced and the distortion in the amplifier is less than without the gain reduction. Become smaller.

The output of preamplifier 306 is sent to bandpass and image reject notch filter 312 with the same basic requirements to minimize the passage of potential interfering signals. The filter 312 is external to the preamplifier and mixer 305 and is made up of a plurality of discrete elements including capacitors, inductors and varactor diodes.

The output of the bandpass and image rejection notch filter 312 is sent back to a mixer 308 in the amplifier and mixer 305. The mixer 308 outputs the output of the filter 312 to the local oscillator and the frequency synthesizer 3
It mixes with the output of 42 and has a selected frequency higher than the desired receiver carrier by 45.75 MHz. Therefore, the output of mixer 308 is 45.7
5 MHz. Also, 91.5 MH is higher than the input frequency.
Above z is the image signal from mixer 308, which is removed by filter 304 and filter 312. Therefore, when the frequency of the frequency synthesizer 342 is tuned to receive signals at different carrier frequencies, the bandpass and image rejection filters 304 and 312 also properly pass only the desired signal and filter the mixer image. Must be tuned not to pass.

The frequency synthesizer 342 receives an input frequency reference signal (usually 16 bits) and outputs status signals "Automatic Frequency Control (AFC) Error" and "Frequency (FREQ) Lock". In addition, the tuned signal used by the voltage controlled oscillator (VCO) in frequency synthesizer 342 is transmitted from frequency synthesizer 342 to bandpass and image reject notch filter 304.
And 312. The local oscillator signal is output from frequency synthesizer 342 to mixer 308.

Then, 45.75 MH of mixer 308
The z output signal is passed through a SAW (surface acoustic wave) filter 316, which limits the bandwidth of the signal to only one (1) channel (6 MHz for the NTSC standard) and A frequency linear attenuation known as the Nyquist slope of Linear attenuation by the SAW filter 316 converts the signal from the vestigial sideband signal to one equivalent to a single sideband with carriers, so that the frequency response of the demodulated signal is flat across the video bandwidth. It is. SAW filter 316 is very lossy (on the order of 25 dB), so the input to SAW filter 316 is amplified by a preamplifier (not shown) by a corresponding value to minimize noise effects.

The output of SAW filter 316 is input to an intermediate frequency (IF) amplifier 320 in IF and baseband signal processor 310. IF amplifier 320
Provides most of the total gain of the TV tuner 300,
And, it receives the gain control from the AGC 338.

The output of the IF amplifier 320 is the video detector 3
22 and also off chip to an external video carrier filter 324. This is the stage where video demodulation is performed. Video detector 322 is substantially
A mixer having a local oscillator input connected to the output of the video carrier filter 324 through a carrier amplitude limiter 326. The output of carrier limiter 326 is
In-phase representation of the video carrier signal without any modulation. The output of carrier limiter 326 is received by video detector 322, which mixes the output of carrier limiter 326 with the output of IF amplifier 320.

AFC frequency discriminator 340 is used in prior art devices to detect the difference between the carrier frequency contained in the output of carrier limiter 326 and a known legal carrier frequency reference. The output signal of AFC frequency discriminator 340 is an error signal used to drive frequency synthesizer 342 in a direction that reduces errors between the output of carrier limiter 326 and a known legal carrier frequency reference. The output of video detector 322 is a baseband video signal that is combined with some high frequency mixing components. These mixed components are
It is removed by the video baseband filter 330. The output of the video baseband filter 330 is directed to a synchronization pulse clamp (sync clamp) 332, which sets the level of the synchronization pulse to a standard level.

Next, the output of sync clamp 332 is sent to a noise inverter 334, which removes large noise spikes from this signal. The output of noise inverter 334 is sent to video buffer 336, which is configured to drive a fairly high circuit board impedance of approximately 1000-2000 ohms. The output of the noise inverter 334 is also sent to an AGC (Automatic Gain Control) 338, which compares the level of the synchronization pulse with the signal blanking level, measures the incoming signal strength, and at the final output A gain control signal that is used by the IF amplifier 320 and the RF preamplifier 306 to dynamically adjust the gain of the TV tuner 300 to correct the level of the signal.

The audio signal is an FM signal that passes through the video detector 322 and follows the same electrical path as the video. At the output of the video detector 322, the audio signal appears as a 4.5 MHz subcarrier due to the fact that the audio signal enters the prior art TV tuner 300 at a frequency 4.5 MHz higher than the desired video carrier. The audio subcarrier is passed through an FM quadrature demodulator. The FM quadrature demodulator comprises a mixer, an audio second detector 350, a 90 ° (at 4.5 MHz) phase shifter, and an audio carrier phase shifter 360. The output of the audio second detector 350 is a baseband audio signal, which is filtered by a low pass (30 kHz) filter 352 to remove any unwanted high frequency components. The output of the low pass filter 352 is finally passed over an audio buffer 354, which drives an audio amplifier that finally drives a speaker. Serial digital interface 34
4 receives the "serial data" and "serial clock" inputs and provides control and update status for the prior art television receiver.

The baseband and image reject notch filters 304 and 312 typically consist of a plurality of capacitors, inductors, and varactor diodes. Video carrier filter 324 typically consists of three discrete elements, one inductor and two capacitors. Similarly, audio carrier phase shifter 360 also comprises one inductor and two capacitors. In addition to the circuit elements shown as discrete elements in addition to the circuit elements 305, 310 and 315 of FIG. 3, other discrete elements (not shown) are provided for IF and baseband signal processor 310 and frequency Synthesizer 34
2 are connected. Frequency synthesizer 342 typically has several external capacitors, inductors and / or
Or tuned by a varactor diode. Video buffer 336 and audio buffer 354 also typically use external discrete elements such as resistors, capacitors and / or transistors. Video baseband filter 330 and low-pass filter 352 also
External inductors and capacitors can be used.

[0024]

SUMMARY OF THE INVENTION Accordingly, one object of the present invention is to provide a relatively low cost and small physical space TV for use on a printed circuit board.
To provide a tuner.

Another object of the present invention is to reduce the number of external components required at the same time, thereby reducing the complexity of the printed circuit board and the circuit board area required by the TV tuner. It is to provide a TV tuner that meets or exceeds the performance of a state-of-the-art TV tuner.

Yet another object of the present invention is to provide a TV tuner with a serial bus so that it can be controlled by a microcontroller embedded in a television set, personal computer, or other video device.
The purpose is to enable computer control of the V tuner.

Yet another object of the present invention is to provide a TV tuner that adapts computer controlled output impedance characteristics to different load specifications.

[0028]

SUMMARY OF THE INVENTION These and other problems include:
The problem was solved by integrating a TV tuner that takes a wideband frequency as input and performs video demodulation over a very narrow bandwidth. To achieve this, one architecture was chosen that up-converts the RF input signal to a higher internal frequency, and this allows the present invention to minimize filtering at the input stage of the receiver. did.
Therefore, the present invention, except for a single fixed tuned filter,
It can operate without tunable input filtering. This eliminates the need for a precision controlled tunable filter that needs to be adjusted mechanically during manufacturing and is subject to performance variations due to age, temperature, humidity, vibration and power performance. This, along with complexity, was a serious deficiency of conventional tuners that had to be eliminated because it was a source of significant error and distortion.

Further, by removing the input of the high-frequency signal to the integrated circuit of the present invention or the output therefrom,
It is advantageous to perform up-conversion on-chip, as driveability issues associated with high frequency signals and noise coupling issues associated with integrated circuit external interconnects are avoided. The present invention then down-converts from high frequency using an image rejection mixing scheme that provides a tightly controlled down-conversion with little added distortion. This also further minimizes the on-chip filtering effect by minimizing the amount of filtering that must be done at high frequencies. As a result, the high frequency operation of the present invention is limited to the initial stage inside the chip and is performed at very low impedance levels to minimize distortion. After the down conversion, all remaining filtering is performed with a continuous-time mode filter implemented in an integrated circuit.

Another technical advantage of the present invention is that manual tuning of the final design is not required. As in the prior art, no external elements are required to perform the adjustment.

Yet another technical advantage of the present invention is that the output impedance can be selected based on system requirements, thereby minimizing power consumption for a particular application. The invention can drive the coaxial cable with a low output impedance driver, or for board level applications, the invention can be configured to drive a high output impedance. The present invention can also match very accurate load impedances defined by a single external element.

The present invention advantageously utilizes much less board space than prior designs (on the order of 5% to 10% of prior art designs) and consumes only about 1/5 of the power. The present invention also advantageously operates at a single voltage level, versus two or three voltage levels in conventional designs.

Another technical advantage of the present invention is that the need for a metal case is reduced. The accumulation itself allows for sufficient shielding to meet interference criteria. The monolithic television (MTV) tuner embodied in the present invention is intended to replace the TV tuner module currently used in most television broadcast receivers. The MTV tuner integrates all of the tuner's functional elements, except for the crystal frequency reference and the power supply bypass capacitor. The level of integration of the present invention dramatically reduces the cost of a basic TV tuner and increases its manufacturability and reliability. The TV tuner of the present invention
Externally controlled by a computer or a controller via a digital serial bus interface (I ² C). The preferred embodiment of the present invention provides an antenna input that can be connected directly to a standard coaxial cable, thereby enabling both antenna and cable television applications. Further, baseband video and audio outputs are designed for high or low impedance applications. The high impedance mode is a driver for short interconnects on printed circuits and offers the benefit of low operating power. Low impedance mode is
Drives an industry standard studio cable interface that requires a larger power supply.

The preferred embodiment of the present invention is designed to operate at frequencies used for both over-the-air and cable television with National Television Standards Committee (NTSC) coded video. Receiver sensitivity is V
It is set to be limited by the antenna noise temperature for the HF system. The present invention also uses global automatic gain control (AGC).

The baseband video output of the present invention is stabilized, ie, minimizes video amplitude variations with respect to antenna RF signal level, and at lower operating currents, either a low impedance studio standard load or a high impedance load. Can be configured through software for The audio output is wideband hybrid for connection to an external MTS decoder. The audio output can be configured for low impedance studio standard loads or high impedance loads through software control.

Control is achieved through the I ² C bus interface. The bias and control circuit in the preferred embodiment of the present invention comprises NTSC, PAL, SECAM and MT
Includes internal registers that can be updated through the control bus for operating frequencies such as S, video and audio transmission standards, power, and test modes. The status of the bias and control circuit can be checked via a status register accessible through the I ² C bus interface. Status data includes AFC errors, channel locks, and received signal strength indicators.

The operating frequency of the present invention is based on an external crystal or reference frequency generator. According to the invention, minimal external components are used and no adjustment of any components is required.

The present invention can be implemented in a bipolar, BiCMOS, or CMOS process, however, the preferred embodiment of the present invention uses a BiCMOS process to maximize flexibility and design difficulties. Reduce sex.

A preferred embodiment of the invention is that after IF amplification,
Perform sound processing before video demodulation. This method is known as "quasi-split" sound and is not used in most television receivers for cost reasons. Quasi-split sound has no "buzz" in the audio signal while the video image is completely white. In the integrated circuit of the present invention, the quasi-split sound is negligibly low in cost and can therefore be integrated in all embodiments of the present invention. The integration of quasi-split sounds reduces the total number of elements and does not require external manual adjustments.

The foregoing has outlined rather broadly the features and technical advantages of the present invention in order that the detailed description of the integrated television tuner that follows may be better understood. Additional features and advantages of the monolithic television tuner that form the subject of the claims of the present invention are described below. One skilled in the art will recognize that the disclosed concepts and particular embodiments can be readily utilized as a basis for designing modifications or other configurations to achieve the same objects of the invention. . Also, those skilled in the art will recognize that such equivalent constructions do not depart from the spirit and scope of the invention as set forth in the appended claims.

[0041]

BRIEF DESCRIPTION OF THE DRAWINGS For a more complete understanding of the present invention and its advantages, reference is now made to the following description taken in conjunction with the accompanying drawings.

FIG. 1 shows a high-level block diagram of the functional elements included in the preferred embodiment of the TV tuner 100. RF signals received from antennas or other sources are converted to intermediate frequency (IF) signals by RF converter 110 and sent to IF processor 120. The output of the IF processor 120 is a vestigial sideband (VSB) detector 1
30 and to the video processor and driver 150 to generate the video output signal of the present invention. The video signal is detected through a synchronous AM demodulator.

The output of IF processor 120 is also FM
Detector 140 and audio processor and driver 1
60 to generate an audio output signal of the present invention. Audio detection is performed by a quadrature FM detector that utilizes a phase locked loop for a quadrature reference. I ²
The C interface 170 receives “I ² C data” and “I ² C clock” signals from the I ² C interface bus. Frequency reference 190 is an oscillator that is synchronized to an external reference crystal. Bias and control logic circuit 180 determines the internal bias voltage and current and maintains the status and control registers of the television tuner of the present invention.

FIG. 2 shows a pin layout diagram of a preferred embodiment of the integrated circuit of the present invention. The integrated circuit 200 is connected to the supply voltages (VCC1 and VCC2) on pins 11 and 20.
And pins 3, 4, 6-10, 14 and 17
(GND1 to GND9). The input reference signal from the external crystal oscillator is
(Crystal 1) and (Crystal 2 / Reference IN). The integrated circuit 200 has an IIC on pin 16 (serial clock) and pin 15 (serial data).
(I ² C) Connected to the interface bus. The RF input from the antenna or other source is connected to pin 5 (RF input). Integrated circuit 200 includes pins 18 and 19 (video and video reference) and pins 12 and 13.
Output video and audio signals on (audio and audio reference).

FIG. 4 is a detailed electric block diagram of a TV tuner according to a preferred embodiment of the present invention. FIG. 4 shows an up-conversion dual-conversion superheterodyne receiver. The vestigial sideband (VSB) coded video signal is processed through a Nyquist slope receiver attenuation characteristic filter prior to detection.

The RF signal enters TV tuner 100 from antenna 402 (or a cable not shown) and passes through an RF low pass filter (RFLPF) 404 to limit the incoming band to 900 MHz or less. The filtered RF signal is amplified to 20 dB by a gain controlled low noise transconductance amplifier (LNTA) 406.

The signal received by the antenna 402
The input signal filtered by F404 and amplified by LNTA 406 is the standard broadcast television spectrum. Channel 2-13 in the VHF band is 54 MH
Channels 14-83 in the UHF band extend from 470 MHz to 890 MHz.
Hz. Each channel in the VHF and UHF bands has a bandwidth of 6 MHz, the video carrier frequency is 1.25 MHz above the lower band edge, and the color carrier frequency is 3.58 MH above the video carrier.
z and the audio carrier frequency is located 4.5 MHz above the video carrier. For example, channel 2 has a 6 MHz bandwidth of 54-60 MHz, 5
It has a 5.25 MHz video carrier, a 58.83 MHz color subcarrier and a 59.75 MHz audio carrier.

The output of the first local oscillator (LO1) 450 operating between 1200 and 2100 MHz is mixed with the RF signal in a first mixer (MIX1) 408 to produce a first IF video carrier frequency of 1200 MHz. Occurs. This minimizes distortion due to the mixer image and harmonic mixing. The first IF is filtered by the bandwidth limitation of the first mixer 408 to minimize harmonic effects.

In the basic mixer scheme, the mixer receives two inputs, an RF input and a local oscillator (LO) input, and generates an IF output. The RF input has the following general form: [1] V _RF = A cos (ω _RF t) The local oscillator input has the following general form: [2] V _LO = B _COS (ω _LO t) The IF output is given by the following equation. [3] V _IF = V _RF · V _LO = AB / 2 [cos (ω _RF −ω _LO ) t + cos (ω _RF + ω _LO ) t]

In the frequency domain, the IF frequency is f _IF
= | F _RF ± f _LO |. In possible mixers, there are various non-linearities that generate harmonics at multiples of the RF and IF frequencies. Thus, the IF spectrum encompasses harmonics of f _IF = mf _RF ± nf _LO , where m and n are integers.

For example, if the output of local oscillator 450 is 400 MHz and the desired input signal to be demodulated from the antenna is 800 MHz, the basic mixer adds at the output of mixer 408 to obtain a 1200 MHz output. Mixing will be performed. However,
Since frequency synthesizer 450 also has harmonics at 800 MHz and 400 MHz signals are present at the antenna, these signals are also added to produce an 800 MHz output that interferes with the desired signal to be demodulated at 800 MHz. .

Mixer 408 is a subtraction mixer that subtracts the frequency of local oscillator 450 from the input RF spectrum between 0 Hz and 900 MHz filtered by RFLPF 404. For example, if the desired frequency to be demodulated in the RF input is 400 MHz, the local oscillator is set to 1600 MHz and the basic mixer is set to 2
Would produce two IF outputs. [4] f _IF = | 400-1600 | = 1200 MHz
And [5] f _IF = | 400 + 1600 | = 2000 MHz Since the mixer 408 is a subtraction mixer,
The signal is filtered and only the 1200 MHz signal is
Passed through 10. Also, since mixer 408 is a subtraction mixer, the incoming spectrum is inverted at the output, so the video carrier is 1.25 MHz below the upper edge of the 6 MHz channel bandwidth and the color carrier is 3.58 MHz below the video carrier. And the audio carrier is 4.5 MHz below the video carrier. For example, 6 MHz of channel 2 of 54-60 MHz
The bandwidth of z is 1201.25 MHz and 1199.55
It will appear inverted between MHz. The channel 2 video carrier will appear at 1200 MHz, the color carrier will appear at 1196.42 MHz, and the audio carrier will appear at 1195.5 MHz.

As described above, the fact that the harmonic signal exists at a multiple of the desired frequency is a feature of both the input RF signal received by the mixer 408 from the LNTA 406 and the frequency synthesizer 405. By using a sufficiently high frequency output from the subtraction mixer and oscillator 450, the harmonics on the output of the frequency synthesizer 450 are:
The frequency is 1200 MHz or more higher than any of the input frequencies in the range of 0 MHz to 900 MHz from the antenna 402.

For example, the minimum output of the local oscillator 450 is 1200 MHz. The first harmonic of this signal is 2
Present at 400 MHz. Since the highest output signal passing through LNTA 406 is 900 MHz, 2400 MHz
Subtractive mixing of the 900 MHz signal from antenna 402 with the harmonics will produce a signal at | 900-1200 | = 1300 MHz. This is higher than the desired 1200 MHz output from mixer 408 and can therefore be filtered out by a low pass filter. Using a local oscillator with an output higher than the RF input frequency is known as high side injection.

The first IF signal of 1200 MHz is supplied to a second mixer (MIX2) 410 which is an image rejection mixer.
At, the second IF is mixed with the fixed 1180 MHz reference output of the second local oscillator (LO2) 412 to generate a second IF on the 20 MHz image carrier. RF input signal is L
Since the frequency is lower than the O reference, mixing of the two signals will cause a down conversion of the RF input. f _IF = | f _RF
The IF frequency given by ± f _LO | is f _IF = 12
00-1180 = 20MHz and f _IF = 1200 + 11
It has a carrier at 80 = 2380 MHz. In particular, the image rejection mixer has f _IF = | 1160-1180 | =
Eliminate signal energy in the first IF signal in the 1160 MHz area, which produces unwanted signals at 20 MHz.

The image reject mixer receives an input signal from MIX 1 having the following general form: [6] V _RF = A cos (ω _RF t) The local oscillator signal from LO2 is divided into two phase quadrature signals having the following general form: _{[7] V LOI = Bcos (} ω LO t) [8] V LOQ = Bsin (ω LO t) desired output signal resulting from the mixing of the RF input and the phase quadrature signal, V _IF has the following general form ing. [9] The desired mixer signal of V _IF = AB cos (ω _RF −ω _LO ) tω _IF = ω _RF −ω _LO appears at the mixer output, while ω _IF = ω _RF + ω _LO (−1160 + 1180 =
The unwanted mixer image signal of 20) is eliminated.

FIG. 5 shows the RF front end of the present invention (ie, RF converter 110 and IF processor 120).
3 shows another specific example. Mixer 408 and mixer 51
The first IF filter (FIFF) 509 during 0 is 1
Filtering out all frequency components of the image going to 160 MHz, such that mixer 510 is a basic mixer similar to mixer 408 rather than an image rejection mixer. Although an image reject mixer is easy to integrate, it has limited accuracy and may be subject to noise and distortion.

Returning to FIG. 4, the second IF signal is then filtered by a second IF filter (SIFF) 420, a low distortion continuous time bandpass filter, for the final video bandwidth. SIFF 420 also includes an auto-tuned low-pass Nyquist slope filter (N
SF). -6dB on NSF slope
The points are maintained at the image carrier frequency (20 MHz). The output of SIFF 420 is an IF amplifier (IFAMP) which is a gain control amplifier applied to a gain of 80 dB.
422. Then, the output of IFAMP 422 is mixed with the video carrier at video detector (VDET) 426, which combines video subtraction filter (VCEF) 425 and video carrier limiter (VC
LIM) 428 to produce a baseband video output.

The video carrier is controlled by automatic frequency control (AF
C) AFCRE by frequency detector (DET) 440
Compared to the synthesized 20 MHz reference signal from F442,
Generates a frequency error signal that is returned to LO1 for automatic frequency control.

The baseband video from VDET 426 is passed through a low pass video bandwidth filter (VBBF) 430 to remove the detected harmonics. The synchronization pulse is clamped to a reference level by a synchronization clamp (SCMP) 432, and this is a phase lock which is locked to the horizontal synchronization speed, "RXLOCK", or optionally to the vertical speed, to control the clamp timing. Contains loops. Timing is also based on AGC 438 and N
Occurs at SCMP 432 for INV 436. SCMP 432 amplifies the video to its output voltage level with a fixed gain of 20 dB. The clamped video is processed by a noise inverter (NINV) 436 to remove excessively large positive or negative amplitude noise pulses.

The resulting video is tested by AGC 438 to blank out the sync level,
And this adjusts the gain of LNTA 406 and IFAMP 422 to maintain a standard 1 volt peak-to-peak video signal. The video at this stage is buffered by a video buffer (BVUF) 437 for off-chip driving using one of three signal impedance standards. VBUF 437 can also mute this signal.

The audio processing is performed using a narrow band audio IF.
IFAMP422 through filter (AIFF) 427
This is achieved by extracting the output audio signal. AIFF 427 is an audio IF limiter (AIFLIM) 44 to remove AM noise and distortion.
4 generates an FM signal limited by

An audio phase detector (APD) 455,
Audio delay lock loop low pass filter (ADLP
F) 458 and audio phase shifter (APS) 46
0 generates a signal that is delayed by 90 ° at the FM carrier frequency and converts the signal through a limiter in an audio quadrature detector (AQD) for quadrature detection of the FM. Mix with signal. The audio baseband is passed through a 120 kHz low band audio baseband filter (ABF) 464 to remove the detection synthesis component. Audio is 0.35
Audio buffer (A) for off-chip drive with drive impedance selectable at volts (rms) level
BUF) 466.

The control register 471 and the status register 472 correspond to the serial digital interface 17.
0 interface. Next, serial digital interface 170 receives external commands from an external microprocessor or microcontroller through the "serial data" and "serial clock" control lines. In a preferred embodiment of the present invention, serial digital interface 170 is a Philips proprietary inter-integrated circuit (I
It can be an IC or I ² C) interface.

System Block Specifications The overall design of the TV tuner 100 is governed by the limitations that can be reliably designed on an integrated circuit. T
The blocks of the V tuner 100 are defined by reference to actual circuit elements so that capabilities such as noise, maximum signal processing, intermodulation, and gain can be ensured. Perhaps the most important element of such a design is to limit the expected chip performance so that it can be manufactured at high yield and therefore at low cost. Note that all measurements in dBm are referenced to standard video RF and 75 ohm baseband impedance.

RF Low-Pass Filter (RFLPF) Table 1 shows the RF low-pass filter (RFLP) of the preferred embodiment of the present invention.
The operation parameters of the LPF 404 are listed.
The RFLPF 404 is a 900 MHz third-order low-pass filter including a package lead inductance, a printed circuit capacitance, and an on-chip capacitor. For most applications, the RFLPF 404 has a MI
Satisfactory for image rejection at X1 (2400-3300 MHz). An external filter matched to 75 ohm impedance could be added for extreme situations such as very large out-of-band signals. Table 1: RFLPF Specifications Parameter Description Min Typical Max Unit G _V voltage gain -1 0 dB f _C cutoff frequency (-3 d 900 MHz B)

A low noise transconductance amplifier (LN
TA) Table 2 lists the operating parameters of the low noise transconductance amplifier (LNTA) 406 of the preferred embodiment of the present invention. LNTA 406 is the front end of the receiver. The single-ended input signal is converted to a differential current passed through a mixer chain. A critical function of this circuit is to maintain a very low noise figure (NF), a considerable gain (G _V ), and a high third-order intermodulation suppression (IP ₃ ).
The gain is determined for the rest of the circuit, so that the block's noise figure determines the overall receiver noise figure. Since this is a transconductance amplifier,
The gain is limited by the load impedance of the final mixer stage and must be properly limited before the final determination of LNTA 406 performance. AGC capability is provided to reduce the effects of distortions such as intermodulation or cross modulation in the presence of high level signals. This input is
Impedance matched to 75 ohms. Table 2: LNTA Specifications Parameter Description Min Typical Max Unit G _V voltage gain -AGC no 20 30 dB G _VAGC voltage gain - fully AGC -10 0 dB NF noise numeric 4.5 5.4 _dB P _-1dB 1dB gain compression input level -24 dBm IP ₃ 3 order input suppression -10 dBm f _in the input frequency 40 900 MHz I _{DC DC} bias current 20 mA

First Mixer (MIX1) Table 3 shows the first mixer (MIX1) of the preferred embodiment of the present invention.
1) A list of 408 operation parameters. Proper rejection of unwanted images and spurious signals by local oscillator harmonics requires that MIX1 be configured for an up-conversion scheme with high-side injection. Thus, the output of MIX1 for the image carrier is at a microwave frequency of 1200 MHz,
The audio carrier is at 1195.5 MHz, and the input of the local oscillator 450 is 1.2 and 2.1 GH
between z. A unipolar low-pass element is incorporated in the output circuit of MIX1 to help reject downconverted signals due to local oscillator harmonics. MIX1 noise and distortion effects are included in the LNTA 406 specification. This design will optionally include a bond wire parallel LC network to further band limit any image energy at the output of the mixer if necessary. Table 3: MIX1 specifications Parameter Description Min Typical Max Unit f _RF RF input frequency 40 900 MHz f _LO LO1 input frequency 1200 2100 MHz f _IF IF output frequency 1195 1205 MHz V _LO LO1 input voltage 0.2 V (rms)

Second Mixer (MIX2) Table 4 shows the second mixer (MIX2) of the preferred embodiment of the present invention.
It is a list of the operation parameter of 2). The 1.2 GHz first IF (image carrier) is the 20 MHz second I
It is input to MIX2, which is an image rejection mixer that converts down to F. This image rejection is to counteract the need for filtering the first mixer output to reject signals entering the 1160 MHz image from being downconverted by MIX2 if MIX2 was a conventional subtraction mixer. Done. An important feature of MIX2 is that
High image rejection (at least 50 dB) must be achieved without the need for post-production trimming techniques to calibrate MIX2. As shown in FIG. 6, the 50 dB image rejection parallels two Gilbert cell multipliers at the MIX2 input, and a very low component that causes the adder 615 to have a phase error of 0.2 degrees or less over a 20 MHz bandwidth. This can be achieved by using balanced RC phase shifters 613 and 614 of higher order sensitivity. These are traditionally avoided by relatively high noise levels, but in the preferred embodiment of the TV tuner 100, this front provides sufficient gain to reduce noise effects. Table 4: MIX2 Specifications Parameter Description Min Typical Max Unit G _I current gain 0 dB NF noise numerical 10 dB f _RF RF input frequency 1195 1205 MHz f _LO LO2 input frequency 1180 MHz f _IF IF output frequency 15.25 21.25 MHz V _LO LO2 input voltage 0.2 V (rms) A _IM input image rejection 60 dB

Second IF Filter (SIFF) Table 5 shows the second IF filter (SIFF) of the preferred embodiment of the present invention.
IFF) 420 is a list of operation parameters. 20
MHzIF signal, the integrated transconductor - is filtered for the capacitor (G _m -C) channel bandwidth of the filter. Transition Gaussian bandpass filters are used to eliminate adjacent channels and noise. The SIFF 420 keeps distortion due to noise and out-of-band signals below -54 dBc. The in-channel group delay can be maximally flattened to the color subcarrier frequency. The audio signal is SIFF420
Pass through. Since the up-conversion scheme inverts the channel frequency at the output of MIX2, the image carrier is now 20
MHz and the voice carrier is at 15.5 MHz. The adjacent image carrier is now at 14 MHz and the adjacent audio carrier is at 21.5 MHz, as per the filter specifications below. Table 5: SIFF specifications Parameter Description Min Typical Max Unit Pass band limit below f _pl 16.75 MHz A (ω) <0.5 dB Pass band limit above f _ph 19.25 MHz A (ω) <0.5 dB Stop band limit under f _sl 15.25 MHz A (ω)> 54dB f sh up stop band limit 20.75 MHz A (ω)> 54dB G v voltage gain 0 dB IP ₃ 3 order input suppression 20 dBm NF noise numerical 15 dB

IF Amplifier (IFAMP) Table 6 shows the IF amplifier (IFAM) of the preferred embodiment of the present invention.
3 is a list of operation parameters of P). The 20 MHz signal is amplified 80 dB in this block. This gain is set to 0d by applying an automatic gain control (AGC) signal voltage.
B. This AGC is implemented in a fully differential fashion for best on-chip noise rejection. Table 6: IFAMP Specifications Parameter Description Min Typical Max Unit G _V voltage gain -AGC without 80 dB G _V voltage gain - fully AGC 20 dB NF noise numerical 10 dB BW Bandwidth 25 MHz

Image Carrier Extraction Filter (VCFF) Table 7 is a list of operating parameters of the image carrier extraction filter (VCFF) 425 of the preferred embodiment of the present invention. The 15 MHz video carrier is extracted from the IFAMP 422 output for synchronous detection processing. VCEF42
5 transmits any accompanying carrier phase modulation (ICPM) in the Nyquist filter without any group delay distortion for best detector performance. Table 7: VCEF Specifications Parameter Description Min Typical Max Unit G _V voltage gain 0 dB f _O center frequency 20 MHz BW bandwidth 0.5 1 3 MHz NF noise numeric 5 dB

Image Carrier Limiter (VCLIM) Table 8 lists the operating parameters of the image carrier limiter (VCLIM) 428 of the preferred embodiment of the present invention. Synchronous detectors require a fixed amplitude reference carrier.
This is obtained through VCEF 425, and then passed through VCLIM 428. With an input level variation of 20 dB and a modulation depth that can exceed 95 percent, the VCLIM 428 output is stable within -50 dB accuracy. VCEF425 and VCLIM428
The combined circuit delay must not exceed 45 ° at the image carrier frequency so that the detector output level and noise behavior do not degrade. Table 8: VCLIM Specifications Parameter Description Min Typical Max Unit G _V voltage gain 46 dB V _th limiter input threshold 10 mV (pp) V _limit limiter output level 2 V (pp) NF noise numeric 5 dB

Video Detector (VDET) Table 9 shows the video detector (VDET) of the preferred embodiment of the present invention.
T) A list of operation parameters of 426. The video is synchronously detected by applying the output of IFAMP 422 as a first input signal and the output of VCLIM 428 as a second input signal to a double balanced mixer (VDET 426). The output of VDET 426 is 0.5 dB
It is configured with a low band single pole set to pass 4.2 MHz with loss. Table 9: VDET Specifications Parameter Description Min Typical Max Unit G _V voltage gain 0 dB BW output -0.5 dB bandwidth 4.2 MHz V _video video output level 0.1 V (pp) NF noise numeric 5 dB

Video Baseband Filter (VBBF) Table 10 lists the operating parameters of the video baseband filter (VBBF) 430 of the preferred embodiment of the present invention. The detected video is a SIFF420 response and FC
It is band-limited to 4.2 MHz by a pre-distorted multi-pole transition Gaussian response low-pass filter for both compensation of group delay characteristics that meet the C rule. VBB
F430 is VDET426 with minimum passband transient distortion.
Output, 20MHz video carrier and high level 4
Eliminate the 0 MHz modulated signal. Table 10: VBBF Specifications Parameter Description Min Typical Max Unit G _V voltage gain 0 dB f _p passband corner frequency 4.2 MHz A (ω) <0.5dB f s stopband corner frequency 8.0 MHz A (ω)> 54dB NF noise numerical 5 dB

Synchronous Clamp (SCMP) Table 11 shows the synchronous clamp (SCM) of the preferred embodiment of the present invention.
P) 432 is a list of operation parameters. The sync pulse is clamped through two techniques. The diode clamp limits the minimum level at which the sync tip is set. Once this is achieved, after one sync pulse, the sync slicer regenerates only the sync pulses and passes them through a phase locked loop (PLL). PLL
Obtains a horizontal sync rate and generates a sync gate pulse approximately centered within the horizontal sync pulse.

[0077] Once the PLL is locked, the gated clamp circuit using the synchronization gate pulse is enabled to limit the synchronization level, and the diode clamp is disabled. For the processing of scrambled video, a line counter is inserted into the phase locked loop to count the fields, its sync signal is integrated for detecting vertical sync, and only the vertical sync pulse is used for the clamp gate. Used for The synchronization gate pulse is used through a 5 microsecond clocked delay to generate the blanking gate used for AGC processing. SCMP 432 also amplifies the video to its final output level. The output signal "RXLOCK" 434 becomes active when the phase lock loop has locked onto the synchronization pulse. Table 11: SCMP Specifications Parameter Description Min Typical Max Unit G _V voltage gain 20 dB V _Video Video Output Level 1 V (pp) f _PLLH PLL operating frequency 15750 kHz horizontal synchronization mode f _PLLV PLL operating frequency 60 Hz vertical sync mode

Noise Inverter (NINV) Table 12 shows the noise inverter (NINV) of the preferred embodiment of the present invention.
9 is a list of operation parameters of the INV) 436. SC
The MP432 video output will contain very transient noise pulses in both black and white directions. Black noise is typically a large RF transient pulse due to some form of interference. This signal drops below the blanking level during the horizontal line video period and may disrupt downstream sync detection circuits. A comparator detects these levels below the blanking level during a video active or non-blank period, and allows an inverted video path while being added to the original video. This addition negates the effects of noise pulses so that the sync level region is not disturbed. Extreme positive excursions, appearing as white, are also canceled when they exceed a level equivalent to 108 IRE white or below 5 percent modulation.
These are typically due to desensitization of the receiver when pulsed interference is present at out-of-band frequencies, and typically results in shut-off of the synchronous detector. 108 IRE
Positive noise inversion is achieved using a level-based comparator and allowing a video inversion path to the adder as in the negative case described above. Table 12: NINV Specifications Parameter Description Min Typical Max Unit V _Video Video Input Level 1 V (pp) V _WINS White Noise Inverter 95 IRE Insertion Level V _BINS Black Noise Inverter 10 IRE Insertion Level V _WTH White Inverter Threshold 108 IRE V _BTH Black Inverter Threshold 0 IRE Non-blank

Automatic Gain Control (AGC) Table 13 shows the automatic gain control (AGC) of the preferred embodiment of the present invention.
C) is a list of operation parameters of 438; The clamped video is directed to a sample and hold circuit controlled by a blanking gate. The sample and hold circuit output drives a differential amplifier having a reference level at another input equal to the desired output blanking level for a standard NTSC video waveform.

At each horizontal sync pulse, the output of the differential amplifier represents the gain error of the receiver to achieve the video waveform. This is fed back to LNTA 406 and IFAMP 422 to adjust the gain when needed. The timing and gain of the differential amplifier must be set for fast transient response times and absolute stability to compensate for aircraft-induced phasing. This response can be changed to update the gain only during the vertical sync period, as by SCMP432,
Thus, the scrambled video can pass through the tuner without any adverse effects.

The AGC voltage defined by the RF and IF amplification performance and the input signal level is measured through a 4-bit analog-to-digital converter and stored as a data word received signal strength indicator (RSSI) 439. Automatic gain control is first applied to IFAMP 422 to reduce the video level, and is also applied to LNTA 406, which reduces the video level by more than 50 dB to perform the delayed AGC function. Table 13: AGC specifications Parameter Description Minimum Typical Maximum Unit V _Video Video input level 1 V (pp) V _sb Blanking level reference Input synchronization to 0.286 V G _AGC AGC rope gain 70 dB GBW _H AGC gain-bandwidth 150 Hz H mode GBW _V AGC gain-bandwidth 6 Hz V mode Ф _MAGC AGC rope phase margin 60 °

Video Output Buffer (VBUF) Table 14 shows the video buffer (VB) of the preferred embodiment of the present invention.
UF) 437 is a list of operation parameters. Video is an internal level of 1 volt peak-to-peak difference.
VBUF 437 converts this signal to current through a linear transconductance amplifier that drives a load resistor.
Feedback is placed through an external signal to allow continuous adjustment of the amplifier gain and offset so that AC coupling is not required for the next stage. An on-chip integrator is used to limit the feedback level,
And eliminate video signals. Control signal VOUT-MOD
E445 is 75 ohms equal to the current source output, 1000
Enable on-chip load with ohms or no load. The control signal VIDEO-MUTE 446 enables a video mute switch that sets the output to a blanking level. Table 14: VBUF Specifications Parameter Description Min Typical Max Unit V _Video Video Input Level 1 V (pp) V _OV Video Output Level 1 V (pp) ( _RL > 70Ω) V _OVR Video Output DC 0.5 3.5 V Reference ( _RL > 70Ω) Z _OUT output impedance 73 75 77Ω 75Ω mode Z _OUT output impedance 980 1000 1020Ω 1000Ω mode Z _OUT output impedance 10kΩ Current mode

Automatic Frequency Control (AFC) Frequency Detector Table 15 shows the automatic frequency control (AF) of the preferred embodiment of the present invention.
C) A list of operating parameters of the frequency detector (DET) 440. The extracted IF video carrier is 20
MHz and is used to drive LO1 in a direction that equalizes the two. An error in LO1 from the frequency control setting is signal AFC-ERR
OR452. A 20 MHz reference is generated from the master reference through the frequency synthesizer. The 20 MHz reference is compared at the output of the VCEF 425 with the extracted video carrier signal using a digital frequency comparator, and this is a 7 bit up that is added to the lower bits of the frequency code sent from the I ² C interface 170 to LO1 Drive down counter. Thus, the frequency error appears in a counter that can be read through the interface bus. This counter is set to 10000000b when there is no frequency error. This condition sets the status signal FREQ-LOCK 453. Table 15: AFC Specifications Parameter Description Min Typical Max Unit V _CIN Video Carrier Input Level 0.1 V (pp) f _ERROR Frequency Error from Nominal -500 500 kHz r _UPDATE Frequency Update Rate 30 Hz

AFC Reference Oscillator (AFCREF) Table 16 shows the AFC reference oscillator (AFC) of the preferred embodiment of the present invention.
FCREF) 442 is a list of operation parameters.
AFCREF 442 is a fixed frequency synthesizer operating at 20 MHz and is phase locked to a crystal frequency reference. Phase noise is not particularly significant because it is applied to this circuit only at low modulation frequencies. Table 16: AFCREF Specifications Parameter Description Min Typical Max Unit V _C carrier level _{0.2 V (pp) N op 20Hz} < phase noise f _O operating frequency 20 MHz for f _m <1MHz -60 dBc

First Local Oscillator (LO1) Table 17 is a list of operating parameters of the first local oscillator (LO1) 450 of the preferred embodiment of the present invention. L
O1 is a phase locked frequency synthesizer configured for a special level of phase noise to allow the best receiver sensitivity without excessive spurious noise response. It is digitally controlled and utilizes a crystal frequency reference and in-band phase noise control. Frequency control input F from serial digital interface 170 bus
REQUENCY 451 has 16 bits with 62.5 kHz LSB resolution. A further 4 LSBs along with the first 3 LSBs are used by the AFC for fine tuning. Table 17: LO1 Specifications Parameter Description Min Typical Max Unit V _c carrier level _{0.2 V (pp) N op f} m> 1 for -110 dBc phase noise N _op 60 MHz <phase noise f _O operations to f _m <1 -80 dBc MHz Frequency 1200 2100 MHz f _STEP frequency step 3.906 kHz f _REF reference frequency 62.5 kHz

Second Local Oscillator (LO2) Table 18 is a list of operating parameters for the second local oscillator (LO2) 412 of the preferred embodiment of the present invention. L
O2 is a fixed tuning frequency synthesizer set at 1180 MHz. It uses the same frequency reference as LO1 and has similar phase noise characteristics. Table 18: LO2 Specifications Parameter Description Min Typical Max Unit V _C carrier level _{0.2 V (pp) N op f} m> 1MHz phase noise f _O operations to -110 dBc phase noise _{N op 60Hz <f m <1} -80 dBc MHz for Frequency 1180 MHz f _REF reference frequency 62.5 kHz

Audio IF Filter (AIFF) Table 19 is a list of operating parameters of the audio IF filter (AIFF) 427 of the preferred embodiment of the present invention. The audio IF signal is tapped from the output of IFAMP 422 and has a bandwidth of 300 kHz.
It operates through a narrow band filter operating at 5 MHz.
AIFF 427 removes video carrier and burst frequency components. Table 19: AIFF Specifications Parameter Description Min Typical Max Unit G _V voltage gain 0 dB f _O center frequency 15.5 MHz BW bandwidth 300 kHz NF noise numerical 10 dB

Audio IF Limiter (AIFLIM) Table 20 lists the operating parameters of the audio IF limiter (AIFLIM) 444 of the preferred embodiment of the present invention. The audio IF is limited by a possible 40 dB variation in input level based on the difference between the AGC 438 and the broadcast signal source. Output level accuracy in this range is better than -40 dB. Table 20: AIFLIM Specifications Parameter Description Min Typical Max Unit G _V voltage gain 40 dB V _th limiter input threshold 2 mV (pp) V _limit limiter output level 0.2 V (pp) NF noise numerical 10 dB

Audio Phase Detector (APD) Table 21 is a list of operating parameters of the audio phase detector (APD) 455 of the preferred embodiment of the present invention. The audio IF is connected to the APS 46 in the delay locked loop.
APS4 to generate a phase error signal driving 0
60 outputs. The APD 455 is a phase detector configured for fast loop acquisition and minimum phase once acquired. Table 21: APDM specifications Parameter Description Min Typical Max Unit V _IN voltage input level 0.35 V (rms)

The audio DLL low-pass filter (AD
LPF) Table 22 is a list of operating parameters of the audio delay locked loop low pass filter (ADLPF) 458 of the preferred embodiment of the present invention. The ADLPF 458 uses two to achieve full audio bandwidth in the detected signal.
It must not respond to any frequency above 0 Hz.
This low pass filter meets this requirement. Table 22: APLPFM Specifications Parameter Description Min Typical Max Unit G _V voltage gain 0 dB BW loop bandwidth 20 Hz Z _L closed loop zero 10 Hz

Audio Phase Shifter (APS) Table 23 lists the operating parameters of the audio phase shifter (APS) of the preferred embodiment of the present invention. APS
460 is a voltage controlled phase shifter that is locked to the audio carrier through a delay locked loop. APS
460 produces a 90 degree phase shift in the output signal at an audio carrier frequency of 15.5 MHz. Table 23: AQOM Specifications Parameter Description Min Typical Max Unit V _OUT output voltage 0.1 V f _O center frequency 15.5 MHz O _S phase shift 90 Deg.

Audio Quadrature Detector (AQD) Table 24 lists the operating parameters of the audio quadrature detector (AQD) 462 of the preferred embodiment of the present invention. A
QD 462 is a mixer used to compare the incoming audio IF signal with a delayed version of the incoming carrier orthogonal IF signal. This output is the desired audio baseband signal. Table 24: AQD Specifications Parameter Description Min Typical Max Unit G _V voltage gain 0 dB

An audio baseband filter (AB
F) Table 25 is a list of operating parameters for the audio baseband filter (ABF) 464 of the preferred embodiment of the present invention. ABF 464 is a high order Chebyche low pass filter design with a cutoff at 120 kHz. This is wide enough to pass all the multiple audio subcarriers for external decoding. Table 25: ABF Specifications Parameter Description Min Typical Max Unit G _V voltage gain 10 dB f _p passband corner frequency 120 kHz A (ω) <0.5dB f s stopband corner frequency 300 kHz A (ω)> 54dB

Audio Output Buffer (ABUF) Table 26 is a list of operating parameters for the audio output buffer (ABUF) 466 of the preferred embodiment of the present invention. This audio is at an internal level of 0.1 volt peak-to-peak. ABUF466 converts this signal to:
It is converted to current through a linear transconductance amplifier that drives a load resistor. Feedback is arranged through an external signal so that the gain and offset of the amplifier can be continuously adjusted so that no AC coupling is required in the next stage. On-chip integrators are used to limit feedback levels and reject audio signals.

The control signal AOUT-MODE 467 enables an on-chip load of 600 Ω, 10 kΩ or no load equal to the current source output. Control signal AUDIO-MUT
E468 controls an audio mute switch that disables output. Table 26: ABUF specifications Parameter Description Minimum Typical Maximum Unit V _video audio input level 0.1 V (pp) _Vov audio output level 0.35 V (pp) ( _RL > 500Ω) V _OVR audio output DC reference 0.5 _3.5V ( _RL > 500 Ω) Z _OUT output impedance 588 600 612 Ω 600 Ω mode Z _OUT output impedance 9.8 10.0 10.2 kΩ 10 kΩ mode Z _OUT output impedance 100 kΩ Current mode

Serial Digital Interface Tables 27-29 define the registers, register bits, and bit functions addressable by serial digital interface 170 in the preferred embodiment of the present invention. For example, control and status are controlled by an inter-integrated circuit (I ²
C) Can be performed through a bus interface. Serial digital interface 170 includes all registers and provides access to all parallel digital on-chip functions through a serial bus. Table 27: Register Definition Register Name Description Register Read / Write Address FREQO Frequency-Low Byte 00 Read / Write FREQ1 Frequency-High Byte 01 Read / Write CTRL0 Function Control 02 Read / Write CTRL1 Function Control 03 Read / Write STAT0 Function Status 04 Read STAT1 Function Status 05 Read STAT2 Function Status 06 Read CMPY0 Company Code-Low Byte 07 Read CMPY1 Company Code-High Byte 08 Read REV0 Revision Code-Low Byte 09 Read REV1 Revision Code-High Byte 0A Read

The definition of the bits used in each register is as follows. Table 28: Register bit definition Register Bit7 Bi6 Bit5 Bit4 Bit3 Bit2 Bit1 Bit0 FREQ0 F7 F6 F5 F4 F3 F2 F1 F0 FREQ1 F15 F14 F13 F12 F11 F10 F9 F8 CTRL0 PWRDN AMUTE VMUTE MODE3 MODE2 MODE1 MODE0 CTRL1 VRS1 VRS0 ARS1 FLOCK ARS1 FLOCK AFC1 AFC0 STAT1 RSSI3 RSSI2 RSSI1 RSSI0 STAT2 T7 T6 T5 T4 T3 T2 T1 T0 CMPY0 C7 C6 C5 C4 C3 C2 C1 C0 CMPY1 C15 C14 C13 C12 C11 C10 C9 C8 REV0 R7 R6 R5 R4 R3 R1 R11 R10 R10 R9 R8

The status register is read-only. Attempting to write to the status register is useless. The function of each bit is as defined below. Table 29: Bit functions Bit name Function Description F0-15 Receiver frequency f = 0.0625F + K (MHz) (F15 = MSB, F0 = LSB) PWRDN circuit power down When high, the receiver is disabled and video and The audio output is muted and I ² C remains functional. Circuit current is minimal. AMUTE Audio Mute When high, audio output is disabled. VMUTE Video Mute When high, video output is disabled. MODE0-3 Receiver mode selection Receiver operation mode selection (not used for NTSC / PAL / SECAM switching, reserved) VRS0-1 Video output impedance 00 = 75Ω, 01 = 1kΩ, 02 = power selection ARS0 -1 Audio output impedance 00 = 600Ω, 01 = 10kΩ, 02-Dance selection = Current source RLOCK Receiver lock High if AFC is locked to incoming signal. FLOCK Synthesizer Lock High if the frequency synthesizer is locked. AFC0-2 AFC offset Indicates the value and direction of receiver mistuning. A The three MSBs of the FC counter. RSSI0-3 Received signal strength Indicates the relative signal strength of the Indy signal. 0000 is not a valid caterer signal. 1111 is a large signal. (RSSI3 = MSB, RSSI0 = LSB ) T0-7 data are used only for test data for testing purposes C0-15 company code Yu nonunique identifier for Cirrus Logic I ² C component R0-15 revision code part number and Unique identifier for revision

Bias and Control (BC) Table 30 is a list of operating parameters of the bias and control logic (BC) 180 of the preferred embodiment of the present invention.
Bias voltages and currents are generated to limit the particular level required for each circuit element of the present invention. The control signal POWERDOWN 476 controls the power down function through the BC 180. Table 30: BC specification Parameter Description Min Typical Max Unit F _OSC crystal oscillator frequency reference 8 MHz

While the invention and its advantages have been described in detail, various changes, substitutions, and alterations may be made herein without departing from the spirit and scope of the invention as defined by the appended claims. It will be understood that can be.

[0101]

【The invention's effect】

(1) The present invention can provide a TV tuner having a relatively low cost and a small physical space for use on a printed circuit board. (2) The present invention provides a state-of-the-art TV tuner while reducing the number of external components required at the same time, thereby reducing the complexity of the printed circuit board and the circuit board area required by the TV tuner. TV tuners that meet or exceed the performance of the TV tuner can be provided. (3) The present invention enables computer control of a TV tuner via a serial bus so that the TV tuner can be controlled by a microcontroller embedded in a television set, personal computer, or other video device be able to. (4) The present invention can provide a TV tuner that adapts computer-controlled output impedance characteristics to different load specifications.

[Brief description of the drawings]

FIG. 1 is a high-level block diagram of the functional elements of the present invention.

FIG. 2 is a pin layout diagram of the integrated circuit of the present invention.

FIG. 3 is a detailed block diagram of a current television tuner found in the prior art.

FIG. 4 is a detailed block diagram of the circuit element of the present invention.

FIG. 5 shows another embodiment of the RF front end of the present invention.

FIG. 6 is a detailed block diagram of MIX2 of FIG. 4;

[Explanation of symbols]

 Reference Signs List 100 TV tuner 110 RF converter 120 IF processor 130 VSB video detector 140 FM audio detector 150 Video processor and driver 160 Audio processor and driver 170 Serial digital interface 180 Bias and control 190 Frequency reference

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[Procedure amendment]

[Submission date] June 5, 1997

[Procedure amendment 1]

[Document name to be amended] Drawing

[Correction target item name] All figures

[Correction method] Change

[Correction contents]

FIG. 2

FIG.

FIG. 5

FIG. 3

FIG. 4

FIG. 6

Claims (118)

[Claims] A receiver input coupled to an RF signal source; a first reference signal having a first operating frequency; a first input coupled to the receiver input and the reference signal. A first mixer having a second input; a second reference signal having a second operating frequency; a first input and the second reference signal directly coupled to an output of the first mixer. A second mixer having a second input coupled to and comprising an image rejection mixer. 2. The television receiver according to claim 1, wherein said first mixer is a subtraction mixer. 3. The minimum frequency value of the first reference signal is:
3. The television receiver according to claim 2, wherein the input frequency is higher than an input cutoff frequency of the RF signal received from the RF signal source. 4. The television receiver according to claim 1, wherein the first operating frequency is variable between a minimum frequency value and a maximum frequency value. 5. The first mixer subtractively mixes the reference signal and an RF signal received from the RF signal source, thereby providing a first I.sub.M output to the first mixer output.
The television receiver according to claim 4, which generates an F signal. 6. The television receiver according to claim 5, wherein a selected carrier frequency in the RF signal appears at a first predetermined frequency of the first IF signal. 7. The television receiver according to claim 6, wherein said first predetermined frequency is 1200 MHz. 8. The television receiver according to claim 6, wherein the selected carrier frequency is selected by changing the first operating frequency. 9. The television receiver of claim 5, wherein said first mixer is coupled to said receiver input by an input filter that filters all frequency components of said RF signal above an input cutoff frequency. . 10. The television receiver according to claim 9, wherein a minimum frequency value of the first reference signal is higher than the input cutoff frequency. 11. The second mixer mixes the second reference signal and the first IF signal, thereby including all channels of information as being included in the RF signal. 7. The television receiver according to claim 6, which generates a second IF signal. 12. The television receiver according to claim 11, wherein the second operating frequency is fixed. 13. The method according to claim 13, wherein the second mixer is configured to
The television receiver according to claim 12, wherein the image signal of the second reference signal is excluded from the F signal. 14. The television receiver according to claim 13, wherein the selected carrier frequency in the RF signal appears at a second predetermined frequency in the second IF signal. 15. The television receiver according to claim 13, wherein the minimum frequency value of the first reference signal is higher than an input cutoff frequency of the RF signal. 16. The RF signal source further comprising a first IF signal generated by the first mixer, wherein the second mixer mixes the second reference signal and the first IF signal. The television receiver of claim 1, wherein the television receiver generates a second IF signal that encompasses all channels of information as being included in an RF signal received from the television receiver. 17. The television receiver according to claim 16, wherein the second operating frequency is fixed. 18. The method according to claim 18, wherein the second mixer is configured to
The television receiver according to claim 17, wherein the image signal of the first IF signal is excluded from the F signal. 19. The method according to claim 19, wherein a selected carrier frequency in an RF signal received from the RF signal source is equal to the first IF.
Signal at a first predetermined frequency and the second I
19. Appearing at a second predetermined frequency in the F signal.
The television receiver according to the above. 20. The television receiver according to claim 19, wherein the second predetermined frequency is a frequency difference between the first predetermined frequency and the second operating frequency. 21. The apparatus according to claim 21, wherein the second operating frequency is lower than the first predetermined frequency, and the frequency of the image signal rejected by the second mixer is equal to the second predetermined frequency. The television receiver according to claim 20, wherein the television receiver is lower than the second operating frequency. 22. The television of claim 20, wherein the first predetermined frequency is 1200 megahertz, the second operating frequency is 1180 megahertz, and the second predetermined frequency is 20 megahertz. Receiving machine. 23. A receiver input coupled to an RF signal source, a first input coupled to the receiver input and a second input coupled to a first reference signal having a first operating frequency. Physically located on an integrated circuit substrate having
A first mixer wherein the first operating frequency is variable between a minimum frequency value and a maximum frequency value; and a first mixer physically located on the same integrated circuit substrate as the first mixer; and A first input coupled directly to the output of the mixer without leaving the substrate, and a second input coupled to a second reference signal having a second operating frequency; and A second mixer having a fixed frequency; and a television receiver. 24. The television receiver according to claim 23, wherein the minimum frequency value is higher than an input cutoff frequency of an RF signal received from the RF signal source. 25. The first mixer subtractively mixes the RF signal and the first reference signal, thereby generating a first IF signal, wherein a selected carrier frequency in the RF signal is 25. Appear at a first predetermined frequency in a first IF signal, and wherein said first predetermined frequency is greater than said selected carrier frequency.
The television receiver according to the above. 26. The method according to claim 26, wherein the second mixer is provided with the first IF.
Mixing the signal with the second reference signal, thereby generating a second IF signal, and wherein the selected carrier frequency appears in the second IF signal at a second predetermined frequency; and The second predetermined frequency is lower than the first predetermined frequency and the second operating frequency.
6. The television receiver according to 5. 27. The method according to claim 27, wherein the second mixer includes the second IF.
At least one signal associated with the second operating frequency from the signal;
27. The television receiver according to claim 26, wherein two television signals are eliminated. 28. mixing the RF signal with a first reference signal having a first operating frequency, thereby generating a first IF signal; and converting the first IF signal to a second operating frequency. Mixing with a second reference signal having a second IF signal, thereby generating a second IF signal;
The mixing of the reference signal and the first IF signal occurs before any channels are removed from the first IF signal, and the RF signal mixing and the first IF signal mixing operation include:
Steps achieved on the same integrated circuit substrate,
A method for processing a received RF signal comprising: 29. The method of claim 28, further comprising filtering all frequency components above the input cutoff frequency from the RF signal. 30. The method of claim 29, wherein the first operating frequency is higher than the input cutoff frequency. 31. The method of claim 30, wherein mixing the RF signal with the first reference signal is a subtractive mixing of the RF signal and the first reference signal. 32. The step of mixing the RF signal with the first reference signal is up-conversion mixing, and the selected carrier frequency in the RF signal is higher in the first IF signal than the RF signal. 32. The method of claim 31, wherein the method appears. 33. The step of mixing the first IF signal with the second reference signal further comprises the step of removing at least one image signal associated with the first IF signal from the second IF signal. 28. The method according to 28. 34. Filtering all frequency components above the input cutoff frequency from the received RF signal, thereby generating a first filtered signal; amplifying the first filtered signal; Subtractively mixing the first filtered signal with a first reference signal having a first operating frequency, thereby generating a first IF signal; Mixing with a second reference signal having an operating frequency, thereby generating a second IF signal, wherein said mixing of the second reference signal and the first IF signal comprises any channel from the first IF signal. C. Before removing the received RF signal. 35. The method of claim 34, wherein the first operating frequency is higher than the input cutoff frequency. 36. The step of mixing the RF signal with the first reference signal is up-conversion mixing, and a selected carrier signal in the RF signal appears in the first IF signal at a first predetermined frequency. And the first predetermined frequency is RF
36. The method of claim 35, which is higher than the signal. 37. The step of mixing the first IF signal with the second reference signal further comprises the step of removing at least one image signal associated with the first IF signal from the second IF signal. 35. The method of claim 34. 38. The step of mixing the first IF signal with the second reference signal is down-conversion mixing, and the selected carrier signal appearing in the first IF signal at the first predetermined frequency is: The method of claim 34, wherein the second predetermined frequency appears in the second IF signal at a second predetermined frequency and the second predetermined frequency is lower than the first IF signal. 39. The second I
35. The method of claim 34, further comprising filtering the F signal. 40. The band pass filter having a pass band between 15 MHz and 21 MHz.
The method described in. 41. The method of claim 39, wherein the bandpass filter is a Nyquist slope filter having a cut-off frequency of 20 megahertz. 42. An input filter coupled to an RF signal source, a first amplifier coupled to an output of the input filter, a first input and a second input, and the first input. A first mixer coupled to an output of the first amplifier; and generating a first reference signal having a first operating frequency coupled to the second input of the first mixer. A second oscillator having a first input and a second input, the first input being coupled directly to the output of the first mixer without connecting a filter therebetween. A second oscillator coupled to the second input of the second mixer and generating a second reference signal having a second operating frequency; and a second oscillator coupled to an output of the second mixer. Television consisting of a second filter Receiving machine. 43. The television receiver according to claim 42, wherein the input filter is a low-pass filter having a -3 dB cut-off frequency of 900 MHz. 44. The television receiver of claim 42, wherein said first amplifier is a transconductance amplifier having a noise figure between 4.5 dB and 5.4 dB. 45. The television receiver according to claim 42, wherein a gain of the first amplifier is automatically controlled by an automatic gain control circuit of the television receiver. 46. The first mixer outputs a filtered and amplified RF signal having a maximum cutoff frequency.
Mixing with the first reference signal, thereby producing a first IF
43. The television receiver of claim 42, wherein the television receiver is a rising converter that generates a signal, and a lower limit of the first operating frequency is greater than the maximum cutoff frequency. 47. The method according to claim 47, wherein the first oscillator is a phase locked synthesizer and the first operating frequency is 210
43. The television receiver of claim 42, having an upper limit of 0 megahertz and a lower limit of 1200 megahertz. 48. The apparatus of claim 42, wherein the second mixer is an image reject mixer that downconverts a first IF signal output from the first mixer, thereby generating a second IF signal. TV receiver as described. 49. The second oscillator is a fixed tuned frequency synthesizer and the second operating frequency is approximately 1
43. The television receiver according to claim 42, which is at 180 MHz. 50. The filter of claim 42, wherein the second filter is a bandpass filter having a passband upper limit of 20.75 MHz and a passband lower limit of 15.8 MHz.
The television receiver according to the above. 51. The television receiver according to claim 42, wherein said second filter is a Nyquist slope filter. 52. The television receiver of claim 51, wherein the second filter has a -6dB gain point at a visual carrier frequency of an IF signal generated by the second mixer. 53. The visual carrier frequency is approximately 20 megahertz and the second filter is approximately 19.25.
53. The television receiver of claim 52 having a passband corner frequency of megahertz. 54. The second filter measures the level of a synchronization pulse at the input of the second filter and at the output of the second filter, and provides 6 dB at the output of the second filter. 53. The television receiver of claim 52, wherein the television receiver is automatically tuned by adjusting a cutoff frequency of the second filter such that there is a loss of the second filter. 55. The television receiver of claim 42, wherein said television receiver further comprises a second amplifier coupled to an output of said second filter. 56. The television receiver according to claim 55, wherein the gain of the second amplifier is automatically controlled by an automatic gain control circuit of the television receiver. 57. A third filter, wherein the television receiver is further coupled to an output of the second amplifier and is a bandpass filter for extracting a video carrier signal from an output signal of the second amplifier. Having a first input and a second input, the first input being coupled to the output of the second amplifier, and the second input being coupled to the output of the third filter. 56. The television receiver of claim 55, further comprising: a third mixer whose output signal at the output is a video passband signal. 58. The second input of the third mixer is coupled to the third filter by an amplitude limiting circuit, and the amplitude limiting circuit couples a fixed amplitude reference carrier signal to the third filter. 58. The television receiver of claim 57 for providing to a mixer. 59. The television receiver further comprises a frequency detection circuit having a first input and a second input, and wherein the first input of the frequency detection circuit is the output of the third filter. 58. The television receiver of claim 57, wherein the second input of the frequency detection circuit is coupled to a third reference signal having a third operating frequency. 60. The frequency detection circuit compares the frequency of the video carrier signal with the third operating frequency, thereby generating an output control signal, and wherein the output control signal is generated by the first oscillator The television receiver according to claim 59, which is used to control the first operating frequency of the first reference signal. 61. The television receiver further comprises a fourth filter coupled to the output of the third mixer, wherein the fourth filter has all of the higher frequencies than the video passband signal. 58. The television receiver according to claim 57, wherein the television receiver is a low-pass filter that cuts off a signal component of? 62. The television receiver further includes a clamp circuit for clamping a synchronization pulse in the video passband signal from the fourth filter.
The television receiver according to claim 1. 63. The television of claim 62, wherein the television receiver further comprises a noise inverter circuit coupled to the clamp circuit, and the noise inverter circuit removes transient noise pulses from the video baseband signal. Receiving machine. 64. The television receiver of claim 63, wherein the transient noise pulse comprises a transient noise pulse that drops the video baseband signal below a blanking level of the video baseband signal. 65. The television receiver of claim 63, wherein the transient noise pulse comprises a transient noise pulse that raises the video baseband signal above the 108 IRE white level of the video baseband signal. 66. The television receiver further comprises an automatic gain control circuit coupled to the output of the third mixer, wherein the automatic gain control circuit controls a gain of the second amplifier. 58. A television receiver according to item 57. 67. The television receiver according to claim 66, wherein the automatic gain control circuit controls a gain of the first amplifier. 68. The television receiver of claim 66, wherein said automatic gain control circuit comprises a sample and hold circuit coupled to a first input of a differential amplifier. 69. The television receiver of claim 68, wherein a second input of the differential amplifier is coupled to a reference voltage equal to an output blanking level of a standard NTSC video baseband signal. 70. The television receiver of claim 57, wherein the television receiver further comprises a video output buffer amplifier coupled to the output of the third mixer. 71. The television receiver of claim 70, wherein said video output buffer amplifier comprises a linear transconductance amplifier. 72. The television receiver of claim 71, wherein said video output buffer further comprises blanking means for setting an output signal of said video output buffer amplifier to a video blanking level. 73. The television receiver of claim 72, wherein said video output buffer amplifier further comprises switching means for changing an output load of said video output buffer amplifier. 74. The television receiver further comprises a third filter coupled to an output of the second amplifier, wherein the third filter has a video output signal at the output of the second amplifier. 56. The television receiver of claim 55, wherein the television receiver is a narrow band filter for filtering audio component signals from the television. 75. The television receiver of claim 74, wherein the third filter has a center frequency of 15.5 megahertz and a bandwidth of 300 kilohertz. 76. The television receiver further comprises: an audio phase detector circuit coupled to an output of the third filter; a first input and a second input, wherein the first input is 75. The television receiver of claim 74, comprising: a third mixer coupled to the output of a third filter, the second input of which is coupled to an output of the audio phase detector circuit. 77. The audio phase detector circuit comprising a delay locked rope, wherein the output signal of the delay locked loop operates at the output of the third filter in quadrature with the center frequency of the audio component signal. Term 76
The television receiver according to the above. 78. The television receiver further comprises a fourth filter coupled to an output of the third mixer,
77. The fourth filter is a low-pass filter having a cut-off frequency of 120 kilohertz.
The television receiver according to the above. 79. The television receiver of claim 76, wherein said television receiver further comprises an audio output buffer amplifier coupled to an output of said third mixer. 80. The audio output buffer amplifier comprises a linear transconductance amplifier.
The television receiver according to the above. 81. The television receiver of claim 79, wherein said audio output buffer amplifier further comprises mute means for setting an output signal of said audio output buffer amplifier to zero volts. 82. The television receiver of claim 79, wherein said audio output buffer amplifier further comprises switching means for changing an output load of said audio output buffer amplifier. 83. A receiver input coupled to an RF signal source; a first reference signal having a first operating frequency; a first input coupled to the receiver input; and the first reference signal. A first mixer having a second input coupled to the output of the first mixer and passing through one or more channels and performing partial image rejection and processed by the following circuit A first bandpass filter for limiting the overall signal power level to be performed; a second reference signal having a second operating frequency; a first input coupled to an output of the first bandpass filter; An image reject mixer having a second input coupled to the second reference signal. 84. The television receiver according to claim 83, wherein said first mixer is a subtraction mixer. 85. The television receiver according to claim 84, wherein a minimum frequency value of the first reference signal is higher than an input cutoff frequency of an RF signal output from the RF signal source. 86. The television receiver according to claim 83, wherein said first operating frequency is variable between a minimum frequency value and a maximum frequency value. 87. The first mixer subtractively mixes the first reference signal and an RF signal received from the RF signal source, thereby converting a first IF signal to the first mixer output. 87. The television receiver of claim 86, wherein the television receiver generates. 88. The television receiver according to claim 87, wherein the selected carrier frequency in the RF signal appears at a first predetermined frequency in the first IF signal. 89. The method according to claim 89, wherein the first predetermined frequency is 1200
89. The television receiver of claim 88, which is in the megahertz. 90. The television receiver of claim 88, wherein the selected carrier frequency is selected by changing the first operating frequency. 91. The television receiver of claim 87, wherein the first mixer is coupled to the receiver input by an input filter that filters all frequency components in the RF signal above an input cutoff frequency. . 92. The minimum frequency value of the first reference signal is greater than the input cutoff frequency.
The television receiver according to the above. 93. The second mixer mixes the second reference signal with a first filtered signal on the output of the first bandpass filter, thereby generating a second IF signal. 89. The television receiver of claim 88. 94. The television receiver according to claim 93, wherein the second operating frequency is fixed. 95. The television receiver of claim 94, wherein the selected carrier frequency in the RF signal appears at a second predetermined frequency in the second IF signal. 96. The television receiver according to claim 94, wherein the minimum frequency value of the first reference signal is higher than an input cutoff frequency of the RF signal. 97. A receiver input coupled to an RF signal source, a first input coupled to the receiver input, and a second input coupled to a first reference signal having a first operating frequency. And wherein the first operating frequency is variable between a minimum frequency value and a maximum frequency value; anda first mixer configured in the same substrate as the first mixer; and A first bandpass filter coupled to an output of the mixer, and a second reference signal having a first input and a second operating frequency coupled to an output of the first bandpass filter. A second mixer having a second input and wherein the second operating frequency is fixed. 98. The television receiver according to claim 97, wherein the minimum frequency value is higher than an input cutoff frequency of an RF signal output from the RF signal source. 99. The first mixer subtractively mixes the RF signal and the first reference signal, thereby generating a first IF signal, and wherein a selected carrier frequency in the RF signal is 100. The television receiver of claim 98, wherein the television signal appears at a first predetermined frequency in the first IF signal, and the first predetermined frequency is higher than the selected carrier frequency. 100. The method of claim 100, wherein the selected carrier frequency is
100. The television receiver of claim 99, wherein the television receiver appears at the first predetermined frequency in an output passband signal of the bandpass filter. 101. The second mixer mixes the output passband signal with the second reference signal, thereby generating a second IF signal, and wherein the selected carrier frequency is The television receiver of claim 100, wherein the television receiver appears in the second IF signal at a predetermined frequency, and the second predetermined frequency is lower than the first predetermined frequency and the second operating frequency. . 102. mixing the RF signal with a first reference signal having a first operating frequency, whereby the first I
Generating an F signal; filtering the first IF signal in a first bandpass filter to generate an output passband signal having at least four channels; Mixing with a second reference signal having a frequency, thereby generating a second IF signal. 103. The method of claim 102, further comprising filtering all frequency components above the input cutoff frequency from the RF signal. 104. The method according to claim 103, wherein the first operating frequency is higher than the input cutoff frequency. 105. The method of claim 104, wherein mixing the RF signal with the first reference signal is a subtractive mixing of the RF signal and the first reference signal. 106. The step of mixing the RF signal with the first reference signal is up-conversion mixing, and the selected carrier frequency in the RF signal is at a first predetermined frequency in the first IF signal. 106. The method of claim 105, wherein the appearing and first predetermined frequency is higher than the selected carrier frequency. 107. The method of claim 1, wherein the first predetermined frequency appears in an output passband signal of the first bandpass filter.
06. The method of item 06. 108. Filter all frequency components above the input cutoff frequency from the received RF signal, thereby:
Generating a first filtered signal; amplifying the first filtered signal; subtractively mixing the amplified first filtered signal with a first reference signal having a first operating frequency; Thereby, generating a first IF signal; filtering the first IF signal in a bandpass filter to generate an output passband signal having at least four channels; Mixing with a second reference signal having a second operating frequency, thereby generating a second IF signal. 109. The method of claim 108, wherein the first operating frequency is higher than the input cutoff frequency. 110. The step of mixing the amplified first filtered signal with a first reference signal is up-mixing, and the selected carrier signal in the received RF signal is at a first predetermined frequency. 110. The method of claim 109, wherein the first predetermined frequency appears in a first IF signal and is higher than a selected carrier frequency. 111. The selected carrier signal appears at an output passband signal at a first predetermined frequency.
The method described in. 112. The step of mixing the output passband signal with the second reference signal is down-conversion mixing, and the selected carrier signal appearing at the first predetermined frequency in the output passband signal is a second At a given frequency, the second I
109. The method of claim 108, wherein the second predetermined frequency appears in an F signal and the second predetermined frequency is lower than the first predetermined frequency. 113. The method of claim 108, further comprising filtering the second IF signal with a second bandpass filter. 114. A second band pass filter comprising:
114. The method of claim 113, having a passband between megahertz and 21 megahertz. 115. A second band pass filter comprising:
115. The method of claim 114, wherein the method is a Nyquist slope filter having a megahertz cutoff frequency. 116. A method for extracting a single communication channel from a number of such channels, wherein each channel has a defined frequency range and the plurality of channels are communicating within upper and lower frequency boundaries, comprising: Removing all frequencies greater than the frequency boundary from the input signal containing all of the channels of the input signal, and removing all of the channels contained within the input signal by a desired single channel to be extracted from the input signal. Converting to a conversion frequency range so as to be located in the center of the range, and the conversion frequency range being located at a frequency higher than the upper frequency boundary; and four or more channels in each frequency direction from the center of the conversion frequency range Removing all stretched channels from the transformed input signal, and the remaining channels in the input signal. To a second transform frequency range, such that the desired single channel to be extracted from the input signal is defined as having a lower frequency boundary defined as high enough not to cause phase distortion in any subsequent filtering. And a filter having the appropriate bandwidth, noise and distortion to be centered within a second transform frequency range having a high-end frequency boundary defined as exceeding that frequency that can be integrated into the substrate. And removing any remaining frequencies that are not part of the desired single channel. 117. The step of removing the remaining frequencies includes a receiver attenuation characteristic such that the relative values of the low and high frequencies in the desired single channel are modified for vestigial sideband demodulation. 117. The method of claim 116. 118. The low end and high end boundary of the second transform frequency range is defined by an integrated circuit continuous time filter capable of achieving single channel TV signal extraction and receiver attenuation characteristics. the method of.