JPH10302958A - High frequency heating device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To minimize a current flowing in the first semiconductor switching element by forming an oscillator for which the ON-time ratio of a pulse signal to drive the first semiconductor switching element and the second semiconductor switching element can be set. SOLUTION: An oscillator 11 is formed is a such manner that the ON-time ratio of a pulse signal to drive the first semiconductor switching element 3 and the second semiconductor switching element 6 can be set. In this way, the ON-time ratio can be set to realize zero voltage operation at all times for pulsating DC voltage. Input voltage detecting means 13, output command means 16 and reference voltage generating means 15 are provided, and the reference voltage generating means 15 gives a reference signal to the oscillator 11 in accordance with the output command means 16 and the input voltage detecting means 13 so that the ON-time ratio can be set to realize the zero voltage operation at all times for pulsating DC voltage.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power supply for driving a magnetron of a high-frequency heating device using a magnetron.

[0002]

2. Description of the Related Art A conventional high-frequency heating apparatus will be described with reference to the drawings.

As shown in FIG. 15, a conventional high-frequency heating apparatus uses a circuit configuration called a one-piece voltage resonance type circuit. 19 is a 60 Hz or 50 Hz commercial power supply, 2
Reference numeral 0 denotes a rectifier for full-wave rectification of the commercial power supply, which rectifies the commercial power supply to form a DC power supply. 21 is a smoothing capacitor, 22 is a first capacitor, 23 is a leakage type transformer, 24
Is a semiconductor switching element, 25 is a magnetron, 26
Is a drive circuit including an oscillator for driving the semiconductor switching element 24. The leakage type transformer 23 and the first capacitor 22 form a resonance circuit, and by this operation, the voltage waveform of the semiconductor switching element 24 becomes a sine wave. The collector voltage and current waveform of the semiconductor switching element 24 are as shown in FIGS.

The effect of this resonance circuit is that the current starts to flow after the voltage becomes zero, so that the switching loss at the time of ON (the portion where the voltage and the current overlap) is reduced. At the time of off, the current is sharply cut off, but since the voltage rises in a sine wave shape, the slope becomes gentle and the switching loss at the time of off is reduced. Thus, the resonance type circuit has an effect of reducing the switching loss of the semiconductor switching element. However, in the high-frequency heating device having such a configuration, the voltage applied to the semiconductor switching element 24 has a resonance voltage waveform as shown in FIG. I will. In addition, the period during which the semiconductor switching element 24 is off depends on the resonance period determined by the primary winding of the leakage type transformer 23 and the first capacitor 22. It is necessary to detect the timing at which the voltage applied to the terminal becomes zero and supply an ON signal. For this reason, there has been a problem that the control circuit is complicated.

To solve this problem, a high-frequency heating apparatus shown in FIG. 17 was considered prior to the present invention.

In FIG. 17, 1 is a DC power supply and 4 is a first power supply.
, 5 is a second capacitor, 6 is a second semiconductor switching element, 3 is a first semiconductor switching element, 7 is a drive circuit, 2 is a leakage type transformer, 8 is a full wave including a capacitor 8a and a capacitor 8b. The voltage doubler rectifier circuit 9 is a magnetron.

The drive circuit 7 includes an oscillator 11 for generating drive signals for the first semiconductor switching element 3 and the second semiconductor switching element 6 therein. A signal of a predetermined frequency and a duty is generated by the oscillator 11, and a driving signal is given to the first semiconductor switching element 3. The second semiconductor switching element 6 is provided with a signal obtained by adding a delay time by the delay circuit 12 to an inverted signal of the drive signal of the first semiconductor switching element 3.

The operation of this circuit is shown in FIGS.
This will be described with reference to FIG. First, when the first semiconductor switching element 3 is turned on, an equivalent circuit of a main circuit part after the DC power supply 1 is as shown in FIG. 18A, and the collector current Ic is changed to the primary winding of the leakage type transformer 2. It is supplied from the second capacitor 5 through the wire (FIG. 19 (a) state (a)). At this time, the secondary side output of the leakage type transformer 2 starts charging the capacitor 8a of the full wave voltage doubler rectifier circuit. Since the initial voltage V8a is stored in the capacitor 8b, when the voltage V8b of the capacitor 8a is in the relationship of V8a + V8b> Vcut (1) Vcut: cut-off voltage of the magnetron, the magnetron 9 can be oscillated and the magnetron 9 can be oscillated. The current Ia starts to flow as shown in FIG.

When the first semiconductor switching element 3 is turned off, the equivalent circuit becomes as shown in FIG. 18B, and the current flowing through the primary winding of the leakage type transformer 2 is directed toward the first capacitor 4. Start flowing. At this time, the secondary output of the leakage transformer 2 starts charging the capacitor 8b (state (b)). At this time, when the expression (1) is satisfied, the magnetron 9 starts oscillating again, and the anode current Ia starts to flow. In addition, 1 of leakage type transformer 2
The current of the next winding is as shown in FIG. 19C, and the voltage of the first semiconductor switching element 3 is as shown in FIG.

When the voltage of the first semiconductor switching element 3 reaches the initial voltage of the second capacitor 5, the second
Of the semiconductor switching element 6 is turned on, and charging of the second capacitor 5 is started. The equivalent circuit at this time is as shown in FIG. Since the second capacitor 5 has a larger capacitance value than the first capacitor 4, the first semiconductor switching element 3
The voltage gradient rapidly becomes gentle and the state shifts to the state (c) in FIG. The current flowing from the primary side of the leakage type transformer 2 to the second capacitor 5 is:
Conversely, when the current flows from the second capacitor 5 toward the first winding of the leakage type transformer 2, the state (d) is entered. At this point, it is necessary to turn on the transistor constituting the second semiconductor switching element 6. When the transistor forming the second semiconductor switching element 6 is cut off at an arbitrary time T, the state shifts to a state (e) in which a current starts to flow from the first capacitor 4 toward the first winding of the leakage type transformer 2. . At this time, the slope of the voltage of the first semiconductor switching element 3 becomes steep, and decreases toward zero due to the energy of the first capacitor 4. When the first semiconductor switching element 3 is driven again when this voltage becomes zero, the same operation is repeated from the state (a).
A switching operation for reducing switching loss can be realized.

The initial voltage of the above-mentioned second capacitor is:
It is determined by turning on the second semiconductor switching element 6 for an arbitrary time T in the state (d). That is, by increasing the ON time of the second semiconductor switching element 6, the initial voltage of the second capacitor 5 decreases, and as a result, the voltage of the first semiconductor switching element 3 can be reduced. As described above, the first semiconductor switching element 3 which cannot be realized by the conventional circuit configuration
, In other words, the on-time of the second semiconductor switching element 6 can be set arbitrarily, and the capacity of the second capacitor 5 is set to be sufficiently larger than the capacity of the first capacitor 4. By taking the value
The voltage of the first semiconductor switching element 3 can be reduced (clamped).

[0012]

However, the high-frequency heating device having such a configuration has the following problems.

That is, a commercial power supply is rectified by a rectifier.
The magnitude of the pulsation that is hardly smooth as shown in FIG.
Circuit operation at a low voltage, and the operating frequency of the oscillator
Since the frequency is fixed at a predetermined frequency, the first semiconductor
Switching element 3 and second semiconductor switching element 6
Operation is constant over the entire power supply voltage range
Becomes Driving at a constant on-time in this way will
The time when the tron 9 starts or stops oscillating (FIG. 20)
(D) near T4 or T6).
Tends to be difficult. For this reason, in the entire range of the power supply voltage
Power supply peak to achieve zero voltage switching
Near (around T5 in FIG. 20 (d))
With enough energy stored in the first capacitor 4
The second semiconductor switching element 6 must be turned off
Must. For this reason, as shown in FIG.
That flows through the diode constituting the body switching element 3
Style I _DBecomes larger and the first power conversion when performing predetermined power conversion is performed.
The current duty of the semiconductor switching element 3 increases. Ma
In addition, the power that is converted every cycle when the power supply voltage is at the maximum value
At this time, the current flowing through the magnetron 9 at this time is
Current Ia rises, momentarily exceeds the specified value,
There is a problem that it has a bad influence on the service life of the housing 9.

[0014]

A DC power supply obtained by rectifying a commercial power supply, a leakage transformer connected to the DC power supply, and a first transformer connected in series to a primary winding of the leakage transformer. 1 semiconductor switching element;
A first capacitor connected in parallel or in series with the primary winding of the leakage type transformer, and a second capacitor connected in parallel or in series with the primary winding of the leakage type transformer; A series connection of semiconductor switching elements, a driving circuit having an oscillator for driving the first semiconductor switching element and the second semiconductor switching element, a rectifier circuit connected to a secondary side of the leakage transformer, A magnetron connected to the rectifier circuit, an input voltage detecting means, an output command means, and a reference voltage generating means are provided, and the reference voltage generating means is based on the outputs of the output command means and the input voltage detecting means. The oscillator is configured to apply a reference signal, and the oscillator is configured to connect the first semiconductor switching element and the second semiconductor switching element based on the reference signal. By so setting the pulse signal for driving the semiconductor switching element of the on-time ratio, the first semiconductor switching element is time to turn on the change with the change in the supply voltage. Therefore, it is possible to set the on-time ratio so as to realize the zero-voltage operation at all times of the pulsating DC voltage, so that the current flowing through the first semiconductor switching element can be minimized.

Further, the oscillator includes frequency setting means,
Based on the output signal of the frequency setting means, the oscillation frequency of the oscillator is changed, and the lowest frequency limiting means is provided.The oscillator is configured to limit the lowest frequency by the lowest frequency limiting means. It will not be in the audible frequency band.

[0016]

DESCRIPTION OF THE PREFERRED EMBODIMENTS A DC power supply obtained by rectifying a commercial power supply, a leakage transformer connected to the DC power supply, and a first transformer connected in series to a primary winding of the leakage transformer. A semiconductor switching element, a first capacitor connected in parallel or in series with the primary winding of the leakage type transformer, and a second capacitor connected in parallel or in series with the primary winding of the leakage type transformer A series connection of a capacitor and a second semiconductor switching element, a driving circuit having an oscillator for driving the first semiconductor switching element and the second semiconductor switching element, and a driving circuit connected to a secondary side of the leakage transformer. Rectifier circuit, and a magnetron connected to the rectifier circuit, wherein the oscillator includes the first semiconductor switching element and the second semiconductor switching element. By configuring so that the on-time ratio of the pulse signal for driving the body switching element can be set, the on-time ratio can be set so as to realize zero-voltage operation at all times of the pulsating DC voltage. In addition, the current flowing through the first semiconductor switching element can be minimized.

Further, the oscillator is provided with frequency setting means, and the oscillating frequency of the oscillator is changed based on the output signal of the frequency setting means, thereby realizing zero voltage operation at all times of the pulsating DC voltage. Since the ON time ratio can be set to such a value, the current flowing through the first semiconductor switching element can be minimized.

Further, input voltage detecting means, output command means, and reference voltage generating means are provided, and the reference voltage generating means sends a reference signal to an oscillator based on the output of the output command means and the input voltage detecting means. With this configuration, the ON time ratio can be set so as to realize the zero voltage operation at all times of the pulsating DC voltage, so that the current flowing through the first semiconductor switching element is minimized. be able to.

Further, the frequency setting means is configured to change the output based on the output of the input voltage detecting means so as to set the ON time ratio so as to realize the zero voltage operation at all times of the pulsating DC voltage. Therefore, the current flowing through the first semiconductor switching element can be minimized.

[0020] Further, a minimum frequency limiting means is provided, and the oscillator is configured such that the minimum frequency is limited by the minimum frequency limiting means. , The current flowing through the first semiconductor switching element can be minimized, and the operating frequency does not fall below the audible frequency band.

Hereinafter, embodiments of the present invention will be described with reference to the drawings. (Embodiment 1) FIG. 1 is a circuit diagram of a magnetron driving power supply according to an embodiment of the present invention. In FIG. 1, components having the same reference numerals as those of the conventional example are the same components, and a detailed description thereof will be omitted.

FIG. 2 shows that the series connection of the second semiconductor switching element 6 and the second capacitor 5 is connected in series to the primary winding of the leakage type transformer 2 and the first capacitor 4 is connected to the leakage. This is an example in which the primary winding of the type transformer 2 is connected in parallel, and the circuit operation is shown in FIG.
(B) The operation modes are divided as shown in (c) and have the same operation as the example of FIG.

On the other hand, FIG. 4 shows that the series connection of the second semiconductor switching element 3 and the second capacitor 5 is connected in parallel to the primary winding of the leakage type transformer 2,
5 is connected in series to the primary winding of the leakage type transformer 2 and the circuit operation is shown in FIG.
The operation modes are divided as shown in (a), (b), and (c), and have the same operation as the example of FIG. 1 or FIG.

FIG. 6 shows that the series connection of the second semiconductor switching element 6 and the second capacitor 5 is connected in parallel to the primary winding of the leakage type transformer 2 and the first capacitor 4 is connected to the leakage type. This is an example in which the primary winding of the transformer 2 is connected in parallel, and the circuit operation is shown in FIG.
(B) The operation mode is divided as shown in (c), and FIG.
Has the same action as the example of

The input voltage detecting means 13 rectifies the voltage of the commercial power supply 1 by a diode and generates a voltage synchronized with the pulsation of the power supply voltage. The inverting amplifier circuit 14 inverts and amplifies this voltage and supplies a pulsating signal synchronized with the power supply cycle to the reference signal generating circuit 15. The reference signal generating means 15 superimposes a pulsating signal synchronized with the power supply cycle output from the inverting amplifier circuit 14 on the DC value given from the output command means 16 and supplies the superimposed signal to the oscillation circuit 11 as a reference signal.

FIG. 8 is a circuit diagram showing a detailed configuration of the oscillator 11. The transistor 11a passes a predetermined current I1 determined by the value of the resistor 11h. Transistor 11
Since b and 11c form a current mirror circuit with the transistor 11a, the same current as I1 flows. When the output of the flip-flop circuit 11i is Low, the transistors 11d, 11e, and 11f are in the off state.
The current flowing through the transistor 11b becomes a current for charging the capacitor 11g, and the voltage V3 of the capacitor 11g increases linearly. Next, when V3 reaches the first reference voltage V1, the comparator 11j switches to the flip-flop circuit 11i.
And the output of the flip-flop circuit is Hig
h. At this time, the transistor 11d is turned on, so that the same current I1 as the current flowing through the transistor 11c flows through the transistor 11d. Also,
Since the transistors 11e and 11f form a current mirror with the transistor 11d, the transistors 11e and 11f try to pass the same current as the current I1, respectively. Since the current flowing through the transistor 11b is I1, a current is supplied from the capacitor 11g to flow the current I1 to the transistors 11e and 11f. As a result, the capacitor 11g is discharged by the current I1, and the voltage V3 decreases linearly. Next, when V3 falls to the second reference voltage V2, the comparator 1
1k sends a signal to the flip-flop circuit 11i, and as a result, the output of the flip-flop circuit 11i becomes low. At this time, since the transistor 11d is turned off again, the capacitor 11g is charged and V3 rises. By repeating the above operation, a triangular wave having a constant frequency is formed in V3. The comparator 111 compares the triangular wave with the reference signal provided from the reference signal generating means 15 to generate a pulse signal.

FIG. 9A shows a voltage waveform of a commercial power supply.
It changes at a frequency of 60 Hz or 50 Hz. FIG. 3B shows a current waveform input from a commercial power supply. FIG. 3C shows a DC voltage waveform obtained by full-wave rectifying a commercial power supply.
This is a waveform having a large ripple that fluctuates in synchronization with the cycle of the commercial power supply. FIG. 4D shows the waveform of the current Ia flowing through the magnetron 9. As described above, when the output voltage of the high-voltage rectifier circuit 8 exceeds the cutoff voltage of the magnetron 9, the current starts flowing. FIG. 3E is a diagram showing the relationship between the triangular wave generated by the oscillator 11 and the reference signal output from the reference signal generating means 15. The reference signal output from the reference signal generating means 15 is synchronized with the commercial power supply, and becomes minimum at time T2 when the commercial power supply has the maximum value. FIG. 10 is a diagram showing the output pulse signal of the oscillator, the current flowing through the first semiconductor switching element, and the applied voltage at each of the times T1 to T3. Since the triangular wave generated by the oscillator 11 and the reference signal generated by the reference signal generating means 15 have the above-described relationship, the time width of the output pulse signal of the oscillator 11 at each time has the following relationship.

T1> t2 <t3 (2) Here, considering the transmission power per cycle, the following equation is obtained.

[0029] ^{P = E 2 × t 2/} 2 × L × T (3) E; supply voltage T; 1 cycle of time t; criteria as in the first time this embodiment the semiconductor switching element is on By changing the output signal of the signal generation means 15 in synchronization with the power supply voltage, the transmission at the peak point (time T2) of the commercial power supply is compared with when the first semiconductor switching element is driven for a fixed ON time. Power can be reduced. On the other hand, at a time (time T1 or T3) that deviates from the peak point, the transmission power can be increased. Therefore, even at the minimum time width t1 required for the first semiconductor switching element to perform the zero-voltage switching operation at the time T2, the zero-voltage operation can be stably performed in the entire region of the power supply voltage. Current flowing through the diode constituting the semiconductor switching element can be minimized. For this reason, there is an effect that the current duty of the first semiconductor switching element can be minimized. Further, since the transmission power at the peak point is suppressed, the current flowing through the magnetron is as shown in FIG. 9C, and the maximum value can be suppressed within the standard value of the magnetron.

(Embodiment 2) In FIG. 11, the same components as those in FIG. 1 are denoted by the same reference numerals. A combination of the series connection of the second semiconductor switching element 6 and the second capacitor 5 and the connection of the first capacitor 4 can be considered in the same manner as in the first embodiment.
Since they have the same function as described above, their description is omitted in this embodiment.

The oscillator 11 has frequency setting means, and acts so as to change the oscillation frequency of the oscillator 11 according to the output of the power supply voltage detecting means. FIG. 12 is a circuit diagram showing a detailed configuration of the oscillator. Components having the same functions as those in FIG. 8 are denoted by the same reference numerals. Transistor 11
Since m is configured to input the output signal of the power supply voltage detecting means to the base terminal, the voltage V4 applied across the resistor 11h is V4 = V5-V _BE (4) V _BE ; the voltage between the base and the emitter of the transistor 11m. . The following relationship holds between the currents I1 and V4 flowing through the transistor 11a: I1 = V4 / R (11h) (5) R (11h); Therefore, by varying V5, the charging / discharging current of the capacitor 11g changes, and the frequency of the triangular wave changes accordingly. Therefore, the frequency of the pulse signal output from the oscillator 11 also changes accordingly. On the other hand, the transmission power per cycle is indicated by (3) as described above. If this is expressed by the time ratio D during which the first semiconductor switching element is turned on during one cycle and the operating frequency, P = E ² × D ² / (2 × F) (6) D; the first semiconductor switching element Is on: D = t / TF; frequency F = 1 / T. Since the frequency setting means is configured to set the operating frequency by the output of the power supply voltage detecting means, it works to reduce the operating frequency as the power supply voltage deviates from the maximum value. Therefore, the transmission power at a time that deviates from the maximum value of the power supply voltage can be increased. Therefore, the zero-voltage switching operation can be easily realized in the entire range of the power supply voltage. In addition, the transmission power at a time outside the maximum power supply voltage can be increased, so the transmission power at the maximum power supply time can be suppressed, and the maximum value of the current flowing through the magnetron can be suppressed. Can be. Therefore, the current flowing through the magnetron can be suppressed within the standard value.

(Embodiment 3) In FIG. 13, the same components as those of the above embodiment are denoted by the same reference numerals. Also,
As in the first or second embodiment described above, a combination of the series connection of the second semiconductor switching element 6 and the second capacitor 5 and the connection of the first capacitor 4 can be considered. Therefore, the description is omitted in the present embodiment. The oscillator 11 has a minimum frequency limiting means, and acts to limit the lower limit of the frequency so that the oscillation frequency of the oscillator does not fall below a predetermined value. FIG. 14 is a detailed circuit diagram of the oscillator 11, and the same components as those in the above-described embodiment are denoted by the same reference numerals. The transistor 11n is turned on when the base voltage of the transistor 11m is about to become equal to or lower than V6, and between the voltages V6 and V5 determined by the resistors 11o and 11p, V5 = V6- _{VBE (11n)} (7) (where V6> V5) V _{BE (11n)} ; base-emitter voltage of the transistor 11n Therefore, the base voltage of the transistor 11m does not become lower than the value represented by the equation (7). Therefore, the minimum value of the voltage V4 of the resistor 11h is also limited, and the transistor 1
The minimum value of the current I1 flowing through 1a is similarly limited. Therefore, the current for charging and discharging the capacitor 11g is I1
Therefore, the minimum value is limited. As a result, the lowest frequency of the triangular wave generated by the oscillator 11 is limited. This translates into first and second
The minimum value of the frequency at which the semiconductor switching element is turned on and off is limited. Therefore, by selecting the value of I1 and the capacity of the capacitor 11g so that this frequency does not fall below the audible frequency constant of 20 kHz, the first
In addition, it is possible to prevent a switching sound generated when the second semiconductor switching element is turned on and off from being heard.

[0033]

As described above, according to the present invention, a DC power supply obtained by rectifying a commercial power supply, a leakage type transformer connected to the DC power supply, and a first winding of the leakage type transformer are provided. A first semiconductor switching element connected in series to a first winding of the leakage type transformer; a first capacitor connected in parallel or in series with the first winding of the leakage type transformer; A drive circuit having an oscillator for driving the first semiconductor switching element and the second semiconductor switching element, and a series connection of a second capacitor and a second semiconductor switching element connected in parallel or in series; A rectifier circuit connected to a secondary side of the leakage transformer; and a magnetron connected to the rectifier circuit, wherein the oscillator is configured to control the first semiconductor switch. By setting the ON time ratio of the pulse signal for driving the switching element and the second semiconductor switching element to be set, the ON time ratio is set so as to realize the zero voltage operation at all times of the pulsating DC voltage. Therefore, the current flowing through the first semiconductor switching element can be minimized.

Further, the oscillator includes frequency setting means,
Since the oscillation frequency of the oscillator is changed based on the output signal of the frequency setting means and the lowest frequency limiting means is provided, and the oscillator is configured to limit the lowest frequency by the lowest frequency limiting means, the operating frequency is audible. There is an effect that the frequency band does not occur and noise accompanying the switching of the semiconductor switching element can be made inaudible.

[Brief description of the drawings]

FIG. 1 is a circuit diagram of a power supply for driving a magnetron of a high-frequency heating device according to a first embodiment of the present invention.

FIG. 2 is a circuit diagram of another magnetron driving power supply of the high-frequency heating device.

FIG. 3 (a) shows a first example of the magnetron driving power supply.
(B) Equivalent circuit diagram when the first semiconductor switching element is off in the same magnetron driving power supply. (C) Second capacitor is charged in the same magnetron driving power supply. Equivalent circuit diagram

FIG. 4 is a circuit diagram of another magnetron driving power supply of the high-frequency heating device.

FIG. 5 (a) shows a first example of the magnetron driving power supply.
(B) Equivalent circuit diagram when the first semiconductor switching element is off in the same magnetron driving power supply. (C) Second capacitor is charged in the same magnetron driving power supply. Equivalent circuit diagram

FIG. 6 is a circuit diagram of another magnetron driving power supply of the high-frequency heating device.

FIG. 7A shows a first example of the magnetron driving power supply.
(B) Equivalent circuit diagram when the first semiconductor switching element is off in the same magnetron driving power supply. (C) Second capacitor is charged in the same magnetron driving power supply. Equivalent circuit diagram to start

FIG. 8 is a detailed circuit diagram of an oscillator of a power supply for driving a magnetron of the present invention.

FIG. 9 is an operation waveform diagram of a main circuit portion of the power supply for driving the magnetron.

FIG. 10 is a waveform chart showing output pulses of an oscillator and currents and voltages of a first semiconductor switching element at times T1, T2, and T3 of the magnetron.

FIG. 11 is a circuit diagram of a power supply for driving a magnetron of a high-frequency heating device according to a second embodiment of the present invention.

FIG. 12 is a detailed circuit diagram of the oscillator of the power supply for driving the magnetron.

FIG. 13 is a circuit diagram of a power supply for driving a magnetron of a high-frequency heating device according to a third embodiment of the present invention.

FIG. 14 is a detailed circuit diagram of the oscillator of the power supply for driving the magnetron.

FIG. 15 is a circuit diagram of a power supply for driving a magnetron of a conventional high-frequency heating device.

FIG. 16 is a waveform chart showing voltage and current of a semiconductor switching element of the power supply for driving the magnetron.

FIG. 17 is a circuit diagram of a power supply for driving a magnetron of another conventional high-frequency heating device.

18A is an equivalent circuit diagram when the first semiconductor switching element is turned on in the magnetron driving power supply. FIG. 18B is an equivalent circuit diagram when the first semiconductor switching element is turned off in the magnetron driving power supply. (C) Equivalent circuit diagram when the second capacitor starts charging in the magnetron driving power supply

FIG. 19 is a waveform diagram of a main circuit portion of the power supply for driving the magnetron.

FIG. 20 is an operation waveform diagram of a main circuit portion of the power supply for driving the magnetron.

FIG. 21 shows times T4 and T of the magnetron driving power supply.
5, Waveform diagrams showing the output pulse of the oscillator at T6 and the current and voltage of the first semiconductor switching element

[Explanation of symbols]

 Reference Signs List 1 DC power supply 2 Leakage type transformer 3 First semiconductor switching element 4 First capacitor 5 Second capacitor 6 Second semiconductor switching element 7 Drive circuit 8 Rectifier circuit 9 Magnetron 11 Oscillator 13 Input power detection means 15 Reference signal generation Means 16 Output command means 17 Frequency setting means 18 Minimum frequency limiting means

 ──────────────────────────────────────────────────の Continued on the front page (72) Inventor Haruo Suenaga 1006 Kazuma Kadoma, Kadoma, Osaka Prefecture Inside Matsushita Electric Industrial Co., Ltd.

Claims (5)

[Claims] 1. A DC power supply obtained by rectifying a commercial power supply,
A leakage type transformer connected to the DC power supply, a first semiconductor switching element connected in series to a primary winding of the leakage type transformer, and connected in parallel or series to a primary winding of the leakage type transformer A first capacitor to be connected, a series connection of a second capacitor and a second semiconductor switching element connected in parallel or in series to the primary winding of the leakage type transformer, A rectifier circuit connected to a secondary side of the leakage type transformer, and a magnetron connected to the rectifier circuit, wherein the oscillator is the first semiconductor. The on-time ratio of a pulse signal for driving the switching element and the second semiconductor switching element can be set. High-frequency heating apparatus. 2. The high-frequency heating apparatus according to claim 1, wherein the oscillator includes frequency setting means, and the oscillation frequency of the oscillator is changed based on an output signal of the frequency setting means. 3. An input voltage detecting means, an output commanding means, and a reference voltage generating means, wherein the reference voltage generating means sends a reference signal to an oscillator based on the output of the output commanding means and the input voltage detecting means. Claim 1 or Claim 2
The high-frequency heating device as described. 4. The high-frequency heating apparatus according to claim 3, wherein the frequency setting means changes the output based on the output of the input voltage detecting means. 5. The high-frequency heating apparatus according to claim 2, further comprising a minimum frequency limiting means, wherein the oscillator limits the minimum frequency by said minimum frequency limiting means.