JPH10275678A - High frequency heating device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To make a current peak value in a forward period and a flyback period of power transmission from a full-wave voltage double rectifier circuit on a secondary side uniform by alternately driving and controlling a first semiconductor switching device arranged in series to the primary side of a leakage type transformer and a second semiconductor switching device connected in series to a second capacitor arranged in parallel. SOLUTION: When a first semiconductor switching means 38 is turned on, the secondary side output of a leakage type transformer 7 starts charging a second high voltage capacitor 59 of a full-wave double rectifier circuit 57. When the total voltage of first and second capacitors 58, 59 exceeds the cut-off voltage of a magnetron 12, current flows in the magnetron 12. When the first switching means 38 is turned on, current flows in a first capacitor 54, and charging of the first high voltage capacitor 58 starts. When the voltage of the first switching means 38 reaches the initial voltage of a second capacitor 55, a second switch 39 is turned on, and charging of the second capacitor 55 starts.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power supply for driving a magnetron of a high-frequency heating device using a magnetron.

[0002]

2. Description of the Related Art As shown in FIG. 9, a conventional high-frequency heating apparatus uses a circuit configuration called a one-piece voltage resonance type circuit. 1 is a commercial power supply, 2 is a rectifier composed of full-wave rectification, which rectifies the commercial power supply to form a DC power supply. The choke coil 3 smoothes the fluctuation of the current, and the smoothing capacitor 4 smoothes the fluctuation of the voltage. These inductors and capacitors form a low-pass filter circuit. Energy is accumulated in the winding of the leakage type transformer 14 from the DC power supply at the point (a). The primary winding 13b of the leakage transformer 14 and the resonance capacitor 6 arranged in parallel with it constitute a tank circuit, and resonate with the energy stored in the winding of the leakage transformer 14. Switching means comprising a transistor 5 and a diode 8 arranged in parallel with the transistor 5 are connected in series to the tank circuit.

The secondary side of the leakage type transformer 14 is
A secondary winding 13a and a heater winding 13c are provided. The voltage induced in the secondary winding 13c is the first high voltage capacitor 1
0, a second high-voltage capacitor 11, a first high-voltage diode 18, and a second high-voltage diode 19, which are rectified by a full-wave voltage doubler rectifier circuit and apply a DC high voltage of about -4 kV to the cathode of the magnetron 12. It has become.

A current transformer 15 is a magnetron 12
Is detected, and the information is insulated and returned to the inverter control circuit 9 having the power control circuit. The inverter control circuit 9 controls the on time of the magnetron 5 so that the signal from the current transformer 15 becomes constant, and controls the power transmitted to the secondary side. In this control law, since the anode current is controlled at a constant value,
The responsibility of the anode current, which will cause serious damage to the magnetron if it increases excessively, can be controlled to an appropriate level,
This is a very advantageous control law in terms of magnetron quality and reliability.

Next, a mechanism for driving the magnetron 5 and supplying power to the secondary side will be described with reference to FIG. The upper time chart shows a signal relationship for generating a pulse signal (e) for driving the magnetron 5 with a certain set high-frequency output, and the lower time chart shows the pulse train.

The upper triangular waveform is sliced by the high frequency output reference signal (f). When the triangular wave (g) signal is lower than (f), a high signal is output, and conversely, a low signal is output. The respective times are t1 and t2, and when t1 is long, the energy stored in the primary winding 13b increases. The operating frequency f is 1 / (t1 + t2).
The vertex of the triangular wave (g) crosses the points (a) and (b) in FIG. 12 and generates a cross trigger under the conditions of (a)> (b) to (b) <(b), and the slope is Turns down. Then, when it crosses the inversion threshold value (c), the mode is naturally changed, and the slope changes to a rising slope. By repeating this, a triangular wave (g) is formed.

Further, the specific circuit configuration and operation will be described with reference to FIG. Point (a) is a complete direct current when the circuit is not operating, but once operated and power is consumed on the secondary side, it has a waveform obtained by full-wave rectification of the sine shape of the power supply. When the effective voltage E is E = 100 V, the peak value is about 141 V when E = 100 V. On the other hand, point (b) is a resonance voltage waveform generated in the above-described tank circuit, which is the voltage waveform of the semiconductor switching means in FIG. 13 (d). Big. Returning to FIG. 11, reference numeral 26 denotes a comparison timing circuit, and a voltage obtained by dividing the point (a) by the resistors 22 and 23;
(B) Compare the voltage with the voltage obtained by dividing the point by resistors 24 and 25,
The cross trigger described above is detected. The inversion threshold (C) is determined by dividing the circuit power supply voltage Vcc by the resistors 20 and 21. Now, when the transistor 31 is on, the electric charge of the capacitor 17 is discharged via the resistors 16 and 30, and the triangular wave (g) enters a falling slope state. Next, when the triangular wave (g) falls and crosses the inversion threshold (c), the output (d) of the comparison timing circuit 26 turns off the transistor 31 via the resistor 29, and the capacitor 17 becomes the resistor 16
And the triangular wave (g) turns to a rising slope. And it reverses again by the above-mentioned cross trigger and turns into a descending slope.

The triangular wave (g) is compared with a high-frequency output reference signal (f) of 28 by a comparator 27 to generate a pulse signal (e). The circuit configuration shown here exists inside the inverter control circuit 9 and the pulse signal drives the magnetron 5 through an appropriate driver circuit.

As described above, the reason for driving the magnetron 5 using the cross trigger and the comparison timing circuit 26 is as shown in FIG. 13 (b) when the transistor current is cut off (a) In addition to reducing the switching loss that occurs as a product of the current and the voltage by increasing the slope of the rise relatively slowly in a sinusoidal manner, the period during which the diode current flows with a predetermined delay from the cross trigger, that is, (a) the voltage By turning on the magnetron 5 when the waveform is 0 V, the switching loss is similarly reduced. This is a major feature of the voltage resonance system.

FIG. 12 is a time chart of waveforms of main parts. (A) is a current waveform of the semiconductor switching means,
(B) shows the current waveform of the magnetron, (c) shows the current waveform of the leakage type transformer, and (d) shows the voltage waveform of the semiconductor switching means. A current flows through the transistor during the period (I), and a resonance voltage of the tank circuit is generated during the period (II). The diode current flows during the period (III), and the transistor is turned on for this period. With such control, the switching loss of the transistor is minimized.

[0011]

However, the conventional magnetron drive power supply has the following phenomena when a single-pole voltage resonance type circuit is used.

(1) The resonance period is a constant period determined by the intrinsic constants L and C. On the other hand, the on-period of the semiconductor switching means, which is a non-resonance period, is arbitrarily determined by the power, and its on / off duty. The ratio differs depending on the design specifications (particularly the resonance circuit constant specifications) and the set output, and in the voltage resonance type, it is consequently determined that ON: OFF = about 2: 1 under conditions satisfying the performance.

(2) As a voltage transmitted by the transformer, a period during which the magnetron 5 is on is referred to as forward, and a period during which the magnetron 5 is turned off and the tank circuit resonates is referred to as flyback. In the forward period (t _FW ), The voltage became the power supply voltage, and during the flyback period (t _FB ), a waveform having a high peak value due to sharp resonance was formed as shown in FIG. 14A, and the positive and negative swings were extremely unbalanced. .

FIG. 14A specifically shows this.
The waveform of the secondary voltage of the transformer of t _FW > t _FB , V _FW
<The relationship of V _FB is clearly visible.

For example, the behavior of the rectifier circuit on the secondary side of the transformer at this time is as shown in FIG. Current I _FW through the first high-voltage diode 19 is charged to a second high-voltage capacitor 11 by the voltage V _FW induced in the secondary winding 13a. When there is a voltage V1 generated by that time had already been stored in the first high-voltage capacitor 10 _{charges, V1 + V FW> Vcut (} 1) Vcut; magnetron if the relationship of the magnetron of the cut-off voltage oscillates. The anode current at that time flows in such a manner that accumulated charges in the first high-voltage capacitor 10 and the second high-voltage capacitor 11 are discharged. Therefore, the second high-voltage capacitor 11 is supplied with electric charge from the primary side and the voltage is maintained, but the first high-voltage capacitor 10 only discharges and oscillates until Vcut or less. However, since the forward period is longer than the flyback period, it is necessary to use a large-capacity first high-voltage capacitor 10. Otherwise, the discharge is performed in a short period of time and becomes equal to or lower than Vcut, so that a small amount of anode current flows in a short period of time.

On the other hand, during the flyback period, FIG.
(C). Voltage V _FB induced in secondary winding 13b
As a result, the current of I _FB is charged in the first high-voltage capacitor 10 through the second high-voltage diode 18. Assuming that there is a voltage V2 generated by the electric charge already stored in the second high-voltage capacitor 11 at that time, the magnetron oscillates if V2 + _VFB > Vcut (2). The anode current at that time flows in such a manner that accumulated charges in the first high-voltage capacitor 10 and the second high-voltage capacitor 11 are discharged. Therefore, the first high-voltage capacitor 10 is supplied with electric charge from the primary side and the voltage is maintained, but the second high-voltage capacitor 11 is only discharged, and the oscillation continues until it becomes equal to or lower than Vcut. However, since the flyback period is shorter than the forward period, the first high-voltage capacitor CH1,
Does not need to be as large as CH2 as the second high-voltage capacitor 11, and is configured to allow sufficient anode current to flow by using a capacitor having a small capacity.

Therefore, within the range of the relationship C _H1 <C _H2 ,
In order for the average value of the anode current (a parameter for determining the high-frequency output) to be a predetermined value and the peak value of the anode current to be equal to or less than the absolute maximum rating, the first high-voltage capacitor 10 and the second high-voltage capacitor 11 Needs to be optimized unbalanced, and two capacitors of different capacitances are required, so that the current peaks in the forward period and the flyback period can be equalized. FIG. 14D shows the waveform of the actual anode current.
If both parts have the same lead pitch, it is necessary to assemble them while taking into account erroneous insertion. This has been a very serious problem in terms of management, manufacturing workability, and quality in view of erroneous insertion.

[0018]

SUMMARY OF THE INVENTION In order to solve the above-mentioned problems, the present invention provides a first semiconductor switching means connected in series to a primary side of a leakage type transformer connected to a DC power supply; A full-wave voltage doubler circuit comprising a first capacitor and a second capacitor, and a first high-voltage capacitor and a second high-voltage capacitor in parallel with the primary side of the transformer, wherein the second capacitor is connected in series. Second
Are connected. The power control circuit for driving the first semiconductor switching means and the second semiconductor switching means is connected to the secondary side of the leakage type transformer and controls the power transmission to the full wave voltage doubler circuit and the magnetron. Not only can the conduction time of the means be controlled, but also the conduction time of the second semiconductor switching means can be controlled. Accordingly, the forward period determined by the conduction time of the first semiconductor switching means and the other flyback periods are set at a predetermined ratio, and the peaks of the anode current during the forward and flyback periods are almost uniformly set. It is possible to

According to the above-mentioned invention, the values of the first high-voltage capacitor and the second high-voltage capacitor are set to the same capacity and the same type to reduce the number of component types and to improve the simplicity of the production management associated therewith. It is possible to prevent erroneous insertion (misplacement), which may occur when constants of different capacities are used, and it is possible to expect improvement in manufacturing quality.

[0020]

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a DC power supply, a leakage type transformer connected to the DC power supply, first semiconductor switching means connected in series to the primary side of the leakage type transformer, and a leakage type transformer. A first capacitor connected in parallel to the primary side of the leakage transformer; a second capacitor connected in parallel to the primary side of the leakage transformer;
A second semiconductor switching means connected in series to the capacitor, a power control circuit for mutually driving the first semiconductor switching means and the second semiconductor switching means, and a secondary side connected to the leakage type transformer. And a first high-voltage capacitor connected to the rectifier circuit, a second high-voltage capacitor connected to the rectifier circuit, and a magnetron connected to the rectifier circuit.

In order to control the power, the power control circuit changes the conduction time of the first semiconductor switching means and also changes the conduction time of the second semiconductor switching means, thereby controlling the ON period of the first semiconductor switching means. And the off-period is optimized to a predetermined ratio, so that the first high-voltage capacitor and the second high-voltage capacitor are of the same type having the same capacity, and the peak current of the anode current of the magnetron is substantially the same. Is realized.

Also, a DC power supply, a leakage type transformer connected to the DC power supply, first semiconductor switching means connected in series to the primary side of the leakage type transformer, and parallel to the primary side of the leakage type transformer. A first capacitor connected in parallel to the primary side of the leakage type transformer, a second semiconductor switching means connected in series to the second capacitor, A power control circuit for mutually driving the semiconductor switching means and the second semiconductor switching means, a full-wave voltage rectifier circuit connected to the secondary side of the leakage transformer, and a first rectifier circuit connected to the rectifier circuit. It comprises a high voltage capacitor, a second high voltage capacitor connected to the rectifier circuit, and a magnetron connected to the rectifier circuit.

When controlling the power, the power control circuit changes the conduction time of the first semiconductor switching means and optimizes the ON time and the OFF time of the conduction time of the second semiconductor switching means to a predetermined ratio. The peak current of the anode current of the magnetron is almost the same, and
An object of the present invention is to realize a high-frequency heating device in which the first high-voltage condenser and the second high-voltage condenser are of a predetermined capacity and use a plurality of ones of the same type or are used alone.

Also, a DC power supply, a leakage type transformer connected to the DC power supply, first semiconductor switching means connected in series to the primary side of the leakage type transformer, and parallel to the primary side of the leakage type transformer. A first capacitor connected in parallel to the primary side of the leakage type transformer, a second semiconductor switching means connected in series to the second capacitor, A power control circuit for mutually driving the semiconductor switching means and the second semiconductor switching means, a first high-voltage capacitor connected to the secondary side of the leakage type transformer, a second high-voltage capacitor, and a first high-voltage capacitor. A rectifier circuit of a full-wave voltage consisting of a diode, a second high-voltage diode, and a magnetron connected to the rectifier circuit connected to the rectifier circuit It is intended.

The power control circuit changes the conduction time of the first semiconductor switching means and the conduction time of the second semiconductor switching means for controlling power, and switches the first high-voltage capacitor and the second high-voltage capacitor. A high-frequency heating device in which the first high-voltage diode and the second high-voltage diode have a configuration in which substantially the same current flows by selecting a predetermined value.

Hereinafter, embodiments of the present invention will be described with reference to the drawings. (Embodiment 1) FIG. 1 is a circuit diagram of a magnetron driving power supply according to an embodiment of the present invention. In FIG. 1, 3 is a DC power supply,
54 is a first capacitor, 55 is a second capacitor, 3
Reference numeral 8 denotes first semiconductor switching means, which has a parallel configuration of a transistor and a diode. Reference numeral 39 denotes second semiconductor switching means, which also has the same configuration as the first semiconductor switching means 38. 9 is an inverter control circuit having the function of a power control circuit, 7 is a leakage type transformer, 57 is a first high voltage diode 18, a second high voltage diode 19, a first high voltage capacitor 10, and a second high voltage capacitor 11. A rectifier circuit for performing full-wave voltage rectification, and 12 is a magnetron.

This portion is the main power circuit of the power supply for driving the magnetron, and the same circuit configuration is shown in FIG.
(B) and (c) can be considered. (A), the second capacitor 55 is arranged in parallel with the primary winding 13b, and the first capacitor 54 and the first semiconductor switching means 38 are arranged in parallel. (B), the primary winding 13b and the first capacitor 54 are arranged in parallel, and the first semiconductor switching means 38
And a second capacitor 55 arranged in parallel. (C) shows a configuration in which a first capacitor 54 and a second capacitor 55 are arranged in parallel with the first semiconductor switching means 38. In each case, the current of the first semiconductor switching means, the current of the magnetron, the primary current of the leakage type transformer, and the voltage of the first semiconductor switching means all have the same waveform.

The inverter control circuit 9 has a first
An oscillation circuit function for generating drive signals for the semiconductor switching means 38 and the second semiconductor switching means 39 is built in. The operation will be described. The oscillation circuit is a self-excited oscillation circuit in which the timing of the waveform change is determined by using the magnitude relation of the internally generated voltage as a cross trigger, and the frequency changes depending on the internally generated voltage waveform as in the conventional example. is not. Therefore, the frequency of the drive signal is constant.

The lower threshold value (d) is a divided voltage value of the control circuit voltage Vcc of the resistors 47 and 48, the upper threshold value (e) is a divided voltage value of the control circuit voltage Vcc of the resistors 45 and 46, and the triangular wave signal ( A) is the voltage across capacitor 52 and is the current source 4
The charge / discharge of the DC current I generated at 9 and 50 changes in a triangular waveform. This is the triangular waveform (a) in FIG. The upper threshold value (e) is a voltage at which the rising ramp waveform is charged by the DC current I and turned up. Then, the mode shifts to the mode of discharging with the DC current I, and the falling ramp waveform returns again at the lower threshold value (d), returns to the mode of the rising ramp waveform, and repeats this cycle.

This circuit operation mechanism can be understood by returning to FIG. 1 and describing the function of the comparison timing circuit 44 in detail. First, the comparison timing circuit 44 switches the switch 5
A description will be made from the state in which the switch 1 is opened under a command signal for opening. At that time, since the DC 2I of the current source 50 does not flow, the DC I of the current source 49 flows into the capacitor 52 and is charged to increase the voltage. Naturally, it is considered that the impedance of the comparison timing circuit 44 viewed from the capacitor 52 is almost infinite, and no current flows. This is the mode of the rising ramp waveform described above. Therefore, the comparison timing circuit 44 compares the upper threshold value (e) with the voltage at the point (a) using a comparator.
A command signal for closing the switch 51 is output, and a short circuit occurs. Then, the DC current of I (current source 2I
The capacitor 52 is discharged and flows so that 2I-I) obtained by subtracting the current source I from the current source 50 establishes the 2I current of the current source 50. This is the mode of the falling ramp waveform described above. When the voltage reaches the lower threshold value (d), a command signal for opening the switch 51 is generated, the cycle goes through, and a self-excited oscillation triangular wave (a) is created.

Next, the operation of the circuit when the semiconductor switching means is turned on / off and the inverter is driven will be described with reference to FIGS. First,
When the first semiconductor switching means 38 is on, an equivalent circuit of a main circuit part after the DC power supply 56 is shown in FIG.
(A), the first semiconductor switching means 3
8 is supplied from the smoothing capacitor 4 through the primary winding of the leakage type transformer 7 (FIG. 4).
(A) State o). At this time, 2 of leakage type transformer 7
The secondary output starts charging the second high-voltage capacitor 59 of the full-wave voltage doubler rectifier circuit 57. Since the initial voltage V2 is stored in the first high-voltage capacitor 58, the magnetron oscillates when the voltage V3 of the second high-voltage capacitor 59 satisfies the relationship of V2 + V3> Vcut (3) Vcut: cut-off voltage of the magnetron. As a result, a current Ia flows through the magnetron 12 as shown in FIG.

When the first semiconductor switch is turned off, the equivalent circuit becomes as shown in FIG. 3B, and the current flowing on the primary side of the leakage type transformer 7 starts flowing toward the first capacitor 54. At this time, the secondary output of the leakage type transformer 7 starts charging the capacitor 58 (state p). At this time, if equation (3) is satisfied, the magnetron 1
2 starts oscillating, and the anode current naturally flows if (Equation 3) is satisfied. The current on the primary side of the leakage type transformer 7 is as shown in FIG. The voltage of the first semiconductor switching means 38 is as shown in FIG. When this voltage reaches the initial voltage of the second capacitor 55, the diode constituting the second semiconductor switching means 39 is turned on, and the charging of the second capacitor 55 is started. The equivalent circuit at this time is as shown in FIG.
Since the capacitance value of the second capacitor 55 is larger than that of the first capacitor 54, the slope of the voltage of the first semiconductor switching means 38 suddenly becomes gentle, and the state shown in FIG. Move to q. When the current flowing from the primary side of the leakage type transformer 7 toward the second capacitor 5 starts flowing from the second capacitor 5 toward the primary side, the state transits to the state r. At this time, it is necessary to turn on the transistor constituting the second semiconductor switching means 39. Any T1
When the transistor constituting the second semiconductor switching means 39 is shut off during the conduction period, the state shifts to a state s in which current starts to flow from the first capacitor 4 toward the primary side of the leakage type transformer 7. At this time, the slope of the voltage of the first semiconductor switching means 38 becomes steep and decreases toward zero due to the energy of the first capacitor 4. When this voltage becomes zero, the state shifts to a state t in which current flows in the direction in which the diode of the first semiconductor switching means 38 conducts, and the forward voltage of the diode (0. (Approximately 7 V) is applied, and substantially zero voltage application is maintained. When the first semiconductor switching means 38 is driven again in this state, the same operation is repeated from the state o, and the switching operation for reducing the switching loss can be realized. The second mentioned above
Is determined by turning on the second semiconductor switching means 39 for an arbitrary time T1 in the state r. In other words, the longer the ON time of the second semiconductor switching means 39, the lower the initial voltage of the second capacitor 5, and as a result, the voltage of the first semiconductor switching means 38 can be reduced.

As described above, the off-time of the first semiconductor switching means 38, that is, the second semiconductor switching means 39, which cannot be realized by the conventional circuit configuration,
On time can be set arbitrarily,
Furthermore, the voltage of the first semiconductor switching means 38 can be reduced (clamped) by setting the capacitance of the second capacitor 55 to a value sufficiently larger than that of the first capacitor 54.

Next, referring to FIG. 5, it will be shown that, even when the power is controlled, only the duty ratio changes without changing the operating frequency of the inverter. (A) Since the figure is described above, the detailed operation thereof will not be described. (A) shows a state of a predetermined high-frequency output, and (b) shows a state of an even higher output. In simple terms, since both slice the triangular wave (a) having the same oscillation frequency with the reference signal (c), the frequency in the state (a) is 1 / (t1 + t2) and the frequency in the state (b) is 1 /
The duty ratio differs at (t1 * + t2 *), and the frequency is constant. Therefore, in this embodiment, the frequency is constant even if the power is controlled, but only the duty ratio changes.

FIG. 6A shows a secondary-side voltage waveform. The secondary-side voltage of the leakage transformer adjusts the on-time of the first semiconductor switching means 38 and the second semiconductor switching means 39. The forward period t _FW and the flyback period t _FB are set to be substantially equal. (B) is a diagram showing the operating voltage of the high-voltage circuit during the forward period, and (c) is a diagram showing the operating voltage of the high-voltage circuit during the flyback period. Since the flow is as described above, the description is omitted. However, since the forward period t _FW and the flyback period t _FB are almost equal as shown in the diagram (d) showing the anode current waveform, the current flowing through the magnetron is almost equal in both operation modes, High voltage condenser 10
If equal barrel CH1 a serving C _H2 second high-voltage capacitor 11, the anode current is equal become anodic current peak flow cutoff voltage Vcut or more periods of the magnetron can be minimized in both operation modes.

(Embodiment 2) In FIG. 7, (Embodiment 2)
The operation of the high-voltage circuit will be described. (A), (b), (c), and (d) in FIG. 7 are diagrams showing the same contents as (a), (b), (c), and (d) shown in FIG. Here, the on period of the second switching means 39 is controlled so that the forward period t _FW is set longer than the flyback period t _FB . The first high voltage capacitor C _H1 is set smaller than the second high voltage capacitor C _H2 . In this way, in the long forward period t _FW , the anode current sufficiently flows from the large second high-voltage capacitor 11 CH ₂ during the t _FW period. On the other hand, since the period of the flyback t _FB is short, a sufficient anode current can be supplied even if the first high-voltage capacitor 10 CH ₁ is small. Here, the on and off periods of the t _FW and t _FB switching means are controlled to obtain C _H2 = C _H1 × 2.
Realize the relationship. By doing so, the number of parts increases by one, but all can be made of the same type.

(Embodiment 3) In FIG. 8, (a) shows the secondary-side voltage waveform, (b) shows the operating current of the high-voltage circuit during forward operation, (c) shows the operating current of the high-voltage circuit during flyback,
(D) is an anode current waveform. Using this, the operation of the high voltage circuit in (Embodiment 3) will be described.
(C) It is assumed from the drawing that the current flowing through the first high-voltage diode 18 is I _FB . This is a current flowing through the flyback period, the first high-voltage capacitor 10C _H1 being discharged under reduced pressure by the anode current in the previous cycle in this case V
Charge current to lift to _FB and Vc of magnetron
It flows as the sum of the anode currents when it exceeds ut. On the other hand, the current flowing through the second high-voltage diode 19 (b) Fig. And I _FW. This is a current flowing in the forward period, as the sum of the anode current when the time reaches the second serving high-voltage capacitor 11 C _H2 over Vcut charging current and a magnetron for lifting up V _FW whose pressure is reduced in the same manner Flows.

Here, the second semiconductor switching means 3
9 (shorter eventually flyback period) if faster timing of turning off the have already been described with respect to a tendency that the initial voltage increases of the second capacitor 55, Then the secondary voltage V _FB is also elevated flyback period The charging current for raising the first high-voltage capacitor C _H1 to V _FB is also increased, and the total current is increased (naturally, this charging current is larger than the anode current).

As described above, by selecting t _FW , t _FB , C _H1 , and C _H2 , the currents I _FW and I _FB flowing through the high-voltage diode are made uniform, and the same type of high-voltage diode is used with a well-balanced loss. And cost performance can be minimized.

[0040]

As described above, the present invention relates to a first semiconductor switching means connected in series to the primary side of a leakage type transformer, and a first capacitor connected in parallel to the primary side of the leakage type transformer. A second capacitor connected in parallel to the primary side of the leakage type transformer, a second semiconductor switching means connected in series with the second capacitor, a first semiconductor switching means and a second semiconductor switching. A power control circuit for mutually driving the means, a full-wave doubler rectifier circuit having a first high-voltage capacitor and a second high-voltage capacitor connected to the secondary side of the leakage transformer, and a rectifier circuit. And a magnetron connected to the rectifier circuit.

In order to control the power, the power control circuit also changes the conduction time of the first semiconductor switching means and the conduction time of the second semiconductor switching means to change the ON time and the OFF time of the first semiconductor switching means. Is optimized to a predetermined ratio, and the first high-voltage capacitor and the second high-voltage capacitor are of the same type having the same capacity, thereby eliminating the operation of creating added value of sorting parts in manufacturing management. be able to.

In the case of high voltage capacitors of the same category, the lead pitch for component insertion is very often the same even if the capacitance value differs. Therefore, if the capacitance values are different, it is impossible to eliminate all parts as long as human work is involved, resulting in a fatal defect in performance at that time. There is a possibility, but there is no fear if the same capacitance value is used according to the present invention. No operational problems occur.

In addition, the anode current peak is designed to be substantially equal and can be minimized, so that the life of the magnetron can be reduced.
Improving reliability can also be realized as an incidental effect.

Next, according to the present invention, since the power is controlled by the power control circuit, the conduction time of the first semiconductor switching means and the conduction time of the second semiconductor switching means are also changed to turn on the first semiconductor switching means. By optimizing the period and the off period to a predetermined ratio, the anode current peak value of the magnetron becomes substantially the same, and the first high-voltage capacitor and the second high-voltage capacitor have a predetermined capacity and use a plurality of the same type or independently. A configuration that does not create added value, such as sorting of parts for manufacturing management, can be eliminated.

Next, the present invention changes the conduction time of the first semiconductor switching means and the conduction time of the second semiconductor switching means in order to control the power by the power control circuit. By selecting the high-voltage capacitor to a predetermined value, the first high-voltage diode and the second high-voltage diode have a configuration in which substantially the same current flows.

Therefore, the currents flowing through the first high-voltage diode and the second high-voltage diode are made uniform, and the high-voltage diodes of the same type can be used in a well-balanced manner and used. Can be minimized.

[Brief description of the drawings]

FIG. 1 is a circuit diagram of a power supply for driving a magnetron of a high-frequency heating device according to a first embodiment of the present invention.

2A is another configuration diagram of the magnetron driving power supply circuit. FIG. 2B is another configuration diagram of the magnetron driving power supply circuit. FIG. 2C is another configuration diagram of the magnetron driving power supply circuit.

FIG. 3 (a) shows a first example of the magnetron driving power supply.
FIG. 4B is a diagram showing an equivalent circuit of a main circuit portion when the semiconductor switching means is turned on. (B) When only the first capacitor is charged and discharged when the first semiconductor switching means is turned off in the magnetron driving power supply. (C) In the magnetron driving power supply, the main circuit portion when the first capacitor and the second capacitor are charging and discharging when the first semiconductor switching means is off. Diagram showing the equivalent circuit of

FIG. 4 is a characteristic diagram of each part of the magnetron driving power supply.

FIG. 5A is a time chart of a voltage in a high-frequency output state of the power control circuit. FIG. 5B is a time chart of a voltage of the power control circuit in a high-frequency output state higher than FIG. 5A.

FIG. 6 is a diagram illustrating the principle of a power supply for driving the magnetron.

FIG. 7 is a diagram illustrating the principle of a power supply for driving a magnetron according to a second embodiment of the present invention;

FIG. 8 is a diagram illustrating the principle of a magnetron driving power supply according to a third embodiment of the present invention.

FIG. 9 is a circuit diagram of a power supply device for driving a magnetron of a conventional high-frequency heating device.

FIG. 10 is a time chart of the voltage of each part of the power control circuit.

FIG. 11 is a main part circuit diagram of the power control circuit.

FIG. 12 is a characteristic diagram of the magnetron driving power supply.

FIG. 13 is a diagram showing a relationship between current and voltage of the semiconductor switching element.

FIG. 14 is a principle explanatory view showing a defect of a magnetron driving power supply in a conventional high-frequency heating device.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 Commercial power supply 7 Leakage type transformer 9 Power control circuit 10 1st high voltage capacitor 11 2nd high voltage capacitor 12 magnetron 18 1st high voltage diode 19 1st high voltage diode 38 1st semiconductor switching means 39 2nd semiconductor switching Means 54 First capacitor 55 Second capacitor 56 DC power supply 57 Full-wave voltage doubler rectifier

 ──────────────────────────────────────────────────の Continuing on the front page (72) Inventor Kenji Yasui 1006 Kazuma Kadoma, Osaka Pref. Matsushita Electric Industrial Co., Ltd.

Claims (3)

[Claims] 1. A DC power supply, a leakage transformer connected to the DC power supply, and first semiconductor switching means connected in series to a primary side of the leakage transformer.
A first capacitor whose voltage resonates with energy stored in the leakage type transformer when the first semiconductor switching means is off, a second capacitor that clamps a voltage of the first capacitor, and a second capacitor. A second semiconductor switching means connected in series to the capacitor, a power control circuit for alternately driving the first semiconductor switching means and the second semiconductor switching means, and a secondary side of the leakage type transformer. A rectifier circuit connected thereto; and a magnetron connected to the rectifier circuit, wherein the power control circuit controls the first power to control power.
And the conduction time of the second semiconductor switching means is also changed so that the first high-voltage capacitor and the second high-voltage capacitor are of the same type having the same capacitance and the first semiconductor switching means. A high-frequency heating apparatus having a configuration in which the peak value of the anode current of the magnetron is substantially the same during the ON period and the OFF period of the means. 2. A DC power supply, a leakage transformer connected to the DC power supply, and first semiconductor switching means connected in series to a primary side of the leakage transformer.
A first capacitor whose voltage resonates with energy stored in the leakage type transformer when the first semiconductor switching means is off, a second capacitor that clamps a voltage of the first capacitor, and a second capacitor. A second semiconductor switching means connected in series to the capacitor, a power control circuit for alternately driving the first semiconductor switching means and the second semiconductor switching means, and a secondary side of the leakage type transformer. A rectifier circuit connected thereto; and a magnetron connected to the rectifier circuit, wherein the power control circuit controls the first power to control power.
The conduction time of the semiconductor switching means is changed and the conduction time of the second semiconductor switching means is also changed, and a plurality of first high voltage capacitors and second high voltage capacitors of the same capacity and of the same type are used. A high-frequency heating apparatus used alone or configured so that the peak value of the anode current of the magnetron becomes substantially the same during the ON period and the OFF period of the first semiconductor switching means. 3. A DC power supply, a leakage transformer connected to the DC power supply, and first semiconductor switching means connected in series to a primary side of the leakage transformer.
A first capacitor whose voltage resonates with energy stored in the leakage type transformer when the first semiconductor switching means is off, a second capacitor that clamps a voltage of the first capacitor, and a second capacitor. A second semiconductor switching means connected in series to the capacitor, a power control circuit for alternately driving the first semiconductor switching means and the second semiconductor switching means, and a secondary side of the leakage type transformer. A rectifier circuit connected thereto; and a magnetron connected to the rectifier circuit, wherein the power control circuit controls the first power to control power.
By changing the conduction time of the semiconductor switching means and the conduction time of the second semiconductor switching means to select the first high-voltage capacitor and the second high-voltage capacitor to predetermined values, the first high-voltage diode and The second
The high-frequency heating device has a configuration in which the same high-current diodes flow.