JP3216562B2 - High frequency heating equipment - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power supply for driving a magnetron of a high-frequency heating device using a magnetron.

[0002]

2. Description of the Related Art As shown in FIG. 9, a conventional high-frequency heating apparatus uses a circuit configuration called a one-piece voltage resonance type circuit. 1 is a commercial power supply, 2 is a rectifier composed of full-wave rectification, which rectifies the commercial power supply to form a DC power supply. The choke coil 3 smoothes the fluctuation of the current, and the smoothing capacitor 4 smoothes the fluctuation of the voltage. These inductors and capacitors form a low-pass filter circuit. Energy is accumulated in the winding of the leakage type transformer 14 from the DC power supply at the point (a). The primary winding 13b of the leakage transformer 14 and the resonance capacitor 6 arranged in parallel with it constitute a tank circuit, and resonate with the energy stored in the winding of the leakage transformer 14. Switching means comprising a transistor 5 and a diode 8 arranged in parallel with the transistor 5 are connected in series to the tank circuit.

The secondary side of the leakage type transformer 14 is
A secondary winding 13a and a heater winding 13c are provided. The voltage induced in the secondary winding 13c is the first high voltage capacitor 1
0, a second high-voltage capacitor 11, a first high-voltage diode 18, and a second high-voltage diode 19, which are rectified by a full-wave voltage doubler rectifier circuit and apply a DC high voltage of about -4 kV to the cathode of the magnetron 12. It has become.

A current transformer 15 is a magnetron 12
Is detected, and the information is insulated and returned to the inverter control circuit 9 having the power control circuit. The inverter control circuit 9 controls the on time of the magnetron 5 so that the signal from the current transformer 15 becomes constant, and controls the power transmitted to the secondary side. In this control law, since the anode current is controlled at a constant value,
The responsibility of the anode current, which will cause serious damage to the magnetron if it increases excessively, can be controlled to an appropriate level,
This is a very advantageous control law in terms of magnetron quality and reliability.

Next, a mechanism for driving the magnetron 5 and supplying power to the secondary side will be described with reference to FIG. The upper time chart shows a signal relationship for generating a pulse signal (e) for driving the magnetron 5 with a certain set high-frequency output, and the lower time chart shows the pulse train.

The upper triangular waveform is sliced by the high frequency output reference signal (f). When the triangular wave (g) signal is lower than (f), a high signal is output, and conversely, a low signal is output. The respective times are t1 and t2, and when t1 is long, the energy stored in the primary winding 13b increases. The operating frequency f is 1 / (t1 + t2).
The vertex of the triangular wave (g) crosses the points (a) and (b) in FIG. 12 and generates a cross trigger under the conditions of (a)> (b) to (b) <(b), and the slope is Turns down. Then, when it crosses the inversion threshold value (c), the mode is naturally changed, and the slope changes to a rising slope. By repeating this, a triangular wave (g) is formed.

Further, the specific circuit configuration and operation will be described with reference to FIG. Point (a) is a complete direct current when the circuit is not operating, but once operated and power is consumed on the secondary side, it has a waveform obtained by full-wave rectification of the sine shape of the power supply. When the effective voltage E is E = 100 V, the peak value is about 141 V when E = 100 V. On the other hand, point (b) is a resonance voltage waveform generated in the above-described tank circuit, which is the voltage waveform of the semiconductor switching means in FIG. 13 (d), and its peak voltage is about 500 V, which is much higher than point (a). Big. Returning to FIG. 11, reference numeral 26 denotes a comparison timing circuit, and a voltage obtained by dividing the point (a) by the resistors 22 and 23;
(B) Compare the voltage with the voltage obtained by dividing the point by resistors 24 and 25,
The cross trigger described above is detected. The inversion threshold (C) is determined by dividing the circuit power supply voltage Vcc by the resistors 20 and 21. Now, when the transistor 31 is on, the electric charge of the capacitor 17 is discharged via the resistors 16 and 30, and the triangular wave (g) enters a falling slope state. Next, when the triangular wave (g) falls and crosses the inversion threshold (c), the output (d) of the comparison timing circuit 26 turns off the transistor 31 via the resistor 29, and the capacitor 17 becomes the resistor 16
And the triangular wave (g) turns to a rising slope. And it reverses again by the above-mentioned cross trigger and turns into a descending slope.

The triangular wave (g) is compared with a high-frequency output reference signal (f) of 28 by a comparator 27 to generate a pulse signal (e). The circuit configuration shown here exists inside the inverter control circuit 9 and the pulse signal drives the magnetron 5 through an appropriate driver circuit.

As described above, the reason for driving the magnetron 5 using the cross trigger and the comparison timing circuit 26 is as shown in FIG. 13 (b) when the transistor current is cut off (a) In addition to reducing the switching loss that occurs as a product of the current and the voltage by increasing the slope of the rise relatively slowly in a sinusoidal manner, the period during which the diode current flows with a predetermined delay from the cross trigger, that is, (a) the voltage By turning on the magnetron 5 when the waveform is 0 V, the switching loss is similarly reduced. This is a major feature of the voltage resonance system.

FIG. 12 is a time chart of waveforms of main parts. (A) is a current waveform of the semiconductor switching means,
(B) shows the current waveform of the magnetron, (c) shows the current waveform of the leakage type transformer, and (d) shows the voltage waveform of the semiconductor switching means. A current flows through the transistor during the period (I), and a resonance voltage of the tank circuit is generated during the period (II). The diode current flows during the period (III), and the transistor is turned on for this period. With such control, the switching loss of the transistor is minimized.

[0011]

However, the conventional magnetron drive power supply has the following phenomena when a single-pole voltage resonance type circuit is used.

(1) The resonance period is a constant period determined by the intrinsic constants L and C. On the other hand, the on-period of the semiconductor switching means, which is a non-resonance period, is arbitrarily determined by the power, and its on / off duty. The ratio differs depending on the design specifications (particularly the resonance circuit constant specifications) and the set output, and in the voltage resonance type, it is consequently determined that ON: OFF = about 2: 1 under conditions satisfying the performance.

(2) As a voltage transmitted in the transformer, a period in which the magnetron 5 is on is called forward, and a period in which the tank circuit resonates when it is turned off is called flyback. In the forward period (tFW), voltage is applied. Is a power supply voltage, and has a waveform having a high peak value due to sharp resonance during the flyback period (tFB), as shown in FIG. 14A, and the swing to the positive and negative sides is very imbalanced.

FIG. 14A specifically shows this.
The waveform of the secondary side voltage of the transformer is tFW> tFB, VFW
<The relationship of VFB is clearly visible.

For example, the behavior of the rectifier circuit on the secondary side of the transformer at this time is as shown in FIG. The current IFW is charged to the second high-voltage capacitor 11 through the first high-voltage diode 19 by the voltage VFW induced in the secondary winding 13a. Assuming that there is a voltage V1 generated by the electric charge already stored in the first high-voltage capacitor 10 at that time, the magnetron oscillates in a relationship of V1 + VFW> Vcut (1) Vcut; cutoff voltage of the magnetron. The anode current at that time flows in such a manner that accumulated charges in the first high-voltage capacitor 10 and the second high-voltage capacitor 11 are discharged. Therefore, the second high-voltage capacitor 11 is supplied with electric charge from the primary side and the voltage is maintained, but the first high-voltage capacitor 10 only discharges and oscillates until Vcut or less. However, since the forward period is longer than the flyback period, it is necessary to use a large-capacity first high-voltage capacitor 10. Otherwise, the discharge is performed in a short period of time and becomes equal to or lower than Vcut, so that a small amount of anode current flows in a short period of time.

On the other hand, during the flyback period, FIG.
(C). Voltage VFB induced in secondary winding 13b
As a result, the current of IFB is charged in the first high-voltage capacitor 10 through the second high-voltage diode 18. Assuming that there is a voltage V2 generated by the electric charge already stored in the second high-voltage capacitor 11 at that time, the magnetron oscillates in a relationship of V2 + VFB> Vcut (2). The anode current at that time flows in such a manner that accumulated charges in the first high-voltage capacitor 10 and the second high-voltage capacitor 11 are discharged. Therefore, the first high-voltage capacitor 10 is supplied with electric charge from the primary side and the voltage is maintained, but the second high-voltage capacitor 11 is only discharged, and the oscillation continues until it becomes equal to or lower than Vcut. However, since the flyback period is shorter than the forward period, the first high-voltage capacitor CH1,
Does not need to be as large as CH2 as the second high-voltage capacitor 11, and is configured to allow sufficient anode current to flow by using a capacitor having a small capacity.

Therefore, within the range of the relationship of CH1 <CH2,
In order for the average value of the anode current (a parameter for determining the high-frequency output) to be a predetermined value and the peak value of the anode current to be equal to or less than the absolute maximum rating, the first high-voltage capacitor 10 and the second high-voltage capacitor 11 Needs to be optimized unbalanced, and two capacitors of different capacitances are required, so that the current peaks in the forward period and the flyback period can be equalized. FIG. 14D shows the waveform of the actual anode current.
If both parts have the same lead pitch, it is necessary to assemble them while taking into account erroneous insertion. This has been a very serious problem in terms of management, manufacturing workability, and quality in view of erroneous insertion.

[0018]

SUMMARY OF THE INVENTION In order to solve the above-mentioned problems, the present invention provides a DC power supply and a lead connected to the DC power supply.
A cage transformer, a first semiconductor switching means connected in series to said DC power supply to the primary side of the leakage transformer, said first semiconductor switching hands
When the stage is off, the energy accumulated in the leakage transformer
A first capacitor which resonates with energy, the first co
Series connection of a second capacitor provided to clamp the voltage of the capacitor and a second semiconductor switching means.
And the first semiconductor switching means and the second half.
Power control circuit for alternately driving conductor switching means
And a first high-voltage capacitor and a second high-voltage capacitor connected to the secondary side of the leakage type transformer.
A rectifier circuit for full-wave voltage doubler that is connected to the rectifier circuit
A second magnetron, and the second capacitor and
The series connection of the second semiconductor switching means has one end thereof.
With the leakage type transformer and the first semiconductor switch.
The other end is located at the middle point of the
Is connected to the negative side, and the power control circuit
The first semiconductor switching means and
And the conduction time of the second semiconductor switching means is changed.
The first high-voltage condenser and the second high-voltage condenser
And the first semiconductor device as the same product with the same capacity.
In the on period and the off period of the switching means,
The anode current peak values of the electrodes are substantially the same . The power control circuit for driving the first semiconductor switching means and the second semiconductor switching means is connected to the secondary side of the leakage type transformer and controls the power transmission to the full wave voltage doubler circuit and the magnetron. Not only can the conduction time of the means be controlled, but also the conduction time of the second semiconductor switching means can be controlled. Accordingly, the forward period determined by the conduction time of the first semiconductor switching means and the other flyback periods are set at a predetermined ratio, and the peaks of the anode current during the forward and flyback periods are almost uniformly set. It is possible to

According to the above-mentioned invention, the values of the first high-voltage capacitor and the second high-voltage capacitor are set to the same capacity and the same type to reduce the number of component types and to improve the simplicity of the production management associated therewith. It is possible to prevent erroneous insertion (misplacement), which may occur when constants of different capacities are used, and it is possible to expect improvement in manufacturing quality.

[0020]

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a DC power supply, a leakage type transformer connected to the DC power supply, and first semiconductor switching means connected in series to the DC power supply on the primary side of the leakage type transformer. , The first semiconductor switch
When the switching means is off, the leakage type transformer
A first capacitor which resonates with the product energy, before
A second capacitor and a second semiconductor switching means provided to clamp the voltage of the first capacitor
Of the series connection, a power control circuit and the first semiconductor switching means and a second semiconductor switching means for Alternating drive is connected to the secondary side of the leakage transformer, first
And a second high-voltage capacitor.
And a magnetron connected to the rectifier circuit , wherein the second capacitor and the second
One end of the series connection of the semiconductor switching means is
Leakage type transformer and the first semiconductor switching
At the middle point of the means, connect the other end to the + or-side of the DC power supply.
And the power control circuit controls the power.
The first semiconductor switching means and the
Changing the conduction time of the second semiconductor switching means,
The first high-voltage capacitor and the second high-voltage capacitor
The first semiconductor switch of the same type with the same capacity
The magnetron is turned on and off during the on and off periods of the
In this configuration, the node current peak values are substantially the same .

In order to control the power, the power control circuit changes the conduction time of the first semiconductor switching means and also changes the conduction time of the second semiconductor switching means, thereby controlling the ON period of the first semiconductor switching means. And the off-period is optimized to a predetermined ratio, so that the first high-voltage capacitor and the second high-voltage capacitor are of the same type having the same capacity, and the peak current of the anode current of the magnetron is substantially the same. Is realized.

Further, a DC power source, and re <br/> Keji transformer connected to a DC power source, a first semiconductor switching means connected in series to said DC power supply to the primary side of the leakage transformer, The first semiconductor switching means is
Energy stored in the leakage transformer
A first capacitor which resonates at over said first Conde down
A second capacitor provided to clamp the voltage of the
And a series connection of the second semiconductor switching means, and a power control circuit and the first semiconductor switching means and a second semiconductor switching means for Alternating drive is connected to the secondary side of the leakage transformer, first High voltage condenser
And a magnetron connected to the rectifier circuit, the second capacitor and a second semiconductor switch.
One end of the leakage means is connected in series with the leakage type
An intermediate point between the lance and the first semiconductor switching means
And the other end is connected to the + side or the − side of the DC power supply.
Together with the power control circuit for controlling power
First semiconductor switching means and said second semiconductor
By changing the conduction time of the switching means, the first high voltage
Capacitor and second high-voltage capacitor with the same capacity and the same type
The first is to use a plurality of
The on-off period and the off-period of the semiconductor switching means are set to the above-mentioned timing.
A structure in which the peak current of the anode current of
It was a good thing.

When controlling the power, the power control circuit changes the conduction time of the first semiconductor switching means and optimizes the ON time and the OFF time of the conduction time of the second semiconductor switching means to a predetermined ratio. The peak current of the anode current of the magnetron is almost the same, and
An object of the present invention is to realize a high-frequency heating device in which the first high-voltage condenser and the second high-voltage condenser are of a predetermined capacity and use a plurality of ones of the same type or are used alone.

A DC power supply, a leakage transformer connected to the DC power supply, and one of the leakage transformers.
A first semiconductor switching means connected in series with the DC power source to the next side, said first semiconductor switching
Means accumulated in the leakage type transformer when off
A first capacitor that resonates with energy ;
A series connection of a second capacitor provided to clamp the voltage of the capacitor and the second semiconductor switching means .
Connections and a power control circuit and the first semiconductor switching means and a second semiconductor switching means for Alternating drive, the first high-pressure co <br/> Nden support connected to the secondary side of the leakage transformer When the second high capacitor and the first high-pressure diode
It includes a rectifier circuit of the full-wave voltage doubler consisting of de second high voltage diode, and a magnetron connected to the rectifier circuit, before
A second capacitor and a second semiconductor switching device;
One end of the series connection of the
And an intermediate point between the first semiconductor switching means and the other end.
Is connected to the + or-side of the DC power supply.
The power control circuit controls the first power to control power.
Semiconductor switching means and the second semiconductor switch
Changing the conduction time of the switching means,
The capacitor and the second high-voltage capacitor are selected to have a predetermined capacity.
By doing so, the first high-voltage diode and the second high-voltage diode
In this configuration, almost the same current flows through the diode .

The power control circuit changes the conduction time of the first semiconductor switching means and the conduction time of the second semiconductor switching means for controlling power, and switches the first high-voltage capacitor and the second high-voltage capacitor. By selecting a predetermined capacity , it is possible to realize a high-frequency heating device in which the first high-voltage diode and the second high-voltage diode have a configuration in which substantially the same current flows.

Hereinafter, embodiments of the present invention will be described with reference to the drawings. Embodiment 1 FIG. 1 is a circuit diagram of a magnetron driving power supply according to an embodiment of the present invention. In FIG. 1, 3 is a DC power supply, 54 is a first capacitor, 55 is a second capacitor, and 38 is first semiconductor switching means, which is a parallel configuration of a transistor and a diode. Reference numeral 39 denotes second semiconductor switching means, which also has the same configuration as the first semiconductor switching means 38. 9 is an inverter control circuit having the function of a power control circuit, 7 is a leakage type transformer, 57 is the first high voltage diode 18, and the second high voltage diode 1
9, a rectifier circuit for performing full-wave voltage rectification comprising a first high-voltage capacitor 10 and a second high-voltage capacitor 11; and 12, a magnetron.

This portion is the main power circuit of the power supply for driving the magnetron, and the same circuit configuration is shown in FIG.
(B) and (c) can be considered. (A), the second capacitor 55 is arranged in parallel with the primary winding 13b, and the first capacitor 54 and the first semiconductor switching means 38 are arranged in parallel. (B), the primary winding 13b and the first capacitor 54 are arranged in parallel, and the first semiconductor switching means 38
And a second capacitor 55 arranged in parallel. (C) shows a configuration in which a first capacitor 54 and a second capacitor 55 are arranged in parallel with the first semiconductor switching means 38. In each case, the current of the first semiconductor switching means, the current of the magnetron, the primary current of the leakage type transformer, and the voltage of the first semiconductor switching means all have the same waveform.

The inverter control circuit 9 has a first
An oscillation circuit function for generating drive signals for the semiconductor switching means 38 and the second semiconductor switching means 39 is built in. The operation will be described. The oscillation circuit is a self-excited oscillation circuit in which the timing of the waveform change is determined by using the magnitude relation of the internally generated voltage as a cross trigger, and the frequency changes depending on the internally generated voltage waveform as in the conventional example. is not. Therefore, the frequency of the drive signal is constant.

The lower threshold value (d) is a divided voltage value of the control circuit voltage Vcc of the resistors 47 and 48, the upper threshold value (e) is a divided voltage value of the control circuit voltage Vcc of the resistors 45 and 46, and the triangular wave signal ( A) is the voltage across capacitor 52 and is the current source 4
The charge / discharge of the DC current I generated at 9 and 50 changes in a triangular waveform. This is the triangular waveform (a) in FIG. The upper threshold value (e) is a voltage at which the rising ramp waveform is charged by the DC current I and turned up. Then, the mode shifts to the mode of discharging with the DC current I, and the falling ramp waveform returns again at the lower threshold value (d), returns to the mode of the rising ramp waveform, and repeats this cycle.

This circuit operation mechanism can be understood by returning to FIG. 1 and describing the function of the comparison timing circuit 44 in detail. First, the comparison timing circuit 44 switches the switch 5
A description will be made from the state in which the switch 1 is opened under a command signal for opening. At that time, since the DC 2I of the current source 50 does not flow, the DC I of the current source 49 flows into the capacitor 52 and is charged to increase the voltage. Naturally, it is considered that the impedance of the comparison timing circuit 44 viewed from the capacitor 52 is almost infinite, and no current flows. This is the mode of the rising ramp waveform described above. Therefore, the comparison timing circuit 44 compares the upper threshold value (e) with the voltage at the point (a) using a comparator.
A command signal for closing the switch 51 is output, and a short circuit occurs. Then, the DC current of I (current source 2I
The capacitor 52 is discharged and flows so that 2I-I) obtained by subtracting the current source I from the current source 50 establishes the 2I current of the current source 50. This is the mode of the falling ramp waveform described above. When the voltage reaches the lower threshold value (d), a command signal for opening the switch 51 is generated, the cycle goes through, and a self-excited oscillation triangular wave (a) is created.

Next, the operation of the circuit when the semiconductor switching means is turned on / off and the inverter is driven will be described with reference to FIGS. First,
When the first semiconductor switching means 38 is on, an equivalent circuit of a main circuit part after the DC power supply 56 is shown in FIG.
(A), the first semiconductor switching means 3
8 is supplied from the smoothing capacitor 4 through the primary winding of the leakage type transformer 7 (FIG. 4).
(A) State o). At this time, 2 of leakage type transformer 7
The secondary output starts charging the second high-voltage capacitor 59 of the full-wave voltage doubler rectifier circuit 57. Since the initial voltage V2 is stored in the first high-voltage capacitor 58, the magnetron oscillates when the voltage V3 of the second high-voltage capacitor 59 satisfies the relationship of V2 + V3> Vcut (3) Vcut: cut-off voltage of the magnetron. As a result, a current Ia flows through the magnetron 12 as shown in FIG.

When the first semiconductor switch is turned off, the equivalent circuit becomes as shown in FIG. 3B, and the current flowing on the primary side of the leakage transformer 7 starts flowing toward the first capacitor 54. At this time, the secondary output of the leakage type transformer 7 starts charging the capacitor 58 (state p). At this time, if equation (3) is satisfied, the magnetron 1
2 starts oscillating, and the anode current naturally flows if (Equation 3) is satisfied. The current on the primary side of the leakage type transformer 7 is as shown in FIG. The voltage of the first semiconductor switching means 38 is as shown in FIG. When this voltage reaches the initial voltage of the second capacitor 55, the diode constituting the second semiconductor switching means 39 is turned on, and the charging of the second capacitor 55 is started. The equivalent circuit at this time is as shown in FIG.
Since the capacitance value of the second capacitor 55 is larger than that of the first capacitor 54, the slope of the voltage of the first semiconductor switching means 38 becomes sharply gentle, and the state shown in FIG. Move to q. When the current flowing from the primary side of the leakage type transformer 7 toward the second capacitor 5 starts flowing from the second capacitor 5 toward the primary side, the state transits to the state r. At this time, it is necessary to turn on the transistor constituting the second semiconductor switching means 39. Any T1
When the transistor constituting the second semiconductor switching means 39 is shut off during the conduction period, the state shifts to a state s in which current starts to flow from the first capacitor 4 toward the primary side of the leakage type transformer 7. At this time, the slope of the voltage of the first semiconductor switching means 38 becomes steep, and decreases toward zero due to the energy of the first capacitor 4. When this voltage becomes zero, the state shifts to a state t in which a current flows in the direction in which the diode of the first semiconductor switching means 38 conducts, and the forward voltage (0. (Approximately 7 V) is applied, and substantially zero voltage application is maintained. When the first semiconductor switching means 38 is driven again in this state, the same operation is repeated from the state o, and the switching operation for reducing the switching loss can be realized. The second mentioned above
Is determined by turning on the second semiconductor switching means 39 for an arbitrary time T1 in the state r. That is, the longer the on-time of the second semiconductor switching means 39, the lower the initial voltage of the second capacitor 5, and as a result, the voltage of the first semiconductor switching means 38 can be reduced.

As described above, the off-time of the first semiconductor switching means 38, that is, the second semiconductor switching means 39, which cannot be realized by the conventional circuit configuration,
On time can be set arbitrarily,
Furthermore, the voltage of the first semiconductor switching means 38 can be reduced (clamped) by setting the capacitance of the second capacitor 55 to a value sufficiently larger than that of the first capacitor 54.

Next, referring to FIG. 5, it will be shown that, even when the power is controlled, only the duty ratio changes without changing the operating frequency of the inverter. (A) Since the figure is described above, the detailed operation thereof will not be described. (A) shows a state of a predetermined high-frequency output, and (b) shows a state of an even higher output. In simple terms, since both slice the triangular wave (a) having the same oscillation frequency with the reference signal (c), the frequency in the state (a) is 1 / (t1 + t2) and the frequency in the state (b) is 1 /
The duty ratio differs at (t1 * + t2 *), and the frequency is constant. Therefore, in this embodiment, the frequency is constant even if the power is controlled, but only the duty ratio changes.

FIG. 6A shows a secondary-side voltage waveform. The secondary-side voltage of the leakage transformer adjusts the on-time of the first semiconductor switching means 38 and the second semiconductor switching means 39. The forward period tFW and the flyback period tFB are set to be substantially equal. (B) is a diagram showing the operating voltage of the high-voltage circuit during the forward period, and (c) is a diagram showing the operating voltage of the high-voltage circuit during the flyback period. Since the flow is as described above, the description is omitted. However, since the forward period tFW and the flyback period tFB are substantially equal as shown in FIG. 4D showing the anode current waveform, the current flowing through the magnetron in both operation modes is substantially equal, and the first high-voltage capacitor 10
If the CH1 and the CH2 are equal, the anode current flowing during the period equal to or higher than the cutoff voltage Vcut of the magnetron is equal, and the anode current peak can be minimized in both operation modes.

(Embodiment 2) Referring to FIG. 7, the operation of the high voltage circuit in (Embodiment 2) will be described. (A), (b),
(C), (d) show (a), (b), (c),
It is a figure which shows the content synonymous with (d). Here, the ON period of the second switching means 39 is controlled so that the forward period tFW is set longer than the flyback period tFB. The first high voltage capacitor CH1 is set to be smaller than the second high voltage capacitor CH2. In this way, in the long forward period tFW, the large second high-voltage capacitor 11 CH2
, The anode current sufficiently flows during the period tFW. On the other hand, since the period of flyback tFB is short, the first high-voltage capacitor 1
Even if CH1 is small, a sufficient anode current can be supplied. Here, the relationship of CH2 = CH1 × 2 is realized by controlling the ON and OFF periods of the tFW and tFB switching means. By doing so, the number of parts increases by one, but all can be made of the same type.

(Embodiment 3) In FIG. 8, (a) shows the secondary-side voltage waveform, (b) shows the operating current of the high-voltage circuit during forward operation, (c) shows the operating current of the high-voltage circuit during flyback, d) is the anode current waveform. Using this, the operation of the high voltage circuit in (Embodiment 3) will be described. (C) It is assumed from the drawing that the current flowing through the first high-voltage diode 18 is IFB. This is the current that flows during the flyback period. At this time, when the charge current for raising the first high-voltage capacitor 10CH1 discharged by the anode current and reduced in pressure in the previous cycle to VFB and the voltage becomes higher than the magnetron Vcut. Flows as the sum of the anode currents. On the other hand, the current flowing through the second high-voltage diode 19 from FIG. This is a current flowing in the forward period, and flows at this time as the sum of a charging current for raising the second high voltage capacitor CH2, which has been similarly reduced in pressure, to VFW and an anode current when the voltage exceeds Vcut of the magnetron.

Here, the second semiconductor switching means 3
If the timing for turning off the capacitor 9 is made faster (the flyback period becomes shorter in the end), the initial voltage of the second capacitor 55 tends to increase as described above. The charging current for raising CH1 as one high-voltage capacitor 10 to VFB also increases, and the total current increases (this charging current is naturally assumed to be larger than the anode current).

As described above, by selecting t FW, t FB, CH 1, and CH 2, the currents I FW and I FB flowing through the high-voltage diode are made uniform, so that the same type of high-voltage diode can be used with a well-balanced loss and cost performance. Minimization can be achieved.

[0040]

As described above, according to the first aspect of the present invention,
For example , by making the first high-voltage capacitor and the second high-voltage capacitor of the same type having the same capacity, it is possible to abolish the operation of sorting the parts for manufacturing management, which does not create added value.

In the case of high-voltage capacitors of the same category, the lead pitch for component insertion is very often the same even if the capacitance value differs. Therefore, if the capacitance values are different, it is impossible to eliminate all parts as long as human work is involved, resulting in a fatal defect in performance at that time. There is a possibility, but there is no fear if the same capacitance value is used according to the present invention. No operational problems occur.

The anode current peaks are designed to be substantially equal to each other, and can be minimized.
Improving reliability can also be realized as an incidental effect.

According to the second aspect of the present invention, the peak current of the anode current of the magnetron becomes substantially the same, and
The first high-voltage capacitor and the second high-voltage capacitor may be configured to use a plurality of capacitors of the same type with a predetermined capacity or may be used independently, so that an operation that does not create added value such as sorting of parts in manufacturing management can be eliminated.

According to the third aspect of the present invention, the currents flowing through the first high-voltage diode and the second high-voltage diode are made uniform, and the high-voltage diodes of the same type can be used in a well-balanced manner. It is possible to distribute the responsibilities equally, that is, to minimize cost performance.

[Brief description of the drawings]

FIG. 1 is a circuit diagram of a power supply for driving a magnetron of a high-frequency heating device according to a first embodiment of the present invention.

2A is another configuration diagram of the magnetron driving power supply circuit. FIG. 2B is another configuration diagram of the magnetron driving power supply circuit. FIG. 2C is another configuration diagram of the magnetron driving power supply circuit.

FIG. 3 (a) shows a first example of the magnetron driving power supply.
FIG. 4B is a diagram showing an equivalent circuit of a main circuit portion when the semiconductor switching means is turned on. (B) When only the first capacitor is charged and discharged when the first semiconductor switching means is turned off in the magnetron driving power supply. (C) In the magnetron driving power supply, the main circuit portion when the first capacitor and the second capacitor are charging and discharging when the first semiconductor switching means is off. Diagram showing the equivalent circuit of

FIG. 4 is a characteristic diagram of each part of the magnetron driving power supply.

FIG. 5A is a time chart of a voltage in a high-frequency output state of the power control circuit. FIG. 5B is a time chart of a voltage of the power control circuit in a high-frequency output state higher than FIG. 5A.

FIG. 6 is a diagram illustrating the principle of the power supply for driving the magnetron.

FIG. 7 is a diagram illustrating the principle of a power supply for driving a magnetron according to a second embodiment of the present invention;

FIG. 8 is a diagram illustrating the principle of a magnetron driving power supply according to a third embodiment of the present invention.

FIG. 9 is a circuit diagram of a power supply device for driving a magnetron of a conventional high-frequency heating device.

FIG. 10 is a time chart of the voltage of each part of the power control circuit.

FIG. 11 is a main part circuit diagram of the power control circuit.

FIG. 12 is a characteristic diagram of the power supply for driving the magnetron.

FIG. 13 is a diagram showing a relationship between current and voltage of the semiconductor switching element.

FIG. 14 is a principle explanatory view showing a defect of a magnetron driving power supply in a conventional high-frequency heating device.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 Commercial power supply 7 Leakage type transformer 9 Power control circuit 10 First high-voltage capacitor 11 Second high-voltage capacitor 12 Magnetron 18 First high-voltage diode 19 First high-voltage diode 38 First semiconductor switching means 39 Second semiconductor switching Means 54 First capacitor 55 Second capacitor 56 DC power supply 57 Full-wave voltage rectifier circuit

Continuing on the front page (72) Inventor Kenji Yasui 1006 Kazuma Kadoma, Osaka Prefecture Matsushita Electric Industrial Co., Ltd. (72) Inventor Yoshiro Ishio 1006 Kadoma Kadoma, Kadoma City Osaka Pref. References JP-A-5-199768 (JP, A) JP-A-8-17566 (JP, A) (58) Fields investigated (Int. Cl. ⁷ , DB name) H05B 6/66-6/68

Claims (3)

(57) [Claims] A DC power supply; a leakage transformer connected to the DC power supply; first semiconductor switching means connected in series to the DC power supply on a primary side of the leakage transformer; a first capacitor of the semiconductor switching means to resonate at d <br/> Nerugi over accumulated in the leakage transformer during oFF, the first second capacitor to provided to clamp the voltage of the capacitor and A series connection of a second semiconductor switching means, a power control circuit for alternately driving the first semiconductor switching means and the second semiconductor switching means, and a power control circuit on a secondary side of the leakage type transformer. Connected and the second
1st high voltage condenser and 2nd high voltage condenser
And a magnetron connected to the rectifier circuit, wherein the second capacitor and
The series connection of the second semiconductor switching means has one end thereof.
With the leakage type transformer and the first semiconductor switch.
The other end is located at the middle point of the
Is connected to the negative side, and the power control circuit controls the first semiconductor switching means and the first semiconductor switching means for controlling power.
Between conduction time of the fine said second semiconductor switching means is changed <br/>, on the first semiconductor switching means and said first high-voltage capacitor and the second high-voltage capacitor as the same kind having the same capacity A high-frequency heating apparatus having a configuration in which the anode current peak value of the magnetron is substantially the same during a period and an off period. 2. A DC power supply; a leakage transformer connected to the DC power supply; first semiconductor switching means connected in series with the DC power supply on a primary side of the leakage transformer; a first capacitor of the semiconductor switching means to resonate at d <br/> Nerugi over accumulated in the leakage transformer during oFF, the first second capacitor to provided to clamp the voltage of the capacitor and A series connection of a second semiconductor switching means, a power control circuit for alternately driving the first semiconductor switching means and the second semiconductor switching means, and a power control circuit on a secondary side of the leakage type transformer. Connected and the second
1st high voltage condenser and 2nd high voltage condenser
And a magnetron connected to the rectifier circuit, wherein the second capacitor and
The series connection of the second semiconductor switching means has one end thereof.
With the leakage type transformer and the first semiconductor switch.
The other end is located at the middle point of the
Is connected to the negative side, and the power control circuit controls the first semiconductor switching means and the first semiconductor switching means for controlling power.
Fine said second semiconductor switching means so between conduction time change <br/> is in, and the first high-voltage capacitor and the second high-voltage capacitor of to or used as the same kind or alone more used in the same capacity A high-frequency heating apparatus having a configuration in which an anode current peak value of the magnetron is substantially the same during an on period and an off period of the first semiconductor switching means. 3. A DC power supply; a leakage transformer connected to the DC power supply; first semiconductor switching means connected in series with the DC power supply on a primary side of the leakage transformer; a first capacitor of the semiconductor switching means to resonate at d <br/> Nerugi over accumulated in the leakage transformer during oFF, the first second capacitor to provided to clamp the voltage of the capacitor and A series connection of a second semiconductor switching means, a power control circuit for alternately driving the first semiconductor switching means and the second semiconductor switching means, and a power control circuit on a secondary side of the leakage type transformer. the first is connected
The first high voltage capacitor, the second high voltage capacitor and the first high voltage capacitor
Full-wave multiplier consisting of a voltage diode and a second high voltage diode
Pressure rectifier circuit, and a magnetron connected to the rectifier circuit, wherein the second capacitor and the second semiconductor
One end of the series connection of the switching means is connected to the leakage
And the first semiconductor switching means.
Connect the other end to the + or-side of the DC power supply at the intermediate point
Together are, the power control circuit is varied between conduction time of the first semiconductor switching means and said second semiconductor switching means for controlling the power, wherein the first high-voltage capacitor second high-voltage capacitor the by selecting a predetermined capacity, high-frequency heating apparatus with substantially the same current flows constituting the first high-voltage diode and the second high-voltage diode.