JPH0974760A - Voltage resonance type switching power circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To realize miniaturization/lightening and low cost of a switching power circuit and to improve a characteristics aspect such as extension of regulation range and power conversion efficiency, etc. SOLUTION: A power-factor improvement circuit formed of a filter choke coil LN inserted into a full-wave rectification line in series, a high-speed recovery type diode D2 , a filter capacitor CN forming the filter choke coil LN and a low-pass filter, and a resonance capacitor C2 connected to the high-speed recovery type diode D2 in parallel, is provided. Through electrostatic capacity coupling of a parallel resonance capacitor CrA forming a voltage resonance type, switching output is superposed upon a connection point between the filter choke coil LN and the high-speed recovery type diode D2 , and the switching output is fed-back to a rectified current path, so that a magnetic coupling transformer is omitted, for power-factor improvement.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a voltage resonance type switching power supply circuit for improving power factor, for example.

[0002]

2. Description of the Related Art In recent years, as a power supply device for rectifying a commercial power supply to obtain a desired DC voltage by developing a switching element capable of withstanding a relatively large current and voltage of a high frequency, a switching type power supply is mostly used. It is a device. The switching power supply is used as a power supply for various electronic devices as a high power DC-DC converter while increasing the switching frequency to reduce the size of a transformer and other devices.

In general, when a commercial power supply is rectified, the current flowing through the smoothing circuit has a distorted waveform, which causes a problem that the power factor indicating the efficiency of use of the power supply is impaired. Further, there is a need for a measure for suppressing harmonics generated due to the distorted current waveform.

Therefore, various inventions have been previously proposed by the applicant of the present invention configured to improve the power factor in the switching power supply circuit, and the circuit diagram of FIG. 7 is a switching circuit constructed based on these inventions. An example of the power supply circuit is shown. The switching converter of this power supply circuit is a self-excited voltage resonance type converter with one switching element. As a power factor improving means, the applicant of the present invention previously excited a voltage corresponding to the switching output from the primary side or the secondary side of an insulating converter transformer (hereinafter referred to as an insulating converter transformer) to the choke coil of the rectification line. A switching power supply circuit has been proposed in which a magnetic coupling transformer configured to do so is provided, and thereby a voltage of a switching cycle is fed back and superimposed on a rectified output of a bridge rectifier circuit to improve a power factor. The power supply circuit in this figure is configured by including the magnetic coupling transformer as described above.

In the switching power supply circuit shown in this figure, a common mode choke coil CMC and an across capacitor C _L are provided as a noise filter for removing common mode noise with respect to the AC power supply AC.
The AC power supply AC is full-wave rectified by a bridge rectifier circuit D ₁ including four diodes, and the rectified output is charged in a smoothing capacitor Ci via a power factor correction circuit 20.
The configuration and operation of the power factor correction circuit 20 will be described later.

In this figure, the voltage resonance type switching converter is constituted by one stone provided with a high withstand voltage switching element Q ₁ . This switching element Q ₁
Of the smoothing capacitor Ci via the starting resistor R _S.
Is connected to the positive electrode side of so that the base current at startup can be obtained from the rectifying and smoothing line. Further, a damping resistor R _B is provided between the base of the switching element Q ₁ and the ground.
A resonant circuit for self-excited oscillation including a resonant capacitor C _B and a drive winding N _B is connected in series. In this case, drive winding N
_B is an insulation converter transformer PIT (Power Isolatio
n Transformer), so that the required inductance for setting the switching frequency can be obtained. Further, the clamp diode D _D inserted between the base of the switching element Q ₁ and the negative electrode primary side (ground) of the smoothing capacitor Ci forms a path for the damper current flowing when the switching element Q ₁ is off. The collector of the switching element Q ₁ is connected to one end of the primary winding N ₁ of the insulating converter transformer PRT, and the emitter is grounded.

The resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q _1.
Is connected. The resonance capacitor Cr includes a primary winding N ₁ of an insulation converter transformer PIT, which will be described later, and a quadrature type control transformer PRT (Poewr Regulating Transforme).
The controlled winding N _{R of} r) and the inductance obtained by the series connection of the winding N _S of the magnetic coupling transformer MCT form a resonance circuit of the voltage resonance type converter. Although a detailed description is omitted here, when the switching element Q ₁ is off, the voltage V _CP across the resonance capacitor Cr actually becomes a sinusoidal pulse waveform due to the action of this resonance circuit, which is a voltage resonance type. You can get the action.

The insulating converter transformer PIT is for transmitting the switching output of the switching element Q _{1 to} the secondary side. In this case, the insulating converter transformer PIT is used.
One end of the primary winding N ₁ of T is connected to the collector of the switching element Q ₁ , and the other end is connected to the controlled winding N _R of the orthogonal control transformer PRT and the magnetic coupling transformer MC as shown in the figure.
It is connected in series with the winding N _{S of the} T.

On the secondary side of the insulating converter transformer PIT, the induced voltage induced in the secondary winding N ₂ by the primary winding N ₁ is a DC voltage by a half-wave rectifier circuit composed of a diode D ₃ and a smoothing capacitor C _3. Is converted into the DC output voltage E ₀ .

The control circuit 1 compares the DC voltage output E _O on the secondary side with a reference voltage and supplies a DC current corresponding to the error between them to the control winding N _C of the orthogonal control transformer PRT as a control current. Error amplifier. For example, when the DC output voltage E _O on the secondary side fluctuates, the control circuit 1 changes the current flowing through the control winding N _C.
The inductance L _R of the controlled winding N _R is changed. Since the controlled winding N _R forms the resonance circuit for obtaining the voltage resonance type switching operation as described above, the resonance condition of the resonance circuit is fixed with respect to the fixed switching frequency. Is changed. As a result, the secondary side DC voltage E _O can be made constant. Note that such a constant voltage control method will be referred to as an inductance control method hereinafter.

The power factor is improved by the power factor improving circuit 20. In the power factor correction circuit 20, as shown in the figure, the filter choke coil L _N , the fast recovery diode D ₂ , and the power recovery type diode D ₂ are connected to the line between the positive output terminal of the bridge rectifier circuit D _{1 and} the positive electrode of the smoothing capacitor Ci. Magnetic coupling transformer M acting as a choke coil for improving efficiency
A secondary winding N _{P of} CT is provided in series. The normal mode filter capacitor C _N has been inserted between the positive pole of the filter choke coil L _N and high-speed connection point of recovery diode D ₂ and the smoothing capacitor Ci, a filter capacitor C _N and filter choke coil L _N LC low pass filter is formed.

This LC low-pass filter is designed to prevent high frequency noise of the switching frequency from flowing into the AC line. Further, the high-speed recovery type diode D ₂ corresponds to a high-frequency current having a switching period described later flowing through the full-wave rectified output line.

The resonance capacitor C ₂ is connected in parallel with the secondary winding N _R of the magnetic coupling transformer MCT as shown in the figure, and forms a parallel resonance circuit together with the secondary winding N _R.
The resonance frequency of the parallel resonance circuit is set to be substantially the same as the resonance frequency of the switching power supply. The operation will be described later.

The magnetic coupling transformer MCT is provided to feed back the switching output to the rectification line side, and the primary winding N _S and the secondary winding N _P are formed by a ferrite core, for example, a 1: 1 winding. It is considered to be tightly coupled in a ratio. One end of the primary winding N _S of this magnetic coupling transformer MCT is connected in series with the primary winding N _{1 of the} insulating converter transformer PIT via the controlled winding N _R of the orthogonal control transformer PRT as described above. . As a result, the switching output obtained at the primary winding N ₁ of the insulating converter transformer PIT becomes the primary winding N _{S of the} magnetic coupling transformer MCT.
It is possible to transmit to. Also, magnetic coupling transformer MC
The other end of the primary winding N _S of T is grounded to the ground via the smoothing capacitor Ci.

The resonance capacitor C ₂ is connected in parallel with the secondary winding N _R of the magnetic coupling transformer MCT as shown in the figure, and forms a parallel resonance circuit together with the secondary winding N _R. The resonance frequency of the parallel resonance circuit is set to be substantially the same as the resonance frequency of the switching power supply. The operation will be described later.

The power factor improving operation is as follows. The switching voltage obtained at the primary winding N ₁ of the isolation converter transformer PIT is the orthogonal control transformer PRT.
Is supplied to the primary winding N _S of the magnetic coupling transformer MCT via the controlled winding N _R of the magnetic coupling transformer MCT. And the magnetic coupling transformer M
In CT, the switching voltage obtained in the primary winding N _S is excited in the secondary winding N _P. That is, the switching output is superimposed on the full-wave rectification line. Therefore, the full-wave rectified voltage rectified by the bridge rectifier circuit D ₁ is the secondary winding N of the magnetic coupling transformer MCT.
_The switching voltage is superposed by _P, and the smoothing capacitor Ci is charged. The terminal voltage of the smoothing capacitor Ci is lowered in the switching cycle by the superposition of the switching voltage. Then, the charging current starts to flow during the period when the terminal voltage of the capacitor Ci is lower than the rectified voltage level of the bridge rectifier circuit D ₁ , and the conduction angle of the average AC input current is expanded to approach the AC voltage waveform. This will improve the power factor.

In the power supply circuit for improving the power factor by such a method, the drive current of the insulating converter transformer PIT becomes small when the load is light, so that the switching current flowing in the full-wave rectified output line is also small due to this drive current. become. Therefore, since the level of the charging current becomes small when the load is light and the charging current becomes large when the load is heavy, the phenomenon that the terminal voltage of the smoothing capacitor Ci abnormally rises is solved especially when the load is light, and the normal MS is eliminated.
It is possible to improve regulation, which was difficult with the (magnet-switch) method. Therefore, for example, the fluctuation of the rectified and smoothed voltage Vi is suppressed even if the fluctuation of the _AC input voltage V _AC ± 20%, and it is not necessary to consider the improvement of the breakdown voltage of the switching element or the smoothing capacitor.

Further, in this circuit, an LC low-pass filter circuit (filter choke coil L _N and filter capacitor C _N ) is provided on the rectified output side of the bridge rectifier circuit D ₁ .

With such a structure, the secondary winding of the filter choke coil L _N , the fast recovery type diode D ₂ , and the magnetic coupling transformer MCT is provided in the line between the rectification output terminal of the bridge rectification diode D ₁ and the smoothing capacitor Ci. The line N _P is connected in series and inserted, but the value obtained by combining the resistance components of these elements is
By setting the inrush current at power-on to a level that can be suppressed to a required level, it becomes possible to omit the inrush current limiting resistor that is normally inserted in the AC line, and the power consumption is reduced. Since it is dispersed by the resistance component of each element, heat generation can be suppressed.

Then, one end of the filter capacitor C _N is not directly grounded to the ground, but the smoothing capacitor C _N is connected as shown in the figure.
Although it is connected to the positive electrode of i, the voltage applied across the filter capacitor C _N can be lower than when the LC low-pass filter circuit is inserted on the AC line side.

Further, the resonance capacitor C ₂ connected to the self-inductance of the secondary winding N _P of the magnetic coupling transformer MCT is a switching capacitor fed back to the rectifying and smoothing line when the load of the switching power supply becomes light. The voltage is suppressed, and it is possible to suppress an increase in the terminal voltage Ei of the smoothing capacitor Ci when the load is light.

Here, referring to the perspective view of FIG. 8, the magnetic coupling transformer M used in the switching power supply circuit of FIG.
The structural example of CT and filter choke coil L _N is shown. The magnetic coupling transformer MCT forms an EE type core in which E type cores C _R1 and C _R2 made of a ferrite material are combined so that their magnetic legs face each other as shown in FIG. A gap G is formed in the magnetic leg as shown in the figure. Then, for this central magnetic leg, the primary winding N
_The magnetic coupling transformer MCT is constructed by winding _S and the secondary winding N _P , respectively. Further, the filter choke coil L _N is constructed by winding a single wire such as a polyurethane copper wire directly around a drum-shaped ferrite core D without a bobbin as shown in FIG. 8B.

[0023]

From the viewpoint of the size and cost of the equipment, it is possible to reduce the number of parts in the switching power supply circuit as much as possible and use small or inexpensive parts to reduce the size / size. It is preferable to reduce the weight and cost. For example, if the magnetic coupling transformer MCT provided in the power supply circuit shown in FIG. 7, the flow input current I _O from the smoothing capacitor Ci to the primary winding N _S, the high speed recovery type in the secondary winding N _P Diode D ₂
As a result, a charging current to the smoothing capacitor Ci of a relatively large level, which is intermittent in the switching cycle, flows. Therefore, in the magnetic coupling transformer MCT shown in FIG. 8A, for example, a litz wire formed by bundling urethane-coated copper wires having a diameter of 60 μφ is wound. However, increasing the number of the bundles causes the winding to flow. The DC resistance of the current is suppressed, and the iron loss is reduced. Further, in the magnetic coupling transformer MCT, in order to reduce the iron loss, as shown in FIG. 8A, a gap G is formed with respect to the central magnetic leg to lower the magnetic flux density. In other words, the magnetic coupling transformer MCT is unavoidably made relatively large due to the need for the above-mentioned configuration, which hinders the promotion of miniaturization of the power supply circuit. Further, it is preferable to improve various electric characteristics such as expansion of the regulation range and power conversion efficiency.

[0024]

SUMMARY OF THE INVENTION Therefore, according to the present invention, in a switching power supply circuit having a voltage resonance type converter, a filter choke coil and a high speed recovery type rectification element which are inserted in series in a line between a rectification circuit and a smoothing capacitor. , A filter capacitor provided to form a low-pass filter together with a filter choke coil,
The switching converter is provided with a resonance capacitor connected in parallel with the high-speed recovery type rectifying element, and by capacitive coupling of the resonance capacitor forming a resonance circuit for making the operation of the switching converter a voltage resonance type. It was decided to provide a power factor correction circuit configured to feed back the output of the above to the line between the rectifier circuit and the smoothing capacitor.

Further, in the voltage resonance type converter provided with the voltage doubler rectifier circuit, a filter choke coil inserted in series with a commercial power source, and a filter capacitor provided so as to form a low pass filter together with the filter choke coil, A resonance capacitor that is provided in parallel with the rectifying element of the voltage doubler rectifier circuit and that forms a resonance circuit with the filter choke coil is provided.
A configuration in which the output of the switching converter is applied to a rectifying element forming a voltage doubler rectifying circuit by capacitive coupling of a resonance capacitor forming a resonance circuit for making the operation of the switching converter a voltage resonance type. did.

Further, a voltage provided with a bridge rectifying circuit for full-wave rectifying the commercial power source and a rectifying / smoothing circuit capable of obtaining a rectified / smoothed voltage by switching between a double voltage rectifying circuit for rectifying the double voltage of the commercial power source. In a resonant converter,
A filter choke coil that is inserted in series with a commercial power supply line, a filter capacitor that is provided so as to form a low-pass filter together with this filter choke coil, and a filter choke that is provided in parallel with a rectifying element that forms a rectifying and smoothing circuit. A resonance capacitor that forms a resonance circuit together with a coil is provided, and the output of the switching converter is rectified and smoothed by the capacitive coupling of the resonance capacitor that forms the resonance circuit for making the operation of the switching converter a voltage resonance type. It is configured to be applied to the rectifying element that forms the.

According to the above structure, the present invention provides switching in various types of voltage resonance type switching power supply circuits through capacitive coupling of a capacitor forming a resonance circuit for making the switching converter a voltage resonance type. Since the power factor is improved by feeding back the output to the current path on the rectifier circuit side and superimposing it, it becomes possible to eliminate the magnetic coupling transformer MCT.

[0028]

DESCRIPTION OF THE PREFERRED EMBODIMENTS The circuit diagram of FIG. 1 shows a switching power supply circuit according to an embodiment of the present invention. In this switching power supply circuit, as in the case of FIG. An excitation type voltage resonance type switching converter is used. The same parts as those in FIG. 7 are designated by the same reference numerals and the description thereof will be omitted.

The power factor correction circuit 10 shown in FIG.
Sea urchin bridge rectifier circuit D₁ Positive output and negative (primary side
Filter capacitor C between ground)_N Is inserted,
Bridge rectifier circuit D₁ One end of the positive output of the
Tachoch coil L_N Are connected. And the above
Luta choke coil L_N And filter capacitor C_N And to
A more normal mode LC low pass filter is formed.
And high frequency noise of the switching frequency on the AC line
It is designed to prevent inflow. Note that the filter
Yoke coil L_N Is the same as described above with reference to FIG.
It is assumed that the one used here is used. In addition,
Cover type diode D₂ The anode is a filter choke
Coil L _N Connected to the other end of the
It is connected to the positive electrode of the sensor Ci. That is, filter choke
Coil L_N And fast recovery type diode D₂ Is full-wave rectification
It is inserted in series with the line. This fast Rica
Burr type diode D₂ Is a switch that flows through the full-wave rectification line.
It is provided so as to correspond to the high frequency current of the pitching cycle. Further
Fast recovery type diode D₂ For resonance
Indexer C₂ Are connected in parallel to form a parallel connection circuit.
Have been. This resonance capacitor C₂ For example Phil
7 is used as a prior art.
Secondary winding N of magnetically coupled transformer MCT_P Connected in parallel
Resonance capacitor C₂ Having the same effect as
It is said.

In the power factor correction circuit 10 having such a connection form, the LC low pass filter circuit (the filter choke coil L _N and the filter capacitor C _N ) is provided on the rectified output side of the bridge rectifier circuit D ₁ . As a result, the filter choke coil L _N and the fast recovery diode D ₂ are connected in series and inserted in the line between the rectification output terminal of the bridge rectification diode D ₁ and the smoothing capacitor Ci. If the value obtained by synthesizing each resistance component of is set to a value that can suppress the inrush current when the power is turned on to the required level, the inrush current limiting resistor that should be inserted in the AC line can be omitted. Alternatively, it is possible to reduce the resistance, and the power consumption is dispersed by the resistance components of the above-mentioned elements, so that heat generation is also suppressed.

By connecting one end of the filter capacitor C _N to the positive electrode of the smoothing capacitor Ci without directly grounding it, the voltage applied to the both ends of the filter capacitor C _N is the LC low-pass filter circuit on the AC line side. It can be lower than that in the case of inserting into, and it is not necessary to adopt, for example, a safety standard certified product.

Further, the parallel resonance capacitor C shown in FIG.
r _A is the primary winding N ₁ of the isolation converter transformer PIT
And orthogonal control transformer PRT (Poewr Regulating Trans)
The inductance formed by the former controlled winding N _R forms a resonance circuit of the voltage resonance type converter, and, for example, a film capacitor is used.
For example, although the parallel resonant capacitor Cr in the circuit shown in FIG. 7 is provided between the collector of the switching element and the primary side ground, the parallel resonant capacitor Cr _A in the present embodiment is not limited to the switching element Q ₁ Is provided between the collector of the RRR circuit, the filter choke coil L _N in the RRR circuit 10 and the connection point of the fast recovery diode D ₂ . In this case, the filter choke coil L _N acts as a load for the switching output fed back via the parallel resonance capacitor Cr _A as described later. The parallel resonance capacitor C of FIG.
When the capacitances of r _A and the parallel resonance capacitor Cr of FIG. 7 are compared, Cr _A > Cr is selected.

According to the above-mentioned connection form,
An operation is obtained in which the charging / discharging current of the parallel resonance capacitor Cr _A is superimposed on the connection point between the filter choke coil L _N and the fast recovery diode D ₂ . That is, through the capacitive coupling of the parallel resonance capacitor Cr _A ,
The switching output of the switching element Q ₁ is fed back to the rectified voltage flowing through the inductance of the filter choke coil L _N. As a result, in substantially the same manner as described with reference to FIG. 7, the terminal voltage of the smoothing capacitor Ci is lowered in the switching cycle by the superposition of the switching voltage, the conduction angle of the AC input current is expanded, and the average waveform thereof is sinusoidal. Approaching the waves will improve the power factor. By the action of the noise filter including the common mode choke coil CMC and the across capacitor C _L and the normal mode low-pass filter including the filter choke coil L _N and the filter capacitor C _N , the commercial AC power supply AC has a high frequency component of the switching cycle. Will not flow in.

Then, the power factor correction circuit 1 having such a configuration.
0, the magnetic coupling transformer MCT, which is a relatively large component, is compared with the circuit shown in FIG.
Will be omitted, and the size of the substrate and the cost can be reduced accordingly. Further, in the present embodiment, the switching converter uses a self-excited voltage resonance type with one switching element. However, compared with a self-excited current resonance type switching converter, for example, at least a half bridge is used in the current resonance type switching converter. Since two switching elements connected to each other and a self-excited oscillation drive circuit for the two switching elements are required, the present embodiment can have a simpler circuit configuration.

FIG. 2 is a waveform diagram showing the operation of the main part of the power supply circuit of this embodiment in a switching cycle. In the case of the present embodiment, the time constant of the series resonance circuit for self-excited oscillation including the resonance capacitor C _B and the drive winding N _B is selected so that the switching frequency of the switching element Q ₁ is, for example, 50 KHz. There is. For example, if the switching element is turned off at time t ₁ shown in this figure,
The input current I to the smoothing capacitor Ci is produced by the action of the resonance circuit composed of the primary winding N _{1 of the} insulating converter transformer PRT, the combined inductance of the controlled winding N _R of the orthogonal control transformer PRT, and the capacitance of the parallel resonance capacitor Cr _{A. O} indicates the period t ₁ to t _{2 as} shown in FIG.
At, a current close to the resonance waveform will flow. Further, the waveforms shown in the next period t _{2 to} t ₃ of the same FIG. 2B are from the clamp diode D _D to the switching element Q ₁
Corresponding to the damper current flowing through the base-emitter of the. The voltage across the parallel resonance capacitor Cr (the potential between the collector and ground of the switching element Q ₁ ) V _CP when the switching element Q ₁ is off is shown in FIG.
The waveform becomes sinusoidal as shown in the period t _{1 to} t ₂ of (a), and the voltage resonance type operation is obtained. The switching element Q ₁ has a time point t ₃ at which the current I ₀ crosses zero.
Changes on in, but in the period t ₃ ~t ₄ is on period of the switching element Q _1, the voltage across V _CP of the resonance capacitor C ₁ is 0V continues as shown in FIG. 2 (a), the current I ₀ is the primary winding N ₁ and the controlled winding N _R
A waveform as shown in FIG. 2 (b) is obtained by the inductance action of. The charging / discharging current Ir of the parallel resonant capacitor Cr in the switching cycle has a waveform as shown in FIG.

FIG. 3 is a waveform diagram showing the operation of each part of the switching power supply circuit having the configuration shown in FIG. For example, when the AC input voltage V _AC is supplied as shown in FIG. 3A, the current I flowing through the fast recovery diode D ₂ in the full-wave rectification line
_As shown in FIG. 3B, ₂ is a high-speed recovery type due to the action of the fed-back switching output in the τ period when the absolute value of the _AC input voltage V _AC is higher than the voltage Ei across the smoothing capacitor Ci. The switching operation of the diode D ₂ results in a waveform in which high frequency components of the switching cycle are superimposed. Further, the voltage V ₂ corresponding to the voltage across the resonance capacitor C ₂ has a waveform shown by a solid line on which a high frequency component of the switching cycle is superimposed as shown in FIG. By the way, the rectified and smoothed voltage Ei _A (voltage across the smoothing capacitor Ci) shown in FIG. 7 described above as the prior art is different from the full-wave rectified voltage V ₁ which is the output voltage of the bridge rectifier circuit D ₁ with respect to the magnetic coupling transformer. The average value of the voltage values obtained by superimposing the high frequency voltage of the switching cycle by the secondary winding N _P of the MCT is shown in FIG.
The level is as shown in (c). On the other hand,
The rectified and smoothed voltage Ei in the power supply circuit is the peak value of the voltage V ₂ across the resonance capacitor C ₂ . Therefore, as shown in FIG. 3C, the rectified smoothed voltages Ei and E
When i _A is compared, the relationship of Ei> Ei _A is obtained, and the DC input voltage to the switching converter is increased, which expands the regulation range. For example, in the power supply circuit of FIG. 7, the regulation range is compensated by the action of the magnetic coupling transformer MCT, but in the case of the present embodiment, the regulation range is compensated even if the magnetic coupling transformer MCT is omitted. become. Then, as shown in FIG. _3D , the AC input current I _AC is made to flow in the τ period, which causes the average of the AC input current to approach the AC input voltage waveform. The conduction angle is enlarged to the extent that the power factor is improved. For example, specifically, as shown in the figure, when the half cycle of the commercial AC power supply is 10 ms, the inductance of the filter capacitor L _N and the capacitance of the resonance capacitor C ₂ should be selected so that the τ period becomes 5 ms. For example, a power factor of about 0.8 can be obtained.

Next, the circuit diagram of FIG. 4 shows the configuration of a switching power supply circuit according to another embodiment of the present invention. The same parts as those in FIG. In this switching power supply circuit, the orthogonal control transformer PR
T is provided on the secondary side. The controlled winding N _R of the orthogonal control transformer PRT is connected in series with the secondary winding N ₂ of the insulating converter transformer PIT, and the other end thereof is connected to the anode of the rectifying diode D _3. It Further, in this case, the secondary winding N ₂ of the insulation converter transformer PIT and the orthogonal control transformer PRT are connected.
A resonance capacitor C _O is provided in parallel with the controlled winding N _R connected in series. This capacitor C _O is
The capacitance and the inductance component of the secondary winding N ₂ and the controlled winding N _R form a resonance circuit on the secondary side so that voltage resonance can be obtained and the secondary side is excited. The output voltage waveform is also designed to approach a sine wave. As a result, for example, the capacitor C
Compared to the case where _O is not provided, it is possible to cope with a wider range of load fluctuations. For example, in the case of the circuit of FIG. 1, constant voltage control by the inductance control method is performed on the primary side of the insulating transformer PIT, whereas in this case, the control circuit 1 outputs the DC output voltage E. By supplying a control current varied based on _O to the control winding N _C , the inductance of the resonance circuit is controlled on the secondary side.
Even in this way, constant voltage control can be performed (secondary side inductance control method). The configuration of the power factor correction circuit 10 in this embodiment is the same as that shown in FIG. 1, and the magnetic coupling transformer M as described above is used.
The power factor is improved and the regulation range and the power conversion efficiency are also improved by the configuration in which the CT is omitted.

FIG. 5 is a circuit diagram showing the configuration of a switching power supply circuit according to still another embodiment of the present invention. The same parts as those in FIGS. 1 and 4 are designated by the same reference numerals and the description thereof will be omitted.
The switching power supply circuit of this embodiment is suitable for being applied to a power supply circuit corresponding to a so-called wide range corresponding to AC input voltage AC100V system to AC200V system.

In this drawing, an LC low pass filter including a filter choke coil and a filter capacitor C _N is provided for the AC power supply AC. Also,
In the present embodiment, the bridge rectifier circuit D ₁ includes four fast recovery type diodes D _F1 , D _F2 , D _F3 and D as shown in the figure.
It is formed by bridge-connecting _F4 . 1 and 4
The resonance capacitor C ₂ shown in (1) corresponds to the case where a voltage doubler rectifier circuit is formed as described later in this JJK.
It is divided into two resonance capacitors C _2A and C _2B and connected in parallel to the fast recovery type diodes D _F1 and D _F2 , respectively. Then, the power factor correction circuit 11 of the present embodiment is formed as shown in the drawing depending on the connection form of these elements described above. In this case, the parallel resonance capacitor Cr _A is connected to the collector of the switching element Q ₁ and the connection point of the fast recovery type diodes D _F1 and D _F2 . Further, the positive pole rectification output terminal of the bridge rectification circuit D ₁ (connection point of fast recovery type diodes D _F2 and D _F4 )
Smoothing capacitors Ci _A and Ci _B are provided in series between the smoothing capacitors Ci _A and Ci _B , and one pole of the AC power supply AC is connected to the switch SW with respect to the connection point of the smoothing capacitors Ci _A and Ci _B.
Connected via ₁ . The switch SW ₁ is actually a hybrid IC using an electromagnetic power relay or a triac, and is controlled to be turned on when the AC input voltage is 100V system and turned off when the AC input voltage is 200V system. The operation of the power supply circuit associated with the on / off switching of the switch SW ₁ will be described later.

In this case, the switching element Q ₁
Is driven by a drive transformer DT configured by winding a drive winding N _B around a power choke coil PCC. The power choke coil PCC is connected to the rectified smoothed voltage line (the positive terminal of the smoothing capacitor Ci _A ) and the end of the primary winding N ₁ of the orthogonal converter transformer PRT as shown in the figure. Therefore, in the case of the present embodiment, the resonance circuit for the voltage resonance type is the parallel resonance capacitor C
r _A , the primary winding N ₁ and the power choke coil P
It will be formed by the combined inductance of CC.

Further, in the present embodiment, the winding direction is orthogonal to the primary and secondary windings N ₁ and N ₂ of the insulating converter transformer PRT for transmitting the switching output to the secondary side. Since the control winding N _C is provided in this manner, the configuration is such that it also serves as an orthogonal type control transformer. A secondary winding N ₂ provided with a center tap on the secondary side of the insulating converter transformer PRT.
On the other hand, a double-wave rectification circuit composed of rectification diodes D _3A and D _3B and a smoothing capacitor C ₃ is provided to obtain a DC output voltage E _O. A parallel resonance capacitor C _O is connected to the secondary winding N ₂ to form a resonance circuit. In this case, the converter circuit P is controlled by the control circuit 1 according to the fluctuation of the DC output voltage E _O.
By supplying the control current to the control winding N _C of the RT, the leakage magnetic flux of the converter transformer PRT is changed. As a result, the resonance condition of the resonance circuit composed of the secondary winding N ₂ and the parallel resonance capacitor C _O is varied, and constant voltage control is performed.

The rectifying / smoothing operation on the primary side of the switching power supply circuit compatible with a wide range configured as described above is as follows. For example, when the AC input voltage is 200 V, the switch SW ₁ is turned off as described above. In this case, the commercial AC power supply AC is a bridge rectifier circuit D
With the full-wave rectified by _1, the series connected smoothing capacitors Ci _A, a current path of the rectifying and smoothing circuit that charge and discharge according to the full-wave rectified output is made relative to Ci _B is formed. As a result, a voltage corresponding to the AC input voltage is obtained as the rectified and smoothed voltage Ei (voltage across the smoothing capacitors Ci _A and Ci _B connected in series).

On the other hand, when the AC input voltage is set to 100 V, the switch SW ₁ is turned on, so that the fast recovery type diode among the four fast recovery type diodes forming the bridge rectifier circuit D ₁ is used. Diodes D _F1 and D
_A voltage doubler rectifier circuit using _F2 is formed. That is, the charging path during the positive period of the AC power supply AC is AC power supply AC → winding a (CMC) → filter choke coil L _N → rectifying diode D ₁₁ → smoothing capacitor Ci _A → winding b (CMC).
→ The AC power supply AC is used, and the smoothing capacitor Ci _{A is} charged by this current path. In addition, AC power supply A
The charging path during the period when C is negative is AC power supply AC → winding b (C
MC) → smoothing capacitor Ci _B → rectifier diode D ₁₂ →
The filter choke coil L _N → winding a (CMC) → AC power source AC, and the smoothing capacitor Ci _B is charged. As a result, as the rectified smoothed voltage, the voltages across the smoothing capacitors Ci _A and Ci _B are combined,
Double voltage will be obtained. As a result, as the rectified and smoothed voltage Ei, a voltage value of 200V system which is almost twice the AC input voltage is obtained. As described above, in the present embodiment, the switch SW ₁ is configured to perform the double voltage rectifying and smoothing operation when the input commercial AC power source AC is the 100V system and the normal full-wave rectifying and smoothing operation when the commercial AC power source AC is the 200V system. The automatic power supply switching is configured so that a switching power supply circuit corresponding to a wide range AC input voltage can be obtained.

In the present embodiment, the switching element Q is connected to the connection point of the fast recovery type diodes D _F1 and D _F2 in the power factor correction circuit 11 described above.
The collector of ₁ is connected via the parallel resonance capacitor Cr _A , whereby the switching output of the switching element Q ₁ is fed back to the path of the rectified current via the capacitive coupling of the parallel resonance capacitor Cr _A. To be superimposed. Therefore, the power factor is improved by the same operation as that of the power factor correction circuit 10 of FIGS. 1 and 4 described so far, and the regulation range is expanded and the power conversion efficiency is improved. Further, in the configuration of the present embodiment, the power factor correction circuit 11 is designed to improve the power factor in both cases of the normal full-wave rectifying and smoothing operation and the double voltage rectifying and smoothing operation. The noise due to the high frequency current of the switching cycle flowing in the rectification current path is caused by the LC low pass filter including the filter choke coil L _N and the filter capacitor C _N , the common mode choke coil CMC and the across capacitor C _L.
The noise is prevented by the noise filter composed of the above, and noise is prevented from flowing into the commercial AC power supply AC.

FIG. 6 is a circuit diagram showing still another embodiment of the present invention. The same parts as those in the circuit diagrams of FIGS. 1, 4, and 5 showing the above-described respective embodiments are designated by the same reference numerals, and the description thereof will be omitted. The switching power supply circuit of the present embodiment corresponds to the condition that the AC input voltage is 100 V and the load power is 150 W or more, which is a relatively heavy load. Therefore, in this embodiment, a voltage doubler rectifier circuit and a voltage resonance type switching converter that operates two switching elements by push-pull coupling are employed as described below.

In the power factor correction circuit 12 of this figure, the LC low pass filter (L _N , C) is connected to the AC power source AC.
_N ) are provided. Then, one pole of the AC power supply AC is connected to the connection point of the anode of the rectifier diode D ₁₁ for double voltage rectification and the cathode of D ₁₂ via the filter choke coil L _N , and the other pole is connected to the smoothing capacitor. C
It is connected to the connection point of i _A and Ci _B. The smoothing capacitors Ci _A and Ci _B are connected in series between the rectifying smoothing line and the ground as shown in the figure. The cathode side of the rectifying diode D ₁₁ is a smoothing capacitor Ci _A.
Of the rectifier diode D ₁₂ is connected to the ground.

In the circuit of this embodiment, the filter choke coil L _N acts as a load for the switching output fed back from the switching converter, and the switching output is superimposed on the rectifying diodes D ₁₁ and D ₁₂ as described later. Is configured. Corresponding to this, in this embodiment, the rectifier diodes D ₁₁ and D ₁₂ are of the fast recovery type. Further, resonance capacitors C _2A and C _2B are connected in parallel to the rectifier diodes D ₁₁ and D ₁₂ , respectively, and a part of the switching output is bypassed by the capacitors C _2A and C _2B . As described above, in the present embodiment, the power factor correction circuit 12 as shown in the figure is combined with the voltage doubler rectifier circuit, and this circuit configuration has the power supply circuit previously shown in FIG. In, the switch SW ₁ is turned on to form a circuit equivalent to the case where the voltage doubler rectifier circuit is formed. Therefore, the voltage doubler operation is the same as the operation described above in the embodiment of FIG. 5, and the description thereof is omitted here.

Further, the switching converter is formed as a self-oscillation type voltage resonance type switching converter in which two stones of the switching element Q ₁ and the switching element Q ₂ are driven by push-pull, and corresponds to the condition under heavy load. I am trying. The drive winding N _B , the resonance capacitor C _B , the resistor R _B , and the clamp diode D _D provided for the switching element Q ₂ are denoted by the same reference numerals and the description thereof is omitted. In this case, the insulating converter switching element Q ₁ with respect to the transformer PIT, the drive windings of Q ₂ N _B, the N _B is wound. The collector of the switching element Q ₁ is a parallel resonance capacitor C
It is connected to the connection point of the rectifying diodes D ₁₁ and D ₁₂ via r _A, and the collector of the switching element Q ₂ is connected to the rectifying diode D ₁₁ via the parallel resonance capacitor Cr _A.
And D ₁₂ are connected to each other. This allows
The switching outputs of the switching elements Q ₁ and Q ₂ are fed back to the rectified current path via the capacitive coupling of the parallel resonance capacitors Cr _A and Cr _B , respectively, and are superimposed, and the switching outputs of the circuit shown in FIG. The power factor is improved by the same action as the case.

Further, in this case, an orthogonal control transformer PRT in which the control winding N _C and the controlled winding N _R are wound so that their winding directions are orthogonal to each other is provided. The controlled winding N _R of the orthogonal control transformer PRT is inserted between the center tap of the primary winding N ₁ of the insulating converter transformer PIT and the line of the rectified and smoothed voltage Ei (the positive terminal of the smoothing capacitor Ci _A ). The controlled winding N _R and the isolated converter transformer PIT are thereby provided.
Primary winding N ₁ and parallel resonant capacitors Cr _A , Cr _B
And form a voltage resonance type resonance circuit.
Then, the control circuit 1 controls the secondary side DC output voltage E _O.
Is supplied to the control winding N _C of the quadrature type control transformer PRT, the same operation as the primary side inductance control method described with reference to FIG. The inductance of the primary winding N ₁ of the converter transformer PIT is varied so that constant voltage control is performed.

In this embodiment, two switching elements are provided for heavy loads,
For example, in the case of handling a heavy load with a current resonance type, a switching converter of four full bridge couplings is configured, and thus the present embodiment has a simpler circuit configuration.

Further, the voltage resonance type switching power supply circuit of the present invention comprises a power factor correction circuit connection configuration, a constant voltage control method,
The number of switching elements of the voltage resonance type converter is not limited to the combinations shown in the above-described embodiments, and the invention can be applied to a power supply circuit configured by other combination patterns. Further, although not shown here, the present invention can be applied to a voltage resonance type converter by a separately excited type. Further, other circuit parts such as multiple secondary side DC output voltage may be changed according to actual use conditions.

[0052]

As described above, the voltage resonance type switching power supply circuit of the present invention, including the one using the voltage doubler rectifier circuit, is essentially the electrostatic capacitance of the resonance capacitor of the resonance circuit for obtaining the voltage resonance type operation. Since the switching output is fed back to the path of the rectified current via capacitive coupling and superimposed, the power factor is improved, so the number of components is reduced compared to the case where the power factor is improved by the magnetic coupling transformer. Therefore, the present invention has an effect that it is possible to promote the miniaturization of the circuit size and the cost reduction in addition to the fact that the present invention has a simple circuit configuration of the voltage resonance type. Further, by removing the magnetic coupling transformer, the power loss is also reduced, so that the power conversion efficiency is improved, and the level of the rectified and smoothed voltage is increased, so that the regulation range is expanded.

[Brief description of drawings]

FIG. 1 is a circuit diagram of a voltage resonance type switching power supply circuit according to an embodiment of the present invention.

FIG. 2 is a waveform diagram showing the operation of the voltage resonance type switching power supply circuit according to the present embodiment in a switching cycle.

FIG. 3 is a waveform diagram showing an operation of the voltage resonance type switching power supply circuit according to the present embodiment in a commercial power supply cycle.

FIG. 4 is a circuit diagram showing a voltage resonance type switching power supply circuit of another embodiment.

FIG. 5 is a circuit diagram showing a voltage resonance type switching power supply circuit as still another embodiment.

FIG. 6 is a circuit diagram showing a voltage resonance type switching power supply circuit as still another embodiment.

FIG. 7 is a circuit diagram showing a voltage resonance type switching power supply circuit as a prior art.

FIG. 8 is a perspective view showing structures of a magnetic coupling transformer and a filter choke coil.

[Explanation of symbols]

1 Control circuit 10, 11, 12 Power factor correction circuit L _N Filter choke coil C _N filter capacitor D ₁ Bridge rectifier circuit D ₂ Fast recovery type diode CH Choke coil PIT Isolation converter transformer PRT Quadrature control transformer Q ₁ , Q ₂ switching Element Ci, Ci _A , Ci _B Smoothing capacitor Cr ₁ , Cr _A , C _1B Parallel resonance capacitor N ₁ Primary winding C ₂ Resonance capacitor C _O Resonance capacitor

Claims (9)

[Claims] 1. A rectifying means for rectifying a commercial power source, a smoothing means for smoothing an output of the rectifying means, and a voltage outputted from the smoothing means for intermittently outputting to a primary winding of an insulating converter transformer. Switching means, a resonance circuit formed by the inductance component including the primary winding of the insulating converter transformer and the capacitance of the resonance capacitor to make the operation of the switching means a voltage resonance type, and the capacitive coupling of the resonance capacitor. A power factor improving circuit for returning the output of the switching means to the line between the rectifying means and the smoothing means, and the power factor improving circuit outputs the rectified output current of the rectifying means by the output of the switching means. The high-speed recovery type rectifying element that connects and disconnects the Voltage resonance type switching power supply circuit, characterized in that it is configured with a capacitor. 2. The voltage resonance type according to claim 1, wherein a low pass filter formed so as to block a switching output flowing into a commercial power supply line is provided on an input side of the power factor correction circuit. Switching power supply circuit. 3. A voltage doubler rectifying means for outputting a rectified and smoothed voltage obtained by rectifying a voltage of a commercial power source, and a voltage output from the voltage doubler rectifying means is intermittently output to a primary winding of an insulating converter transformer. Formed by the inductance component including at least the primary winding of the insulating converter transformer and the capacitance of the resonance capacitor to make the operation of the switching device a voltage resonance type, and by the capacitive coupling of the resonance capacitor. A resonance circuit provided so as to apply the output of the switching means to a rectifying element forming the voltage doubling rectification means, a filter choke coil inserted in series with the commercial power supply in a line, and the filter choke coil A filter filter provided to form a low-pass filter. A voltage resonance type switching characterized by comprising a capacitor and a resonance capacitor which is provided in parallel with the rectifying element of the voltage doubler rectifying / smoothing means and which forms a resonance circuit together with the filter choke coil. Power supply circuit. 4. The voltage resonance type switching power supply circuit according to claim 3, wherein a high-speed recovery type rectifying element is used in the voltage doubling rectifying means. 5. A rectifying / smoothing means capable of obtaining a rectified / smoothed voltage by enabling switching between a bridge rectifier circuit for full-wave rectifying a commercial power source and a voltage doubler rectifier circuit for rectifying a commercial power source by a double voltage, The switching means is adapted to intermittently output the voltage output from the voltage rectifying / smoothing means to the primary winding of the insulating converter transformer, and the inductance component including at least the primary winding of the insulating converter transformer and the capacitance of the resonance capacitor. Resonance provided so as to make the operation of the switching means a voltage resonance type, and to apply the output of the switching means to a rectifying element forming the rectifying and smoothing means by capacitive coupling of the resonance capacitor. A circuit and a filter choke coil inserted in series with the line of the commercial power supply, A filter capacitor provided so as to form a low-pass filter together with the filter choke coil; a resonance capacitor provided in parallel with the rectifying element forming the rectifying / smoothing means to form a resonance circuit together with the filter choke coil; A voltage resonance type switching power supply circuit comprising: 6. The high-speed recovery type rectifying element is used in the rectifying / smoothing means.
The voltage resonance type switching power supply circuit described in. 7. A capacitor is connected in parallel to the secondary winding of the insulation converter transformer, and a resonance circuit is formed by the inductance component of the insulation converter transformer. The voltage resonance type switching power supply circuit according to any one of claims 1 to 6. 8. The voltage resonance type switching power supply circuit according to claim 1, wherein the switching means is configured by push-pull coupling two switching elements. 9. A constant voltage control means configured to perform constant voltage control by varying the inductance of the insulating converter transformer based on a DC output voltage obtained on the secondary side of the insulating converter transformer. 9. The voltage resonance type switching power supply circuit according to claim 1, wherein: