JPH09307479A - Spread spectrum signal synchronizing device - Google Patents
Spread spectrum signal synchronizing deviceInfo
- Publication number
- JPH09307479A JPH09307479A JP8116169A JP11616996A JPH09307479A JP H09307479 A JPH09307479 A JP H09307479A JP 8116169 A JP8116169 A JP 8116169A JP 11616996 A JP11616996 A JP 11616996A JP H09307479 A JPH09307479 A JP H09307479A
- Authority
- JP
- Japan
- Prior art keywords
- complex baseband
- baseband signal
- correlation
- code
- symbol
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
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- Synchronisation In Digital Transmission Systems (AREA)
Abstract
Description
【０００１】[0001]
【発明の属する技術分野】この発明は例えばＣＤＭＡ
（符号分割多元接続）移動通信の基地局よりの信号に拡
散符号を同期させるために用いられるスペクトラム拡散
信号同期装置に関する。BACKGROUND OF THE INVENTION 1. Field of the Invention
(Code Division Multiple Access) The present invention relates to a spread spectrum signal synchronizer used to synchronize a spread code with a signal from a base station of mobile communication.
【０００２】[0002]
【従来の技術】ＣＤＭＡデジタルセルラ電話の規格であ
るＥＩＡ／ＴＩＡ／ＩＳ−９５ Ｍｏｂｉｌｅ Ｓｔａ
ｔｉｏｎ−Ｂａｓｅ Ｓｔａｔｉｏｎ Ｃｏｍｐａｔｉ
ｂｉｌｉｔｙ Ｓｔａｎｄａｒｄ ｆｏｒ Ｄｕａｌ−
Ｍｏｄｅ ＷｉｄｅｂａｎｄＳｐｒｅａｄ Ｓｐｅｃｔ
ｒｕｍ Ｃｅｌｌｕｌａｒ Ｓｙｓｔｅｍ（以下ＩＳ−
９５と記す）で基地局の送信信号の波形品質が規格化さ
れている。この波形品質の測定において入力スペクトル
拡散信号の拡散符号に、測定装置の拡散符号を同期させ
る必要がある。2. Description of the Related Art EIA / TIA / IS-95 Mobile Station, which is a standard for CDMA digital cellular telephones.
Tion-Base Station Compati
beauty Standard for Dual-
Mode Wideband Spread Spec
rum Cellular System (hereinafter IS-
95)), the waveform quality of the transmission signal of the base station is standardized. In measuring the waveform quality, it is necessary to synchronize the spreading code of the measuring device with the spreading code of the input spread spectrum signal.
【０００３】ＩＳ−９５規格の基地局の送信信号のうち
の１つのチャネルはパイロットチャネルと呼ばれ、この
パイロットチャネルの信号は同相成分（Ｉ）のＰＮ符号
と直交成分（Ｑ）のＰＮ符号とによりＱＰＳＫ変調した
ものであって、Ｉ成分のＰＮ符号とＱ成分のＰＮ符号は
ゴールド符号と呼ばれ、相互相関が一様に著しく小さな
値である。One of the transmission signals of the base station of the IS-95 standard is called a pilot channel, and the signal of this pilot channel is composed of a PN code of an in-phase component (I) and a PN code of a quadrature component (Q). The PN code of the I component and the PN code of the Q component, which are QPSK-modulated by, are called Gold code, and the cross-correlation is a uniformly small value.
【０００４】このパイロット信号の拡散符号に同期させ
るため従来においては図２に示す構成で行われていた。
入力端子１１から中間周波に変換されたパイロット信号
が入力され、Ａ／Ｄ変換部１２でその拡散符号に対しオ
ーバサンプリング状態でデジタル値系列に変換される。
この変換におけるサンプリング周波数はパイロット信号
の拡散符号（ＰＮ符号）のチップ周波数の８倍、つまり
４倍のオーバサンプリングとされる。In order to synchronize with the spread code of the pilot signal, the configuration shown in FIG. 2 has been conventionally used.
The pilot signal converted into the intermediate frequency is input from the input terminal 11, and is converted into a digital value sequence in the A / D converter 12 in the oversampling state with respect to the spread code.
The sampling frequency in this conversion is 8 times the chip frequency of the spread code (PN code) of the pilot signal, that is, 4 times oversampling.
【０００５】このデジタル信号系列は直交変換部１３で
直交変換され、同相成分Ｉ1 （ｋ）と直交成分Ｑ
1 （ｋ）よりなる複素ベースバンド信号系列に変換され
る。一方拡散符号生成部１４からパイロット信号のＩ成
分のＰＮ符号（拡散符号）ＩR （ｋ）と、Ｑ成分のＰＮ
符号（拡散符号）ＱR （ｋ）の系列が生成され、各同相
成分Ｉ1 （ｋ）とＩR （ｋ）との相関が相関計算部１５
で計算され、各直交成分Ｑ1（ｋ）とＱR （ｋ）との相
関が相関計算部１６で計算され、同相成分Ｉ1 （ｋ）と
直交成分ＱR （ｋ）との相関が相関計算部１７で計算さ
れ、直交成分Ｑ1 （ｋ）と同相成分ＩR （ｋ）との相関
が相関計算部１８で計算される。相関計算部１５，１６
の各計算結果の差が加算部１９で演算され、相関計算部
１７，１８の各計算結果の和が加算部２１で演算され、
これ等加算部１９，２１の各出力はそれぞれ乗算部２
２，２３で自乗演算がされた後、加算部２４で加算され
る。このようにして信号Ｉ1 （ｋ），Ｑ1 （ｋ）と拡散
符号ＩR （ｋ），ＱR （ｋ）との複素相関が計算され
る。This digital signal sequence is orthogonally transformed by the orthogonal transformation unit 13, and the in-phase component I 1 (k) and the orthogonal component Q are obtained.
It is converted into a complex baseband signal sequence consisting of 1 (k). On the other hand, the spread code generator 14 outputs the I component PN code (spread code) I R (k) of the pilot signal and the Q component PN.
Code (spreading code) Q R series (k) is generated, the correlation is the correlation calculator 15 and the phase component I 1 (k) and I R (k)
In is calculated, the correlation between each quadrature component Q 1 (k) and Q R (k) is calculated by the correlation calculation unit 16, correlation is a correlation between in-phase component I 1 (k) and the quadrature component Q R (k) The correlation is calculated by the calculation unit 17, and the correlation between the quadrature component Q 1 (k) and the in-phase component I R (k) is calculated by the correlation calculation unit 18. Correlation calculation units 15 and 16
The difference between the calculation results of 1 is calculated by the adder 19, the sum of the calculation results of the correlation calculators 17 and 18 is calculated by the adder 21,
The outputs of the adders 19 and 21 are respectively supplied to the multiplier 2
After the square operation is performed in 2 and 23, the addition is performed in the adding unit 24. Signal I 1 in this manner (k), Q 1 (k ) and the spread code I R (k), the complex correlation between Q R (k) is calculated.
【０００６】この複素相関の計算値Ａ（ｋ）が最大にな
るように、つまりＡ（ｋ）がピーク値となるように拡散
符号生成部１４の生成位相が１チップづつ順次シフトさ
れる。The generated phase of the spread code generator 14 is sequentially shifted by one chip so that the calculated value A (k) of the complex correlation becomes maximum, that is, A (k) becomes a peak value.
【０００７】[0007]
【発明が解決しようとする課題】相関計算部１５〜１８
では各サンプル値系列系に対して１サンプルずつシフト
しながら計算するのでＦＩＲフィルタと同じ構造である
ため、この相関処理に時間がかかるという問題があっ
た。Correlation calculation units 15 to 18
However, since the calculation is performed while shifting by one sample for each sample value series system, it has the same structure as the FIR filter, so there is a problem that this correlation processing takes time.
【０００８】[0008]
【課題を解決するための手段】この発明によれば、入力
スペクトラム拡散信号の複素ベースバンド信号系列から
その直交変調のシンボル判定点が推定され、複素ベース
バンド信号系列中から前記推定シンボル判定点に近いサ
ンプルが取出され、その取出されたものと拡散符号との
相関が計算される。According to the present invention, a symbol decision point for orthogonal modulation is estimated from a complex baseband signal sequence of an input spread spectrum signal, and the estimated symbol decision point is determined from the complex baseband signal sequence. A close sample is taken and the correlation between the taken and the spreading code is calculated.
【０００９】この場合、入力スペクトラムに拡散信号が
前記ＣＤＭＡ移動通信のパイロット信号の場合は、複素
ベースバンド信号系列中の実部（同相成分）についての
みのチップ判定点付近の拡散符号との相関計算がなされ
る。In this case, when the spread signal in the input spectrum is the pilot signal of the CDMA mobile communication, the correlation calculation with the spread code near the chip decision point of only the real part (in-phase component) in the complex baseband signal sequence is calculated. Is done.
【００１０】[0010]
【発明の実施の形態】この発明を前記ＣＤＭＡ移動通信
の基地局のパイロット信号に拡散符号を同期させる場合
に適用した実施例を図１Ａに、図２と対応する部分に同
一符号を付けて示す。この発明では複素ベースバンド信
号Ｉ1 （ｋ），Ｑ1 （ｋ）から、そのＱＰＳＫ変調のシ
ンボルを判定するシンボル判定点がシンボル判定点推定
部３１で推定される。具体的には複素ベースバンド信号
の各サンプルＩ1 （ｋ），Ｑ1（ｋ）がそれぞれ乗算部
３２，３３で自乗され、これら自乗演算結果が加算部３
４で加算され、つまり複素ベースバンド信号Ｉ
1 （ｋ），Ｑ1 （ｋ）の振幅の自乗系列を得、これを離
散的フーリエ変換部３５で離散的フーリエ変換され、そ
の変換結果中のシンボル（チップ）周波数成分が抽出さ
れ、その抽出成分の逆正接（ａｒｃｔａｎ）が逆正接部
３６で求められ、そのゼロとなる点の一方がシンボル判
定点と推定される。DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS An embodiment in which the present invention is applied to a case where a spreading code is synchronized with a pilot signal of a base station of the CDMA mobile communication is shown in FIG. 1A, and parts corresponding to those in FIG. . In the present invention, the symbol decision point estimation unit 31 estimates the symbol decision point for deciding the symbol of the QPSK modulation from the complex baseband signals I 1 (k) and Q 1 (k). Specifically, the samples I 1 (k) and Q 1 (k) of the complex baseband signal are squared by the multiplication units 32 and 33, respectively, and the squared calculation results are added by the addition unit 3.
4 added, that is, the complex baseband signal I
The squared series of the amplitudes of 1 (k) and Q 1 (k) is obtained, this is subjected to discrete Fourier transform in the discrete Fourier transform unit 35, and the symbol (chip) frequency component in the transform result is extracted and extracted. The arctangent (arctan) of the component is obtained by the arctangent unit 36, and one of the zero points is estimated to be the symbol decision point.
【００１１】前記例では複素ベースバンド信号系列のサ
ンプリング周波数はチップ周波数の８倍とされている。
この場合は、ｓｉｎ，ｃｏｓの値として、例えばｃｏｓ
について１，０．７０７，０，−０．７０７，−１，−
０．７０７，０，０．７０７の８値を順次、Ｉ1 （ｋ）
2 ＋Ｑ1 （ｋ）2 の系列に順次乗算し、同様にｓｉｎの
８値を順次Ｉ1 （ｋ）2 ＋Ｑ1 （ｋ）2 の系列に順次乗
算すれば離散的フーリエ変換の結果のチップ周波数成分
が得られる。In the above example, the sampling frequency of the complex baseband signal sequence is 8 times the chip frequency.
In this case, as the values of sin and cos, for example, cos
About 1,0.707,0, -0.707, -1,-
Eight values of 0.707, 0, 0.707 are sequentially set to I 1 (k)
If the sequence of 2 + Q 1 (k) 2 is sequentially multiplied, and the eight values of sin are also sequentially multiplied by the sequence of I 1 (k) 2 + Q 1 (k) 2 , the chip frequency of the discrete Fourier transform result is obtained. The ingredients are obtained.
【００１２】推定部３１で推定されたチップ判定点に近
いサンプルが複素ベースバンド信号系列中の実部（同相
成分）Ｉ1 （ｋ）から間引き部３７で取出される。実部
Ｉ1（ｋ）の系列の各サンプルが例えば図１Ｂに示す×
点位置であり、推定されたシンボル判定点が○点位置で
あったとすると、×点位置のサンプル中の、○点位置に
近いものを各○点ごとに１つ図１Ｃに示すように取出
す。サンプリング周波数をシンボル周波数の整数倍にし
ておけば、最初のシンボル判定点とサンプル点とのずれ
を決定すれば、その後はそのサンプルに対し、４ｋごと
のサンプルを順次取出せばよい。このシンボル判定点に
近い実部Ｉ（ｋ）のサンプル系列Ｉ1 （４ｋ）と拡散符
号生成部１４からの拡散符号ＩR （４ｋ），ＱR （４
ｋ）との相関計算がそれぞれ相関計算部３８，３９で行
われる。これら相関計算結果は乗算部４１，４２でそれ
ぞれ自乗され、これら自乗演算の結果が加算部４３で加
算される。Ｉ1 （４ｋ）とＩR （４ｋ），ＱR （４ｋ）
との相関値Ａ′（４ｋ）が得られる。このＡ′（４ｋ）
が最大（ピーク）となるように拡散符号生成部１４での
拡散符号の生成位相がシフトされる。A sample near the chip decision point estimated by the estimation unit 31 is extracted by the thinning unit 37 from the real part (in-phase component) I 1 (k) in the complex baseband signal sequence. Each sample of the series of the real part I 1 (k) is shown in FIG.
If it is a point position and the estimated symbol determination point is a ∘ point position, one sample near the ∘ point position in the sample at the ∘ point position is taken out for each ∘ point as shown in FIG. 1C. If the sampling frequency is set to an integral multiple of the symbol frequency, the deviation between the first symbol determination point and the sampling point can be determined, and thereafter, every 4k samples can be sequentially taken out from the sample. Sample sequence I 1 (4k) and the spread code from the spread code generating section 14 I R of the real part I (k) closest to the symbol decision point (4k), Q R (4
The correlation calculation with k) is performed by the correlation calculation units 38 and 39, respectively. These correlation calculation results are squared by the multiplication units 41 and 42, and the results of these square calculations are added by the addition unit 43. I 1 (4k) and I R (4k), Q R (4k)
A correlation value A '(4k) with is obtained. This A '(4k)
The spreading code generation phase in the spreading code generation unit 14 is shifted so that the value becomes maximum (peak).
【００１３】前記パイロット信号では同相側のＰＮ符号
と、直交側のＰＮ符号とで相関値が一様に小さいため、
複素ベースバンド信号の位相が正しくない場合でも同相
側のＰＮ符号との同期をとるだけでも正しい同期が得ら
れる。このような関係にない場合、つまり例えばＱＰＳ
Ｋ変調信号を拡散符号で拡散したスペクトラム拡散信号
との同期をとる場合はＩ1 （４ｋ），Ｑ1 （４ｋ）とＩ
R （４ｋ），ＱR （４ｋ）との複素相関を求めればよ
い。この場合も相関計算はシンボル周期（前記例では４
ｋ）ずつずらしながら行うため、サンプル周期ずつずら
しながら行う場合より相関計算の計算量が少なくて済
む。上述においてはＱＰＳＫ変調の例をしたが、その他
の直交変調の場合にもこの発明は適用できる。また上述
において直交変換部１３を含むこれより以後の処理はソ
フトウェア処理によってもよい。In the pilot signal, since the correlation value between the in-phase PN code and the orthogonal PN code is uniformly small,
Even if the phase of the complex baseband signal is not correct, correct synchronization can be obtained only by synchronizing with the PN code on the in-phase side. When there is no such relationship, that is, for example, QPS
When synchronizing with a spread spectrum signal obtained by spreading a K modulation signal with a spread code, I 1 (4k), Q 1 (4k) and I 1 (4k)
It suffices to find the complex correlation with R (4k) and QR (4k). Also in this case, the correlation calculation is performed in the symbol period (4 in the above example).
k) Since the calculation is performed while shifting by each step, the calculation amount of the correlation calculation can be smaller than that when performing by shifting by each sampling period. Although the example of QPSK modulation has been described above, the present invention can be applied to other orthogonal modulation. Further, in the above description, the processing thereafter including the orthogonal transformation unit 13 may be software processing.
【００１４】[0014]
【発明の効果】以上述べたようにこの発明によれば、シ
ンボル判定点を推定し、これに近いサンプルについての
み相関計算を行うため、各サンプルについて相関計算を
行う場合より演算量が少なくて済む。特に前記パイロッ
ト信号の場合は入力信号の複素ベースバンド信号の実部
についてのみ、シンボル判定点付近のサンプルとの相関
計算を行うようにしたため、計算量が従来より著しく少
なくなる。As described above, according to the present invention, the symbol decision point is estimated, and the correlation calculation is performed only for the samples close to the symbol decision point. Therefore, the amount of calculation is smaller than that in the case of performing the correlation calculation for each sample. . In particular, in the case of the pilot signal, since the correlation calculation with the sample in the vicinity of the symbol decision point is performed only for the real part of the complex baseband signal of the input signal, the amount of calculation becomes significantly smaller than in the conventional case.
【図１】Ａはこの発明の実施例の機能構成を示すブロッ
ク図、Ｂは複素ベースバンド信号の実部の系列と、推定
シンボル判定点の例を示す図、Ｃは取出されたシンボル
判定点近くのサンプル系列の例を示す図である。FIG. 1 is a block diagram showing a functional configuration of an embodiment of the present invention, B is a diagram showing an example of a real part sequence of a complex baseband signal and an estimated symbol decision point, and C is a symbol decision point taken out. It is a figure which shows the example of a nearby sample series.
【図２】従来のスペクトル拡散信号の同期装置を示すブ
ロック図。FIG. 2 is a block diagram showing a conventional spread spectrum signal synchronizer.
Claims (3)
スバンド信号からシンボル判定点を推定する手段と、 上記推定されたシンボル判定点に近い上記入力複素ベー
スバンド信号のサンプル値を取出す手段と、 その取出されたサンプル値と拡散符号との相関を計算す
る手段と、 その相関計算結果にもとづき上記拡散符号を上記入力複
素ベースバンド信号の拡散符号に同期させる手段と、 を具備するスペクトラム拡散信号同期装置。1. A means for estimating a symbol decision point from an oversampled input complex baseband signal, a means for extracting a sample value of the input complex baseband signal close to the estimated symbol decision point, and a means for extracting the sample value. A spread spectrum signal synchronizer comprising: means for calculating the correlation between the sample value and the spread code; and means for synchronizing the spread code with the spread code of the input complex baseband signal based on the correlation calculation result.
信号の実部に対して行う手段であることを特徴とする請
求項１記載のスペクトラム拡散信号同期装置。2. The spread spectrum signal synchronizer according to claim 1, wherein said extracting means is means for performing the real part of said complex baseband signal.
複素ベースバンド信号の振幅の２乗を求める手段と、そ
の２乗結果を離散的フーリエ変換する手段と、そのフー
リエ変換の結果中のチップ周波数成分の逆正接を求める
手段と、その逆正接がゼロとなる時点を上記チップ判定
点と推定する手段とよりなることを特徴とする請求項１
又は２記載のスペクトラム拡散信号同期装置。3. The chip decision point estimating means is means for obtaining the square of the amplitude of the input complex baseband signal, means for discrete Fourier transforming the squared result, and chips in the result of the Fourier transform. 2. A means for obtaining an arctangent of a frequency component, and means for estimating a time point at which the arctangent becomes zero as the chip determination point.
Alternatively, the spread spectrum signal synchronizer according to item 2.
Priority Applications (3)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP8116169A JPH09307479A (en) | 1996-05-10 | 1996-05-10 | Spread spectrum signal synchronizing device |
| US08/847,597 US5799038A (en) | 1996-04-30 | 1997-04-25 | Method for measuring modulation parameters of digital quadrature-modulated signal |
| EP97107203A EP0805573A3 (en) | 1996-04-30 | 1997-04-30 | Method for measuring modulation parameters of digital quadrature-modulated signal |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP8116169A JPH09307479A (en) | 1996-05-10 | 1996-05-10 | Spread spectrum signal synchronizing device |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| JPH09307479A true JPH09307479A (en) | 1997-11-28 |
Family
ID=14680507
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| JP8116169A Withdrawn JPH09307479A (en) | 1996-04-30 | 1996-05-10 | Spread spectrum signal synchronizing device |
Country Status (1)
| Country | Link |
|---|---|
| JP (1) | JPH09307479A (en) |
Cited By (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JP2002314458A (en) * | 2001-04-11 | 2002-10-25 | Denso Corp | Demodulator for wireless communication unit adopting cdma system |
-
1996
- 1996-05-10 JP JP8116169A patent/JPH09307479A/en not_active Withdrawn
Cited By (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JP2002314458A (en) * | 2001-04-11 | 2002-10-25 | Denso Corp | Demodulator for wireless communication unit adopting cdma system |
| JP4572482B2 (en) * | 2001-04-11 | 2010-11-04 | 株式会社デンソー | CDMA wireless communication demodulator |
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| Date | Code | Title | Description |
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| A300 | Withdrawal of application because of no request for examination |
Free format text: JAPANESE INTERMEDIATE CODE: A300 Effective date: 20030805 |