JPH08335828A - Voltage control oscillator and integrated bias circuit - Google Patents

Abstract

PURPOSE: To provide a compact voltage control oscillator and a compact integrated bias circuit which have the reduced phase noise changes caused by the oscillation frequency and also can be tuned to the wide bands. CONSTITUTION: A tuning circuit consists of a serial resonance circuit including of a varactor diode 2 and a serial inductor 3 and an impedance transformer 4 which inverts the impedance with desired frequency. A negative resistance amplifier circuit consists of a field effect transistor 1 and a feedback inductor 5 and then partly feeds the oscillation power that is supplied to a load resistor 7 to the input side via a capacitor 6. Then the transformer 4 having transformation ratio (n) applies the impedance inversion to the Junction capacity Cj and the equivalent serial resistance Rs of the diode 2 to transform them into a parallel circuit of inductance Cj /n<2> and conductance n<2> Rs . The transformer 4 also applies the impedance inversion to the serial inductance Ls and transforms it into the parallel capacitance n<2> Ls .

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a voltage controlled oscillator (VCO) applied to a microwave / millimeter wave band and an integrated bias circuit applied to a microwave circuit in general.

[0002]

2. Description of the Related Art For example, a document (Kimishima)
M. et al. : "A Semi-Monolithi
c Wideband VCO with Outpu
t Power Control Cobilit
y Using an Active Power S
"plitter", IEEE MTT-S Diges
t, HH-6, pp. 1317-1320, 1992).
As shown in FIG. 22A, in the microwave band VCO of the conventional example shown in FIG. 22, the first tuning circuit is composed of a parallel resonant circuit including a varactor diode 2a and a parallel inductor 3a.
The second tuning circuit is composed of a parallel resonant circuit including a varactor diode 2b and a parallel inductor 3b. The amplifier circuit is composed of the field effect transistor 1 and outputs the oscillation power to the load resistor 7. Further, for example, Japanese Patent Laid-Open No. 4-3120
The microwave band VCO of the conventional example shown in Japanese Patent No. 04 is shown in FIG.
(A) As in the tuning circuit, an open end having an electrical length of approximate 180 degrees by a transmission line 4a and the desired oscillation frequency characteristic admittance Y _t with an electrical length of approximate 90 degrees varactor diode 2 and the desired oscillation frequency It is composed of the line 20. The amplifier circuit includes a field effect transistor 1 and a series feedback inductor 5, and outputs oscillation power to the load resistor 7. Further, generally, in the microwave band VCO of the conventional example, when the varactor diode 2 is mounted on the substrate 122 as shown in FIG. 24A, a fixed stray capacitance parallel to the varactor diode 2 is provided between the mounting pattern 127 on the substrate and the ground conductor. Figure 24 shows Ca
It will be added as shown in (b). Further, when the varactor diode 2 is tightly coupled to the field effect transistor 1 to form a wide band oscillator, the microstrip line 1
Bias circuits 124 for the field effect transistor 1 and the varactor diode 2 are used as the low impedance line 23.
And 125 are high impedance lines with little high frequency leakage.

The conventional microwave band VCO employs a system in which the difficulty of wideband oscillation and the amount of change in phase noise due to the oscillation frequency are in a trade-off relationship, and the wideband characteristic and the noise characteristic are traded off.

In the above conventional example, as shown in FIG. 22 (a), the admittances Y _s5 and Y _g3 that allow the first and second parallel resonant circuits to be seen from the source and gate terminals of the field effect transistor 1 are:
It looks like this: Y _s5 = 1 / jωL _p1 + 1 / (R _s + 1 / JωC _j ) = 1 / jωL _p1 + {jωC _j + (ωC _j ) ² R _s } / {1+ (ωC _j R _s ) ² } ≈1 / jωL _{p1 + jωC j + (ωC j} ) 2 R s (∵1» (ωC j R s) 2) Y g3 ≒ 1 / jωL P2 + jωC j + (ωC j) 2 R s wherein the C _j and R _S is a varactor diode Junction capacitance of 2a and 2b and equivalent series resistance, L _P1 and L _P2 are parallel inductors 3
It represents the inductance of a and 3b. Therefore, when changing the oscillation frequency by changing C _j according to the applied voltage so that one of Y _S5 and Y _g3 becomes capacitive and the other becomes inductive,
Since the conductance component (ωC _j ) ² R _{s of} Y _S5 and Y _g3 changes significantly, the Q of the resonance circuit changes greatly.
When Q is large as in (b), the phase noise is low,
When Q is small, the phase noise is high and the change in the phase noise due to the oscillation frequency is large. Further, as shown in FIG. 23B, the junction capacitance (C _j ) and the equivalent series resistance (R _s ) of the varactor diode 2 are impedance-inverted by the action of the transmission line 4a, and the inductance (L _v = − (Y _t ) ²
/ Ω ² C _j ) and conductance (G _v = R _s Y _t ² ), while the open-ended line 20 has an equivalent inductance (L
_r ) and capacitance (C _r ) and conductance (G
_Since it is represented by the parallel circuit of _r ), the admittance Y _s6 looking into the tuning circuit side from the source terminal of the field effect transistor 1 as shown in FIG. 23A is as follows. Y _s6 = (Y _t ) ² · (R _s + 1 / jωC _j ) + (G _r + jωC _r + 1 / jωL _r ) = (Y _t ) ² R _s + G _r + j {(ωC _r −1 / ωL _r ) − (Y _t ) ² / ωC _j } Therefore, | ωC _r −1 / ωL _r | ≧ | (Y _t ) ² / ωC _j
|, And the oscillation frequency band is almost determined by L _r and C _r , and the conductance component (Y _t ) ² R _s + of Y _s6
Since G _r does not depend on C _j , Q of the resonance circuit is constant, and the change in phase noise due to the oscillation frequency is small. If formed as shown in FIG. 24A, the varactor diode 2 is formed.
The junction capacitance change ratio n _a (when C _a is present) is as follows. n _a = (C _jmax + C _a ) / (C _jmin + C _a ) = (n _i · C _jmin + C _a ) / (C _jmin + C _a ) = {n _i · (C _jmin + C _a ) − (n _i −1) _{) -C a} / (C jmin} + C a) = n i - {(n i -1) · C a} / (C jmin + C a) <n i C a = ε 0 ε r S / d C here maximum and minimum values of _jmax and C _jmin is C _j, (in the absence of C _a) n _i is the junction capacitance change ratio of the varactor diode 2, epsilon ₀ and epsilon _r is the dielectric constant of vacuum and the substrate 122, S is attached pattern The area of 127, d represents the thickness of the substrate 122. Thus C _a is larger the n _a is reduced, the band of the oscillation frequency can not be obtained a desired oscillation frequency band becomes narrower.

[0005]

The conventional voltage controlled oscillator as described above adopts a method of trade-off between the wide band characteristic and the noise characteristic. Therefore, a wide band oscillator that is easy to oscillate has many phase noise changes due to the oscillation frequency. Broadband oscillation is difficult for those with little change in phase noise due to oscillation frequency. In addition, since the entire circuit pattern including the bias circuit is formed on the same type of board, the stray capacitance between the mounting pattern on the board and the ground conductor increases, and the change ratio of the junction capacitance decreases and the desired oscillation frequency band is obtained. However, there is a problem that the freedom of design is restricted by the range of desired line impedance that can be realized, especially when applied to a microwave bias circuit that requires a band exceeding an octave, characteristic deterioration occurs at both ends of the band. there were.

A problem to be solved by the present invention is to realize a desired miniaturization in an integrated bias circuit which is generally applied to a microwave circuit, because a voltage controlled oscillator has little change in phase noise due to an oscillation frequency and can be tuned in a wide band. (Miniaturized wideband oscillation system).

[0007]

The voltage controlled oscillator of the present invention comprises a semiconductor oscillator and a variable capacitance element, and an integrated bias circuit is generally applied to microwave circuits. It is characterized in that a means is provided and a miniaturized low noise wideband oscillation system is adopted.

The tuning circuit is composed of a series resonance circuit composed of a variable capacitance element and an inductance and an impedance transformer for performing impedance inversion at a desired oscillation frequency.
Alternatively, a resistor is additionally connected to the tuning circuit in series to form the tuning circuit.

The impedance transformer is formed by a transmission line having an electrical length of about 90 degrees at a desired oscillation frequency. Alternatively, a lumped constant circuit including an inductor and a parallel capacitor or a lumped constant circuit that changes the impedance transformation ratio by using a variable capacitance element instead of the parallel capacitor is used.
Alternatively, it is formed of a quarter-wavelength coupled line or a quarter-wavelength coupled line on a superconducting material that easily satisfies the oscillation condition.
Alternatively, first and second switching devices are provided at both ends of a plurality of impedance transformers having different center frequencies, and one of the impedance transformers is selected and formed according to a desired oscillation frequency.

The series resonant circuit is formed by connecting an even number of series variable circuits in the directions of polarities different from each other by half. Alternatively, a resistance sufficiently larger than the equivalent series resistance of the variable capacitance element is connected in parallel and formed.

The voltage controlled oscillator is formed by connecting first and second tuning circuits to two different oscillation element terminals.
Alternatively, the impedance transformer of each of the first and second tuning circuits is formed of a transmission line having an electrical length of approximately 45 to 90 degrees and approximately 90 to 135 degrees at a desired oscillation frequency. Alternatively, the resonant frequencies are formed differently by different DC voltages applied to the series resonant circuits of the first and second tuning circuits. Alternatively, instead of the feedback capacitor, a variable capacitance element for returning a part of the oscillation power to the input or separately coupling the output of the oscillation power to the load is provided, and the capacitance is controlled according to the oscillation frequency.

The voltage controlled oscillator has a low impedance with a first substrate forming a bias circuit or the like of a high impedance line for supplying a DC voltage to the oscillation element and the variable capacitance element or a matching circuit of the oscillation element when forming an integrated circuit. The dielectric constant or the substrate thickness of the second substrate forming the transformer or the first substrate forming the matching circuit of the oscillating element and the second substrate forming the mounting pattern of the variable capacitance element are different.

When forming an integrated circuit, the integrated bias circuit has a different dielectric constant or substrate thickness between the first substrate forming a matching circuit and the like and the second substrate forming a bias circuit of a high-impedance line such as a spiral inductor. . Alternatively, a second substrate on which a spiral inductor and other bias circuits for high impedance lines are formed is mounted on the first substrate on which a matching circuit for oscillating elements having different dielectric constants or substrate thicknesses is formed. Or, by attaching a wire in the air directly to the matching circuit of the oscillation element formed on the substrate mounted on the base plate or via a relay capacitor mounted on the base plate so as to pass through the through hole provided on the substrate. An inductor is formed as a bias circuit for connecting and supplying power. Alternatively, the substrate and the relay capacitor are mounted on the lower and upper steps of the step provided on the base plate. Alternatively, a matching circuit of the oscillating element and a relay pattern are formed on the lower and upper steps of the step provided on the substrate mounted on the base plate, and the wire is stretched in the air through the relay pattern to the matching circuit. Alternatively, via the relay capacitor mounted on the same substrate forming the matching circuit. Alternatively, a wire is formed vertically above the microstrip line formed on the substrate in the air to form an inductor as a bias circuit for feeding power from above.

[0014]

The voltage controlled oscillator of the present invention comprises the transmission line having the electrical length of about 90 degrees at the desired oscillation frequency, the lumped constant circuit, the quarter wavelength coupling line and other impedance transformers and the series resonance circuit. An equivalent parallel resonance circuit is formed by the tuning circuit consisting of the variable capacitance element and the variable capacitance junction capacitance is changed to enable tuning in a wide band, and by keeping Q of the resonance circuit constant, a change in phase noise due to an oscillation frequency is suppressed. Reduce. Also, by changing the permittivity or substrate thickness of the circuit pattern forming substrate when forming an integrated circuit, or by forming a desired bias circuit with a wire that extends in the air, it is possible to reduce the capacitance to ground and improve the wideband characteristics, and to reduce the mounting area. To reduce.

[0015]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS A voltage controlled oscillator (VCO) according to an embodiment of the present invention has a tuning circuit as shown in FIG. 1A, a tuning circuit, a series resonance circuit composed of a varactor diode 2 and a series inductor 3, and an impedance at a desired oscillation frequency. The impedance transformer 4 has an inverting function. The amplifier circuit includes a field effect transistor 1, a series inductor 5, and a load resistor 7.
And a capacitor 6 for returning a part of the oscillation power output to the input side to the input side. The capacitor 8 blocks the bias voltage for the varactor diode 2.

The VCO of the above embodiment adopts a system (broadband oscillation system) which realizes characteristics that the phase noise due to the oscillation frequency is low, changes little, and can be tuned in a wide band.

In the above embodiment, as shown in FIG. 1A, the admittance Y _s looking into the tuning circuit side from the source terminal of the field effect transistor 1 is the impedance transformer 4 (impedance transformation ratio) as shown in FIG. 1B. n), the junction capacitance (C _j ) of the varactor diode 2 and the equivalent series resistance (R
_s ) impedance reversal inductance (C _j / n
² ) and conductance (n ² R _s ) and impedance reversal capacitance (n ² R _s ) of the series inductor (L _s ).
Since L _s ) is converted in parallel, Y _s = n ² · (R _s + 1 / jωC _j + jωL _s ) = n ² R _s + jn ² (ωL _s −1 / ωC _j ) Therefore, the resonance angular frequency ω _r = 1 / (L _s C _j ) ^1/2 Therefore, when the applied voltage of the varactor diode 2 is changed and C _j is changed, ω _r can be tuned in a wide band, and the conductance component n ² R _s of Y _s becomes constant regardless of C _j. Phase noise changes little with frequency.

[0018] Note that as shown in FIG. 1 impedance transformer 4 in the above embodiment of (a) FIG. 2 (a), formed by the transmission lines 4a characteristics admittance Y _t with an electrical length of approximate 90 degrees at the desired oscillation frequency You may. The transmission line 4a and the series inductor 3 are realized by the microstrip line 102 and the wire 103 formed on the substrate 101, as shown in FIG. In the above embodiment, as shown in FIG. 2A, the admittance Y _s1 looking into the tuning circuit side from the source terminal of the field effect transistor 1 is as follows. ^{_{Y s1 = (Y t) 2}} · (R s + 1 / jωC j + jωL s) = (Y t) 2 R s + j (Y t) 2 (ωL S -1 / ωC j) Accordingly resonance angular frequency omega _r = 1 / (L _s C _j ) ^1/2 , and if C _j is changed in the same manner as above, ω _r can be tuned in a wide band and the conductance component (Y _t ) ² R of Y _s1 can be obtained.
_s is constant regardless of the oscillation frequency, and the change in phase noise due to the oscillation frequency is small.

Further, in the above embodiment of FIG. 1A, the impedance transformer 4 has a characteristic impedance Z _c having an impedance inverting function at a desired oscillation frequency as shown in FIG. 3A.
(When inductance L and parallel capacitance C (L
It may be formed by the lumped constant circuit 4b equivalent to the / 2C) ^1/2 ) transmission line. The circuit can be made smaller. The lumped constant circuit 4b is realized by the wire 103 and the chip capacitor 104 as shown in FIG.

In the above embodiment of FIG. 3A, the lumped constant circuit 4b is a lumped constant circuit 4c which forms a variable capacitance C _v with a variable capacitance element such as a varactor diode as shown in FIG. C _v may be changed and the impedance transformation ratio n may be changed. Since the degree of coupling between the tuning circuit side and the field effect transistor 1 can be changed, wideband oscillation with little change in phase noise due to the oscillation frequency can be performed.

In the above embodiment of FIG. 1A, the impedance transformer 4 has a quarter wavelength coupling line 4d as shown in FIG.
You may form with. Since it has a wide band property in the microwave band and it is not necessary to provide the blocking capacitor 8, the number of parts can be reduced.

Further, in the above embodiment of FIG. 5, the 1/4 wavelength coupling line 4d has a reflection coefficient Γ that allows the tuning circuit side and the amplification circuit side to be seen from the assumed node between the field effect transistor 1 and the 1/4 wavelength coupling line 4d. _{For t} and Γ _a , the oscillation condition (| Γ _t ·
Γ _a | ≧ 1, ∠ (Γ _t · Γ _a ) = 2nπ, n = 0, ±
1, ± 2, ...) may be easily realized with a superconducting material. Loss due to multiple reflection can be reduced and stable oscillation can be achieved.

In the above embodiment of FIG. 1A, the series resonance circuit has varactor diodes 2-1 and 2-2 and series inductor 3-in which the polarities are opposite to each other and are connected in series as shown in FIG. 6A. You may comprise 1 and 3-2 and 3-3 in combination. When the varactor diodes 2-1 and 2-2 are used as resonant elements, the phase noise generated by up-conversion of the baseband noise due to the nonlinearity can be greatly reduced. The varactor diodes 2-1 and 2-2 have a pattern 1 formed on the substrate 101 as shown in FIG.
It is attached to 06 and is connected to the control voltage supply line 107. As shown in FIG. 6C, the temporal changes in the junction capacitance due to the oscillating waves are made opposite to each other to cancel.

Further, in the above embodiment of FIG. 1A, the series resonance circuit includes a resistor 9 (having a resistance R ₁ sufficiently larger than the equivalent series resistance R _s of the varactor diode 2) as shown in FIG. 7A. You may insert in parallel. The tuning circuit is shown in Fig. 8.
The resistor 10 may be inserted in series as in (a).
Unnecessary oscillations can be made less likely to occur. The above embodiment is shown in FIG.
As shown in FIGS. 8A and 8A, when the desired frequency is oscillated, the varactor diode 2 and the series inductor 3 are close to the series resonance state. Therefore, the impedance Z ₁ for the series resonance circuit is equivalent to the equivalent series resistance R _s of the varactor diode 2. A short circuit occurs via (1 to 2 Ω) and Z ₁ ≈R _s . In FIG. 7A, the impedance Z ₁ ′ including the parallel resistor 19 and looking into the series resonance circuit is Z ₁ ′ = Z ₁ · R ₁ / (Z ₁ + R ₁ ) ≈
R _S · R ₁ / (R _s + R ₁ ) = R _s / {1+ (R _s / R
₁ )} ≈R _s (∵Z ₁ ≈R _S , R ₁ >> R _s ), the impedance change due to the addition of the parallel resistor 9 can be ignored, and the Q of the series resonance circuit does not decrease at the desired oscillation frequency.
On the other hand, when the impedances Z ₁ and Z ₁ ′ are displayed on the Smith chart as shown in FIG. 7B, the series resonance frequency is the point where the solid line and the broken line showing the locus with respect to the frequencies of Z ₁ and Z ₁ ′ intersect the real axis. In addition to the desired oscillation frequency, the reactance component of Z ₁ becomes a large value to some extent and Z ₁ ≈R _s does not hold, so that the circuit loss increases and unnecessary oscillation or the like hardly occurs. Further, in FIG. 8A, the impedance Z ₂ ′ including the series resistor 10 and looking into the tuning circuit is the characteristic impedance of the impedance transformer 4 as Z ₀ (number + Ω),
If the impedance considering the impedance transformer 4 and the series resonance circuit is Z ₂ , then Z ₂ ′ = Z ₂ + R ₂ ≈Z ₀ ² / Z ₁
+ R ₂ ≈ Z ₀ ² / R _s + R ₂ ≈Z ₀ ² / R _s (∵Z ₁ ≈R _s
, Z ₀ ² / R _s >> R ₂ ) and the change in impedance due to the addition of the series resistor 10 can be ignored, and the Q of the tuning circuit does not decrease at the desired oscillation frequency. On the other hand, when the impedances Z ₂ and Z ₂ ′ are displayed on the Smith chart as shown in FIG. 8B, the points where the solid line and the broken line showing the loci with respect to the frequencies of Z ₂ and Z ₂ ′ intersect the real axis are equivalent to parallel resonance. Frequency, and the reactance of Z ₁ becomes a large value to some extent other than the desired oscillation frequency, and Z ₁ ≈R _s does not hold.
_The effect of adding ₂ appears (R _s of Z ₀ ² / R _s has a large value), the circuit loss increases, and unnecessary oscillation is less likely to occur.

In the above embodiment of FIG. 2A, the impedance transformer 4 is switched by the diode switch 12 via the capacitors 8 and 11 for blocking the control voltage as shown in FIG. Two impedance transformers 4d-1 and 4d-2 having different center frequencies may be selected so as to select ones having a closer electrical length. Since the impedance inversion band is expanded, the change in Q of the resonant circuit at both ends of the band due to the limitation of the impedance inversion band due to the frequency characteristic of the impedance transformer 4 itself can be reduced,
A wider band oscillator can be realized.

Further, in the above embodiment of FIG.
(A), in respect to the source and the gate terminal of the field effect transistor 1, varactor diodes 2a and 2b, a series inductor 3a and 3b and characteristic admittance Y _t1 and Y
A tuning circuit composed of _t2 transmission lines 4a-1 and 4a-2 may be connected. A frequency band satisfying the oscillation condition can be widened to realize a wider-band oscillator. In the above-described embodiment, as shown in FIG. 10A, the admittance Y in which the tuning circuit side is seen from the source and base terminals of the field effect transistor 1.
_s2 and Y _g are as follows. Y _s2 = (Y _t1 ) ² · (R _s + 1 / jωC _j + jωL _s ) = (Y _t1 ) ² R _s + j (Y _t1 ) ² (ωL _S −1 / ωC _j ) Y _g = (Y _t2 ) ² _{· (R s + 1 / jωC} j + jωL s) = (Y t2) 2 R s + j (Y t1) 2 (ωL S -1 / ωC j) Accordingly resonance angular frequency _{ω r = 1 / (L s} C j) 1 Becomes ^{/ 2} , and if the values of the series inductors 3a and 3b are changed, or if the applied voltages of the varactor diodes 2-1 and 2-2 are changed and C _j1 and C _j2 are changed, ω _r can be tuned in a wide band, and Y _s2 and Y _The oscillation frequency can be changed within the range in which one of _g is capacitive and the other is inductive. As shown in FIG. 10B, for example, L _s1
When ≠ L _s2 , each resonance angular frequency ω _{r of the} two tuning circuits
(Y _s2 and Y _g of the imaginary part I _m (Y _s2) and I _m (Y _g) is 0
The angular frequency at which is always dependent on the tuning voltage at certain intervals (for example, the frequency range in which Y _s2 is inductive and Y _g is capacitive). Further, the conductance components (Y _t1 ) ² R _s and (Y _t2 ) ² R _s of Y _s2 and Y _g are constant regardless of the oscillation frequency, and the change in phase noise due to the oscillation frequency is small. To form the difference between Y _s2 and Y _g , one of the parameters differs from the other, eg L _s1 ≠ L _s2 , C
_It suffices if _j1 ≠ C _j2 or the like.

In the above embodiment of FIG. 10A, the transmission lines 4a-1 and 4a-2 have electrical lengths θ ₁ and θ ₂ (θ ₁ ≠ θ ₂ ) at the desired oscillation frequency as shown in FIG. the transmission line 4a-3 and 4a-4 characteristic admittance Y _t1 and Y _t2 with may be used. Even if the varactor diodes 2a and 2b and the series inductors 3a and 3b are the same, the transmission lines 4a-3 and 4a
Since the admittances Y _s3 and Y _g1 that can be expected via -4 can be made different, the advantage in the configuration is great.

In the above embodiment of FIG. 10A, the tuning circuits connected to the source and gate terminals of the field effect transistor 1 have varactor diodes 2a and 2b as shown in FIG.
And DC blocking capacitors 13a and 13b are inserted between the series inductors 3a and 3b, and a high-frequency blocking inductors 14a and 14b are provided for the control voltage supplied from the terminal 15. One is an adder 17 and a fixed offset voltage from the voltage source 16. And the other is the same as the varactor diode 2a.
And 2b may be applied. With the same control voltage, the junction capacitances C _j1 and C _j2 of the varactor diodes 2a and 2b are made different from each other,
Since the admittances Y _s4 and Y _g2 different from each other are obtained, the tuning circuits may have the same structure, which is a great structural advantage.

Further, in the above embodiment of FIG. 1A, the feedback capacitor 6 may return a part of the oscillation power output to the load resistor 7 to the input side instead of the varactor diode 18, as shown in FIG. . Alternatively, the oscillated power may be output to the load resistor 7 by a coupling circuit including a varactor diode 19, as shown in FIG. Since the junction capacitance of the varactor diodes 18 and 19 is changed so as to be large when the oscillation frequency is low and small when the oscillation frequency is high, the frequency characteristic of the input feedback amount with respect to the oscillation frequency can be corrected and is constant regardless of the oscillation frequency. The amount of input feedback is maintained and the oscillator can operate stably. Also, the frequency characteristic of the amount of coupling to the load due to the oscillation frequency can be corrected,
The output level of the oscillator can be kept constant regardless of the oscillation frequency.

In the voltage controlled oscillator of the above embodiment, FIG.
As described above, the matching circuit 109 of the oscillating element or the bias circuit 110 for supplying the DC voltage to the field effect transistor 1 and the varactor diode 2 is thinner than the first substrate 108 formed of a high impedance line, or has a larger dielectric constant. The impedance transformer 4 having a low impedance microstrip line and an electrical length of about 90 degrees at the desired oscillation frequency may be formed on the second substrate 111. Since the varactor diode 2 is tightly coupled to the field effect transistor 1 so that the oscillator can have a wide band and can be formed as a small impedance transformer having a large transformation, a wide band and compact oscillator can be realized.

Further, in the voltage controlled oscillator which is not limited to the above-mentioned embodiment, as shown in FIG. 16, it is formed on the second substrate 113 which is thicker or smaller in dielectric constant than the first substrate 108 which forms the matching circuit 109 of the oscillating element. The mounting pattern 114 of the varactor diode 2 may be formed. Since the stray capacitance C _a between the mounting pattern 114 and the ground conductor is reduced, it is possible to realize an oscillator in which the degree of decrease in the oscillation frequency band due to C _a is small. C _a = ε ₀ ε _r S / d where ε ₀ and ε _r are vacuum and the dielectric constant of the substrate 113, S is the area of the mounting pattern, and d is the thickness of the substrate 113.

Further, as shown in FIG. 17, a first substrate 108 for forming a matching circuit 109 for an oscillating element is an integrated bias circuit generally applied to a microwave circuit not limited to the above embodiment.
The high impedance line of the bias circuit 110 may be formed by the spiral inductor 115 on the second substrate 113 having a larger thickness or a smaller dielectric constant. Since the stray capacitance is reduced, the broadband property can be improved.

In the above-described embodiment, as shown in FIG. 18, the sub-board 113 mounted on the main board 108 for forming the matching circuit 109 of the oscillating element, etc. (need to have a larger thickness or a smaller dielectric constant than the main board 108) Bias circuit 11 on top
0 may be formed. Further, as shown in FIG. 19A, the matching circuit 109 of the oscillation element formed on the substrate 108 mounted on the base plate 117 is directly connected to the matching circuit 109 of the oscillation element via the relay capacitor 119 mounted on the base plate 117 by a wire 118 extending in the air. An inductor may be used as the bias circuit 110 that supplies power. Further, as shown in FIG. 19B, the wire 118 may be stretched in the air via the upper side of the drill hole 120 provided on the substrate 108 mounted on the base plate 117. In contrast to FIG. 20A showing the mounting cross section in the case of FIGS. 19A and 19B, the board 108 and the interruption are provided on the lower and upper steps of the step provided on the base plate 117 as shown in FIG. 20B. The capacitor 119 may be mounted. Further, as shown in FIG. 20C, the matching circuit 109 of the oscillating element and the relay pattern 121 are formed in the lower and upper steps of the step provided on the substrate 108 mounted on the base plate 117, and the matching circuit 109 is used for the relay. Pattern 12
The wire 118 may be stretched in the air through 1. Further, as shown in FIG. 20D, the relay capacitor 11 mounted on the same substrate 108 as the matching circuit 109 of the oscillation element formed on the substrate 108 mounted on the base plate 117.
The wire 118 may be stretched in the air through the cable 9. In either case, the capacitance to ground can be reduced, the wide band performance can be improved, and the mounting area can be further reduced.

In addition, as shown in FIG. 21 (a), an integrated bias circuit generally applied to microwaves not limited to the above embodiment,
Microstrip line 109 formed on substrate 108
An inductor may be used as a bias circuit 110 that feeds power from above by a wire 118 extending vertically from -1 in the air. Since the direction of the wire 118 and the direction of the high frequency current J on the microstrip line 109-1 and the magnetic field H are orthogonal to each other, the high frequency current J ′ on the wire 118 induced by the high frequency magnetic field H can be minimized. Further, if the wire 118 is connected at the center of the width direction of the microstrip line 109-1, the high frequency current density can be minimized as shown in FIG. 21B, so that the leakage of the high frequency current into the wire 118 can be minimized.

Further, although the case where the field effect transistor is applied to the oscillating element has been described in the above embodiment, it goes without saying that it may be a bipolar transistor or a diode having a negative resistance such as a Gunn diode or an impatt diode.

[0036]

As described above, the voltage controlled oscillator according to the present invention adopts a system in which the phase noise is little changed by the oscillation frequency and can be tuned in a wide band, and the integrated bias circuit generally applied to microwave circuits can be miniaturized to a desired size. , The difficulty of wideband oscillation and the amount of change in phase noise due to the oscillation frequency are in a tradeoff relationship as in the past, and there is a trade-off between wideband characteristics and noise characteristics and there is a restriction on realization. Compared with the method, it has an effect of realizing a stable and excellent wide band and small-sized voltage controlled oscillator and integrated bias circuit.

[Brief description of drawings]

FIG. 1 is a configuration diagram of a voltage-controlled oscillator according to an embodiment of the present invention and an equivalent admittance circuit diagram in which the tuning circuit side is considered.

FIG. 2 is a configuration diagram and an implementation diagram of another embodiment showing the present invention.

FIG. 3 is a configuration diagram and an implementation diagram of another embodiment showing the present invention.

FIG. 4 is a configuration diagram of another embodiment showing the present invention.

FIG. 5 is a configuration diagram of another embodiment showing the present invention.

FIG. 6 is a configuration diagram of another embodiment showing the present invention, a mounting diagram, and a diagram for explaining changes in junction capacitance with time.

FIG. 7 is a configuration diagram of another embodiment showing the present invention and a Smith chart of impedance looking into the series resonance circuit side.

FIG. 8 is a configuration diagram of another embodiment showing the present invention and a Smith chart of impedance looking into the tuning circuit side.

FIG. 9 is a block diagram of another embodiment showing the present invention.

FIG. 10 is a configuration diagram of another embodiment showing the present invention and a diagram for explaining changes in resonance angular frequency with respect to a tuning voltage.

FIG. 11 is a configuration diagram of another embodiment showing the present invention.

FIG. 12 is a configuration diagram of another embodiment showing the present invention.

FIG. 13 is a configuration diagram of another embodiment showing the present invention.

FIG. 14 is a configuration diagram of another embodiment showing the present invention.

FIG. 15 is a mounting view of another embodiment showing the present invention.

FIG. 16 is a mounting view of another embodiment showing the present invention.

FIG. 17 is a mounting view of another embodiment showing the present invention.

FIG. 18 is a mounting view of another embodiment showing the present invention.

FIG. 19 is a mounting view of another embodiment of the present invention.

FIG. 20 is a mounting cross-sectional view of another embodiment showing the present invention.

FIG. 21 is a mounting diagram of another embodiment showing the present invention and a diagram for explaining changes in high-frequency current density in the line width direction.

FIG. 22 is a diagram illustrating a configuration of a conventional voltage controlled oscillator and a diagram illustrating frequency characteristics of phase noise.

FIG. 23 is a block diagram of another conventional example and an admittance equivalent circuit that looks into the tuning circuit side.

FIG. 24 is a mounting diagram of another conventional example and a diagram illustrating stray capacitance.

[Explanation of symbols]

1 field effect transistor, 2, 2-1, 2-2, 2
a, 2b Varactor diodes 3, 3, 3-1, 3-2,
3-3, 3a, 3b Series inductor, 4 Impedance transformer, 4a, 4a-1, 4a-2 Transmission line, 4
b, 4c lumped constant circuit, 4d, 4d-1, 4d-2
1/4 wavelength coupled line, 5 series inductor, 6 feedback capacitor, 7 load resistor, 8 DC blocking capacitor, 9
Parallel resistor, 10 series resistor, 11 DC blocking capacitor, 12 Diode switch, 13a, 13b
DC blocking capacitor, 14a, 14b High frequency blocking inductor, 15 Control voltage supply terminal, 16 Offset voltage source, 17 Adder, 18 Varactor diode, 19
Varactor diode, 101 substrate, 102 microstrip line, 103 wire, 104 chip capacitor, 105 substrate, 106 mounting pattern, 107
Control voltage supply line, 108 first substrate, 109 matching circuit, 109-1 microstrip line, 110
Bias circuit, 111 second substrate, 112 wire,
113 second substrate, 114 mounting pattern, 115
Spiral inductor, 116 wires, 117 base plate, 118 wires, 119 relay capacitor, 120 drill holes, 121 relay pattern. In the drawings, the same reference numerals indicate the same or corresponding parts.

 ─────────────────────────────────────────────────── ─── Continuation of the front page (72) Inventor Akio Iida 5-1-1, Ofuna, Kamakura-shi Electronic Systems Research Center, Mitsubishi Electric Corporation

Claims (25)

[Claims] 1. A voltage controlled oscillator including a semiconductor oscillating element and a variable capacitance element, wherein a tuning circuit including a series resonance circuit including the variable capacitance element and an inductor and an impedance transformer for performing impedance inversion at a desired oscillation frequency is provided. A voltage controlled oscillator characterized in that 2. The voltage controlled oscillator according to claim 1, wherein the impedance transformer is formed by a transmission line having an electrical length of about 90 degrees at a desired oscillation frequency. 3. The voltage controlled oscillator according to claim 1, wherein the impedance transformer is formed by a lumped constant circuit including an inductor and a parallel capacitor. 4. The voltage controlled oscillator according to claim 3, wherein a variable capacitance element is used instead of the parallel capacitor in the lumped constant circuit to change the transformation ratio of the impedance transformer. 5. The voltage controlled oscillator according to claim 1, wherein the impedance transformer is formed of a quarter wavelength coupling line. 6. The voltage controlled oscillator according to claim 5, wherein the quarter-wave coupling line is formed of a superconducting material so as to easily satisfy the oscillation condition. 7. The serial resonance circuit is formed by connecting even numbers of series variable circuits in the directions of polarities different from each other to the variable capacitance element in series.
The voltage controlled oscillator according to 4, 5, or 6. 8. The series resonance circuit is formed by connecting in parallel a resistance sufficiently larger than the equivalent series resistance of the variable capacitance element. Voltage controlled oscillator. 9. The tuning circuit is further formed by additionally connecting resistors in series.
The voltage controlled oscillator according to 4, 5, 6, 7 or 8. 10. The first and second switching devices are provided at both ends of a plurality of impedance transformers having different center frequencies, and one of the impedance transformers is selected according to a desired oscillation frequency. 2, 3, 4, 5, 6,
7. The voltage controlled oscillator according to 7, 8, or 9. 11. The voltage controlled oscillator according to claim 1, wherein the first and second tuning circuits are connected to two different terminals of the oscillation element.
5. The voltage controlled oscillator according to 5, 6, 7, 8, 9 or 10. 12. The impedance transformer of each of the first and second tuning circuits is formed by a transmission line having an electrical length of approximately 45 to 90 degrees and approximately 90 to 135 degrees at a desired oscillation frequency. The voltage controlled oscillator according to claim 11. 13. The voltage controlled oscillator according to claim 11, wherein the resonant frequencies of the first and second tuning circuits are formed differently for different DC voltages applied to the series resonant circuit. . 14. A voltage-controlled oscillator, wherein a variable capacitance element for feeding back a part of oscillation power to an input is provided instead of the feedback capacitor, and the capacitance is controlled according to the oscillation frequency. 3, 4, 5, 6, 7, 8,
The voltage controlled oscillator according to 9, 10, 11, 12 or 13. 15. The voltage controlled oscillator according to claim 1, further comprising a variable capacitance element for separately coupling oscillation power to a load, and controlling the capacitance according to an oscillation frequency.
3, 4, 5, 6, 7, 8, 9, 10, 11, 12, 13
The voltage controlled oscillator according to 14 above. 16. A first circuit for forming a bias circuit or the like of a high impedance line for supplying a DC voltage to an oscillating element and a variable capacitance element or a matching circuit of an oscillating element when forming an integrated circuit
2. The dielectric constant or the thickness of the second substrate forming the low-impedance impedance transformer is different from that of the substrate of claim 2, 5, 6, 7, 8, 9, 10, 1.
The voltage controlled oscillator according to 1, 12, 13, 14 or 15. 17. A voltage controlled oscillator comprising a semiconductor oscillator and a variable capacitance element, wherein a first substrate for forming a matching circuit of the oscillation element and the like for forming an integrated circuit and a second pattern for forming a mounting pattern of the variable capacitance element and the like. A voltage-controlled oscillator characterized in that the dielectric constant or the thickness of the substrate is different. 18. An integrated bias circuit generally applied to a microwave circuit, comprising: a first substrate for forming a matching circuit and the like at the time of forming an integrated circuit; and a second substrate for forming a bias circuit for a spiral inductor and other high impedance lines. An integrated bias circuit having different dielectric constants or substrate thicknesses. 19. When forming an integrated circuit, a second substrate is formed on the first substrate.
The integrated bias circuit according to claim 18, wherein the substrate is mounted. 20. In an integrated bias circuit generally applied to a microwave circuit, a wire is extended in the air through a relay capacitor mounted on a base plate to a matching circuit formed on a substrate mounted on the base plate when the integrated circuit is formed. An integrated bias circuit characterized by forming an inductor that is directly connected to supply power. 21. The integrated bias circuit according to claim 20, wherein, when the integrated circuit is formed, the wire is stretched in the air so as to pass through the through hole provided on the substrate. 22. The integrated bias circuit according to claim 20, wherein the substrate and the relay capacitor are mounted on the lower and upper steps of the step provided on the base plate when the integrated circuit is formed. 23. When forming an integrated circuit, a matching circuit of an oscillating element and a relay pattern are formed on a lower stage and an upper stage of a step provided on the substrate to be mounted on a base plate, and the relay pattern is provided to the matching circuit. 21. The integrated bias circuit according to claim 20, wherein the wire is extended in the air. 24. The integrated bias circuit according to claim 20, wherein when the integrated circuit is formed, the relay capacitor mounted on the same substrate on which the matching circuit is formed is interposed. 25. An integrated bias circuit generally applied to microwave circuits, characterized in that an inductor is formed by vertically extending a wire in the air from a transmission line formed on a substrate at the time of forming the integrated circuit, and feeding the wire from above. Bias circuit.