JPH08308235A - Current resonance type switching power source circuit - Google Patents

Abstract

PURPOSE: To improve in the field of characteristics by expediting the decreases in the size and weight and the cost of a current resonance type switching power source circuit in which the power factor is improved. CONSTITUTION: The current resonance type switching power source circuit is so constituted as to reduce the second serial resonance current I01 in proportion to the rise of a rectified and smoothed voltage Ei by feeding back the switching output to the full-wave rectifying line of a power factor improving circuit (LN, CN, D2 , C2 ) side by the second serial resonance circuit (NR, C1 A) and increasing the inductance LR of the primary controlled winding NR of a magnetically coupled transformer MCT(A) by an MCT control circuit 1, thereby guaranteeing the regulating range at the time of an AC input voltage 100V.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a current resonance type switching power supply circuit whose power factor is improved,
In particular, the present invention relates to a current resonance type switching power supply circuit corresponding to a wide range AC input voltage of 100V to 200V.

[0002]

2. Description of the Related Art In recent years, by developing a switching element capable of withstanding a relatively large current and voltage of high frequency, most of the power supplies of a switching system are used as a power supply device for rectifying a commercial power supply to obtain a desired DC voltage. It is a device. The switching power supply is used as a power supply for various electronic devices as a high-power DC-DC converter, while increasing the switching frequency to downsize transformers and other devices.

By the way, in general, when a commercial power supply is rectified, the current flowing through the smoothing circuit has a distorted waveform, which causes a problem that the power factor indicating the utilization efficiency of the power supply is impaired. In addition, there is a need for measures to suppress harmonics generated by the distorted current waveform.

Also, as a power supply circuit, a so-called wide-range compatible power supply circuit is known which can be applied to a commercial AC power supply input of, for example, 100 V to 200 V.

Therefore, a switching power supply circuit as shown in the circuit diagram of FIG. 10 has been previously proposed by the applicant of the present invention as one of the switching power supply circuits having the above wide range and improved power factor. There is. This power supply circuit is a self-excited current resonance type converter using a half bridge, and AC input voltage V _AC = 80V to 288.
It corresponds to V and guarantees a load power of 120 W or more.

In the switching power supply circuit shown in this figure, the commercial AC power supply AC is full-wave rectified by a bridge rectification circuit D ₁ in which four rectification diodes are bridge-connected.

Then, for the full-wave rectification line between the positive output terminal of the bridge rectifier circuit D _{1 and} the positive electrode of the smoothing capacitor, a filter choke coil L _N , a fast recovery type diode D ₂ and a magnetic field which will be described later are provided. Coupling transformer M
The primary winding N _{P of} CT (D) is inserted in series. Also,
The filter capacitor C _N is the filter choke coil L
_It is inserted between the connection point of _N and the fast recovery type diode D ₂ and the positive electrode of the smoothing capacitor Ci, and the filter capacitor C _N and the filter choke coil L _N form a normal mode LC low pass filter.

This LC low pass filter is designed to prevent high frequency noise of the switching frequency from flowing into the AC line. Further, the fast recovery type diode D ₂ is provided in response to the high-frequency current of the switching cycle described later flowing through the full-wave rectification line.

Further, the parallel resonance capacitor C ₂ is connected in parallel with the primary winding N _P of the magnetic coupling transformer MCT (D) as shown in the figure, the primary winding of the magnetically-coupled transformer MCT (D) N _P Together, they form a parallel resonant circuit. The resonance frequency of the parallel resonance circuit is set to be substantially the same as the resonance frequency of the switching power supply, for example. The operation will be described later. The switching converter of this power supply circuit is provided with _two switching elements Q ₁ and Q _{2 which} are half-bridge coupled as shown in the figure, and each collector is connected between the connection point on the positive side of the smoothing capacitor Ci and the primary side ground. , Connected through the emitter.

[0010] The collectors of the switching elements Q _1, Q ₂ - is between the base, each starting resistor R _S, R _S is inserted, the resistor R _B, a switching element Q ₁ by R _B,
Adjust the base current (drive current) of Q ₂ . Also,
The bases of the switching elements Q _1, Q ₂ - respectively between the emitters damper diode D _D, D _D is inserted. The resonance capacitors C _B and C _B form a series resonance circuit for self-excited oscillation together with the drive windings N _B and N _{B of} the drive transformer PRT described next.

Drive transformer PRT (Power Regulati
ng Transformer) variably controls the switching frequency of the switching elements Q ₁ and Q ₂ , and in the case of this figure, the drive windings N _B and N _B and the resonance current detection winding N _D are wound. control winding N _C is the orthogonal saturable reactor is wound in a direction orthogonal to the windings. _One end of the drive winding N _B on the switching element Q ₁ side of the drive transformer PRT has a resonance capacitor C _B.
To the resistor R _B , and the other end is connected to the emitter of the switching element Q ₁ . Further, one end of the drive winding N _B on the switching element Q ₂ side is grounded to the ground and the other end is connected to the resonance capacitor C _B, and has a polarity opposite to that of the drive winding N _B on the switching element Q ₁ side. The voltage of is output.

Isolation transformer PIT (Power Isolation Tr)
The ansformer transmits the switching outputs of the switching elements Q ₁ and Q _{2 to} the secondary side. One end of the primary winding N ₁ of the insulating transformer PIT is connected to the emitter of the switching element Q ₁ and the switching element Q via the resonance current detection winding N _D.
The switching output is obtained by connecting to the contact of the collector of ₂ . Further, the other end of the primary winding N ₁ is connected to a contact of the high speed recovery type diode D ₂ and the primary winding N _P of the magnetic coupling transformer MCT (D) via a series resonance capacitor C ₁ for switching. The output is fed back to the full-wave rectification line. And
The inductance component of the insulating transformer PIT including the series resonance capacitor C ₁ and the primary winding N ₁ forms a resonance circuit for making the switching power supply circuit a current resonance type. In the case of this switching power supply circuit, on the secondary side of the insulation transformer PIT, the induced voltage induced in the secondary winding N ₂ by the primary winding N ₁ is converted into a DC voltage by the bridge rectifier circuit D ₃ and the smoothing capacitor C _3. Then, the DC output voltage E ₀ is obtained.

The control circuit 1 compares the DC voltage output E _O on the secondary side with a reference voltage and supplies a DC current corresponding to the error to the control winding N _C of the drive transformer PRT as a control current I _C. Error amplifier.

A magnetic coupling transformer MCT (D) is provided in this switching power supply circuit. The magnetic coupling transformer MCT (D) is configured to tightly couple the primary winding N _P and the secondary winding N _S inserted in the full-wave rectification line as shown in the figure. Then, for the secondary winding N _{S of the} magnetic coupling transformer MCT (D), a bridge rectifier circuit D ₄ for full-wave rectifying the output of the secondary winding N _S.
Further, a rectifying / smoothing circuit including a smoothing capacitor C ₄ for voltage superimposition (hereinafter referred to as a voltage superimposing capacitor) for smoothing the full-wave rectified output of the bridge rectifier circuit D ₄ to obtain a DC voltage V ₁₂ is provided. This voltage superimposing capacitor C ₄
Is provided such that the positive electrode terminal is connected to the collector of the switching element Q ₁ and the negative electrode terminal is connected to the positive electrode terminal of the smoothing capacitor Ci. That is, the voltage superimposing capacitor C ₄ is the smoothing capacitor Ci in the rectifying and smoothing line.
Is connected in series to the DC voltage V _{11 with} respect to the voltage V ₁₁ across the smoothing capacitor Ci, as will be described later.
It is supposed that ₁₂ is superimposed and the rectified smoothed voltage Ei supplied to the switching converter is boosted. That is, the rectified and smoothed voltage Ei is expressed as Ei = V ₁₁ + V ₁₂ , which allows the high-voltage operating power supply (rectified and smoothed voltage Ei) to drive the switching power supply. The operation will be described later.

In the switching operation of the switching power supply circuit having the above structure, when a commercial AC power supply is first turned on, a base current is supplied to the bases of the switching elements Q ₁ and Q ₂ via the starting resistors R _S and R _S , for example. However, if the switching element Q ₁ is turned on _first , the switching element Q ₂ is controlled to be turned off. Then, as the output of the switching element Q ₁ , a resonance current flows through the resonance current detection winding N _D to the primary winding N ₁ and the series resonance capacitor C ₁ , but in the vicinity where the resonance current becomes 0, the switching element Q ₂ Is turned on and the switching element Q ₁ is turned off. Then, a resonance current in the opposite direction to the above flows through the switching element Q ₂ . After that, the self-excited switching operation in which the switching elements Q ₁ and Q ₂ are alternately turned on is started. In this way, the switching elements Q ₁ and Q ₂ are alternately opened and closed by using the terminal voltage of the smoothing capacitor Ci as the operating power supply, and thus the primary side winding N _{1 of the} insulation transformer is
Is supplied with a drive current close to the resonance current waveform to obtain an alternating output to the secondary winding N ₂ .

Further, when the DC output voltage E _O on the secondary side decreases, the control circuit 1 controls the current flowing through the control winding N _C so that the switching frequency becomes low (close to the resonance frequency). Is controlled to increase the drive current flowing through the primary winding N ₁ to achieve a constant voltage (referred to as a switching frequency control method).

The power factor improving operation is as follows. In this circuit, the primary winding N ₁ of the isolation transformer PIT
The switching output to be supplied to is directly superposed on the rectified output voltage via the self-inductance of the primary winding N _P of the magnetic coupling transformer MCT (D). As a result, the smoothing capacitor Ci is charged in the state where the switching voltage is superimposed on the full-wave rectified voltage, and the terminal voltage of the smoothing capacitor Ci is lowered in the switching cycle by the superimposed amount of this switching voltage. Then
From the rectified voltage level of the bridge rectifier circuit, the capacitor Ci
The charging current comes to flow during the period when the terminal voltage of is decreased, and the average AC input current approaches the AC voltage waveform, so that the power factor is improved.

In the switching power supply circuit shown in this figure, the rectification smoothing voltage Ei is boosted by the configuration including the magnetic coupling transformer MCT (D) as described above. _AC = 80V
The level of the rectified and smoothed voltage Ei becomes high over the range of up to 290 V, and the rectified and smoothed voltage Ei is 1 even under the condition that the _AC input voltage V _AC is 100 V or less and a heavy load occurs.
00V or more is maintained. That is, the switching current increases under the condition of heavy load with a low AC input voltage such as 100 V AC or less, but the increased switching current excites the rainbow coil N _S of the magnetic coupling transformer MCT (D). Since the secondary side alternating voltage rises, the secondary side alternating voltage is rectified by the bridge rectifying circuit D ₄ to increase the terminal voltage of the voltage superimposing capacitor C ₄ , and as a result, the rectified and smoothed voltage Ei is maintained at 100 V or higher. The regulation characteristic of the secondary-side DC output voltage E _O is guaranteed for a wide-range _AC input voltage V _AC = 80 V to 288 V. And
Since the high voltage rectified and smoothed voltage Ei is obtained by using the magnetic coupling transformer to boost the voltage, it is not necessary to increase the breakdown voltage of the smoothing capacitor Ci. further,
In the switching power supply circuit of this configuration, the AC input voltage V _AC
It is also possible to obtain a high power factor over the range of = 80V to 290V.

The parallel resonance circuit formed by the inductance of the primary winding N _P of the magnetic coupling transformer MCT (D) and the resonance capacitor C ₂ suppresses the switching voltage fed back to the rectification smoothing line when the load becomes light. As a result, the rise of the rectified and smoothed voltage Ei which is the terminal voltage of the smoothing capacitor Ci at the time of light load is suppressed. That is, in the power supply circuit with this configuration, the switching frequency is controlled to increase as the power supply load decreases, but at this time, the switching voltage returned to the charging circuit side is suppressed by the resonance capacitor C ₂ and the terminal voltage rises. Block. Further, when the power supply load increases, the switching frequency lowers, approaches the resonance frequency of the resonance circuit of the primary winding N _{P of} the magnetic coupling transformer MCT (D) and the resonance capacitor C ₂ , and increases the switching voltage fed back. Acts like. Therefore, in this power supply circuit, the voltage fluctuation rate at which the terminal voltage Ei of the smoothing capacitor fluctuates due to the power supply load is reduced, and therefore the DC output voltage E _O can be easily made constant regardless of the load fluctuation.

For example, when it is attempted to obtain a power factor of 0.8 when the _AC input voltage V _AC = 230 V with the circuit configuration specifically shown in FIG. 10, under the condition of load power P _O = 150 W, the rectified smoothed voltage is obtained. The step-up bridge rectifier circuit D ₃ is formed by four Schottky diodes (3 A / 40 V), and the voltage superimposing capacitor C ₄ = 470 μF / 50 V,
Smoothing capacitor Ci = 470 μF / 400 V and resonance capacitor C ₂ = 0.056 μF. As an example of the structure of the magnetic coupling transformer MCT (D) that can be used in the circuit shown in FIG. 10, for example, as shown in FIG. 11, E-type cores C _R1 and C _R2 made of a pair of ferrite materials are used.
To form a gap G of about 1 mm at the joint of the central magnetic leg to form an EE type core, and a primary winding of 30T with a litz wire of 60 μφ × 80 bundles to the central magnetic leg. N _P and 15T secondary winding N _S
Can be respectively wound and configured.

The power supply circuit shown in FIG. 10 has a load power of 1
It was decided that the load power 120
The present applicant has previously proposed a switching power supply circuit that is compatible with a wide range under the condition of W or less and has a power factor improved. Then, FIG. 13 shows a configuration example of a switching power supply circuit based on this application. In this figure, the same parts as those in the circuit diagram shown in FIG. 10 are designated by the same reference numerals, and the description of the configuration of the switching converter, the constant voltage control system, and the like will be omitted.

In the power supply circuit of this figure, a filter choke coil L _N and a fast recovery type diode D ₂ are connected in series and inserted into the full-wave rectification line. Also,
A filter capacitor C _N is connected between the positive output terminal of the bridge rectifier circuit D ₁ and ground. That is, the bridge rectifier circuit D ₁ is inserted between the positive electrode and the negative electrode, and in this case as well, the filter choke coil L _N and the filter capacitor C _N are inserted.
To form a normal mode LC low-pass filter.

In this case, the resonance capacitor C _2A is connected in parallel with the fast recovery diode D ₂ as shown in the figure to form a parallel connection circuit. The capacitance of the resonance capacitor C _{2A in} this case is set so that C ₂ > C _{1 as} compared with the capacitance of the series resonance capacitor C ₁ , and is connected to the inductance of the filter choke coil L _N to form a resonance circuit. Is formed. The resonance frequency of this resonance circuit is set to be lower than the minimum switching frequency set for the switching elements Q ₁ and Q ₂ .

Further, in this case, the series resonance circuit composed of the primary winding N ₁ of the insulating transformer and the capacitor C _{1 is} composed of a filter choke coil L _N and a high speed recovery type diode D.
_The switching output is fed back to the full-wave rectification line by being connected to the connection point of the _second anode, and the series resonance current I ₀ can flow through the resonance capacitor C _2A . .

Further, according to this connection form, the switching voltage from the series resonance circuit is configured to be superimposed on the rectified output voltage via the filter choke coil L _N , whereby the power supply circuit of FIG. As in the case of, the switching voltage is superimposed on the rectification line and the smoothing capacitor Ci is charged, and as a result, the conduction angle of the AC input current is expanded and the power factor can be improved.

[0026]

From the viewpoint of the size and cost of equipment, the switching power supply circuit is reduced in size and weight by reducing the number of parts as much as possible and using small and inexpensive parts. It is preferable to reduce the cost and cost, and it is also preferable to improve the characteristics such as the ripple suppression of the DC output voltage. For example, in the switching power supply circuit shown in FIG. 10, the elements are selected as described above and the AC input voltage V _AC = 2.
When the power factor is configured to be improved to about 0.8 at 30 V, as shown in the relationship diagram of AC input voltage-power factor of FIG. 12, when the AC input voltage V _AC = 100 V, the power factor is almost constant. Although it is about 0.95, if the power factor is improved to this extent, the ripple voltage component of the power supply cycle of the DC output voltage E _O actually increases. Therefore, as means for suppressing the ripple voltage component, for example, it is conceivable to increase the capacitance of the smoothing capacitor C ₃ of the secondary side, an increase in the size of the smoothing capacitor C ₃ of the secondary side in this case In addition, the cost is increased. The same applies to the power supply circuit having the configuration shown in FIG.

[0027]

In view of the above-mentioned problems, the present invention is directed to a switching power supply circuit of a current resonance type, which is formed by a second series resonance capacitor and a resonance winding, and which is a switching converter of a switching converter. A resonance current control is provided in which a second series resonance circuit that supplies an output to a line between the rectifier circuit and the smoothing capacitor is provided, and the level of the resonance current flowing through the second series resonance circuit is changed according to the AC input voltage level. It was decided to provide a circuit.

[0028]

According to the above construction, in the current resonance type switching power supply circuit, the level of the switching current supplied from the second series resonance circuit to the rectification line is controlled so as to decrease in accordance with the rise of the AC input voltage. This makes it possible to keep the power factor constant regardless of the fluctuation of the AC input voltage.

[0029]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS FIG. 1 shows an embodiment of a switching power supply circuit according to the present invention, which corresponds to a wide range AC input voltage of 100V to 200V.
Further, since this power supply circuit is a self-exciting current resonance type converter by half-bridge coupling and adopts a switching frequency control system to make the DC output voltage a constant voltage, the same parts as those in FIG. The description of the switching operation and constant voltage control will be omitted.

In the circuit of this embodiment, the filter capacitor C _N is inserted between the positive electrode output terminal of the bridge rectifier circuit D _{1 and} the positive electrode of the smoothing capacitor Ci. The filter capacitor C _N may be inserted between the positive electrode output terminal of the bridge rectifier circuit D ₁ and the primary side ground, but the current flowing through the filter capacitor C _N has a high switching frequency. As described above, even though the smoothing capacitor Ci is used, the filter capacitor C _N can be regarded as being directly grounded to the ground. By connecting in this way, it is not necessary to select a withstand voltage product for the filter capacitor C _N. The cost can be reduced accordingly. The filter choke coil L _N is connected in series with the fast recovery diode D ₂ in the full-wave rectification line and inserted. That is, the filter choke coil L
_N is connected to the connection point of the positive output terminal of the bridge rectifier circuit D ₁ , the other end is connected to the anode side of the fast recovery diode D ₂ , and the cathode side is connected to the positive electrode of the smoothing capacitor Ci. ing. Further, the resonance capacitor C _2A in this case is connected in parallel to the fast recovery diode D ₂ as in the case of the circuit of FIG. 13 to form a parallel connection circuit.

Further, the series resonant circuit consisting of the primary winding N ₁ and the series resonant capacitor C ₁ of the insulating transformer PIT in this case, the primary winding N ₁ of the end series resonant capacitor C ₁
And the resonance current detection winding N _D of the drive transformer PRT, and are connected to the output points of the switching elements Q ₁ and Q ₂ , and the other end is grounded to the primary side ground. In the present specification, the series resonance circuit composed of the primary winding N ₁ and the series resonance capacitor C _{1 is referred} to as “first series resonance circuit”, and the primary controlled winding N of the magnetic coupling transformer MCT described later is described. It is to be distinguished from the second series resonance circuit formed by _R and the second series resonance capacitor C _1A .

Magnetic coupling transformer MCT in this embodiment
(A) is a saturable reactor in which a primary controlled winding N _R and a secondary controlled winding N _C1 are wound so as to be orthogonal to each other. As a concrete structure, for example, as shown in FIG.
A box-shaped core is formed by combining cores C _R1 and C _R2 of the same shape having four magnetic legs such that the ends of the magnetic legs face each other. Then, as shown in the figure, the primary controlled winding N _R and the secondary control winding N _C1 are wound around the box-shaped core so that the winding directions are orthogonal to each other and straddle two magnetic legs. Configured.

The primary controlled winding N _R of the magnetic coupling transformer MCT (A) has one end connected to the output points of the switching elements Q ₁ and Q ₂ as shown in FIG. The other end is connected to the anode of the fast recovery diode D ₂ via the second series resonance capacitor C _1A and the series connection. In this case, the inductance L _R of the primary controlled winding N _R and the capacitance of the second series resonant capacitor C _1A form the second series resonant circuit, so that the switching output of the switching converter is obtained. Is fed back to the full-wave rectification line via the second series resonance circuit.

Further, an MCT control circuit 1A is provided on the secondary control winding N _C1 of the magnetic coupling transformer MCT (A). In this case, for example, a transistor Q ₃ and resistors R ₁ , R ₂ , The secondary control winding N _C1 is composed of a class A amplifier circuit formed of R ₃ and has one end connected to the collector of the transistor Q ₃ and the other end grounded to the ground. As the configuration of the class A amplifier circuit in the MCT control circuit 1A, the base of the transistor Q ₃ is grounded via the resistor R ₂ and is connected to the positive terminal of the smoothing capacitor Ci via the resistor R ₁ . A base current having a level corresponding to the rectified and smoothed voltage Ei is supplied. The emitter is connected through a resistor R ₃ to a tertiary winding N ₃ formed by winding the primary winding N ₁ of the insulating transformer PIT, a rectifying / smoothing circuit including a rectifying diode D ₄ and a smoothing capacitor C ₄ , Operating power is supplied. The MCT control circuit 1
A is not particularly limited to the configuration of a class A amplifier circuit using a bipolar transistor as shown in this figure.

In the case of the switching power supply circuit having such a configuration, for example, the connection form of the power factor correction circuit portion formed by the filter choke coil L _N , the filter capacitor C _N , the high speed recovery type diode, and the resonance capacitor C _2A is as follows.
This is similar to the power factor correction circuit portion of the power supply circuit shown in FIG. 13 as the prior art, and in this embodiment, the switching output is fed back to the rectification line via the above-mentioned second series resonance circuit, so that FIG. The power factor is improved by the same action as in the case of the power supply circuit.

The operation of the magnetic coupling transformer MCT (A) in this embodiment is as follows. As described above, in the power supply circuit of this embodiment, the AC input voltage V _AC is 100
Although it is supposed to correspond to a wide range of V system to 200 V system, the rectified and smoothed voltage Ei also changes in accordance with the change of the _AC input voltage V _AC . Therefore, for example, the AC input voltage V _AC
Rectifying and smoothing voltage Ei also rises accordingly, but at this time, due to the operation of MCT control circuit 1A, control current I _C1 (which flows through secondary control winding N _C1 of magnetic coupling transformer MCT (A) ( The collector current of the transistor Q ₃ ) is made to increase. Due to this control current I _C1 , in the magnetic coupling transformer MCT (A), the value of the inductance L _R of the primary controlled winding N _R also increases as the _AC input voltage V _AC increases, as shown in FIG. 2A. To increase. And
The resonance condition of the second series resonance circuit changes accordingly, so that the second series resonance current I _O1 flowing through the second series resonance circuit.
Is decreased in proportion to the rise of the rectified and smoothed voltage Ei as shown in FIG.

Then, the second series resonant current I _O1 whose level changes according to the voltage value of the _AC input voltage V _AC input as described above is superimposed on the full-wave rectification line to improve the power factor. As a result, operation waveforms as shown in FIGS. 3 and 4 to be described later are obtained, and as a result, it becomes possible to obtain a substantially constant power factor for a wide range AC input voltage. For example, FIG. 2B is a diagram showing the measurement result of the relationship between the _AC input voltage V _AC and the power factor of the circuit of this embodiment, but as shown in this figure, the AC input voltage V _AC = 80 to 24
The power factor can be maintained at about 0.8 in the range near 0V.

Thus, the magnetic coupling transformer MCT (A) of this embodiment is the magnetic coupling transformer MC shown in FIG.
It is not provided for the purpose of boosting the rectified and smoothed voltage Ei like T (D), but is configured and provided so that the power factor can be constantly controlled with respect to a wide range AC input voltage.

Then, for example, the power factor is 0.8 as described above.
As a specific element selection in the present embodiment for keeping the degree constant, the second series resonance capacitor C _1A = 0.033 μF and the resonance capacitor C _2A = 0.30 when load power P _O = 150 W. 1 μF, smoothing capacitor Ci =
It is set to 470 μF / 400V. As the magnetic coupling transformer MCT (A), for example, in the structure shown in FIG. 7, the primary controlled winding is formed by a 60 μφ × 80 bundle of litz wires for a prismatic magnetic leg having a width of 6 mm × 6 mm. N _R is wound by 20 T, and the secondary control winding N _C1 is formed by winding a 60 μφ single wire by 1000 T.

Further, in the power supply circuit of this embodiment, the power factor can be kept constant at about 0.8 as described above, thereby increasing the electrostatic capacitance of the secondary side smoothing capacitor C _3. Without doing so, it is possible to suppress an increase in the ripple voltage component of the DC output voltage E _O when an AC input voltage of 100 V system is input. For example, the electrostatic capacity of the secondary side smoothing capacitor C ₃ used in this embodiment is 100 μF.
When the voltage is set to / 160V, a large-capacity secondary side smoothing capacitor C ₃ of 100VμF / 160V is required in order to obtain an equivalent ripple voltage level in the power supply circuit shown in FIG. 10 or 13.

Further, in the present embodiment, the lower limit regulation range when the AC input voltage of 100 V system is input is guaranteed even under the condition that the load power is 120 W or more. Therefore, the rectified smoothed voltage Ei is boosted as in the power supply circuit of FIG. It is not necessary to provide a circuit configuration for doing so. Therefore, the magnetic coupling transformer MCT (D) (shown in FIG. 11) provided for boosting shown in the circuit of FIG. 10, the bridge rectifier circuit D ₄ for boosting which is expensive due to the Schottky diode, and the large-capacity voltage superposition. The capacitor C ₄ is deleted, and instead, a magnetic coupling transformer MCT (A) for power factor control, a small-capacity second series resonance capacitor C _1A, and a simple and inexpensive MCT control circuit 1A (transistor Q). ₃ , a class A amplifier circuit composed of resistors R ₁ , R ₂ , and R ₃ , and a low-voltage rectifying / smoothing circuit for supplying operating power to the MCT control circuit 1A (for example, tertiary winding N ₃ = 3T, rectifying diode D ₄ , a smoothing capacitor C ₄ outputs a DC voltage of 12V). As a result, the circuit can be made smaller than the circuit shown in FIG. 10, and the cost can be suppressed accordingly.

Here, the operation waveforms of the respective parts of the switching power supply circuit described in FIG. 1 are shown by the waveform diagrams of FIGS. 3 and 4. Figure 3 shows the case when the AC input voltage V _AC = 100 V, and FIG. 4 shows the case when the AC input voltage V _AC = 230V. For example, when the AC input voltage V _AC = 100V is supplied as shown in FIG.
As described in (a), the second series resonance current I _O1 whose level is higher than that in the case where the AC input voltage of 200 V system is input flows according to the waveform shown in FIG. Supplied on line. The rectified output line voltage at this time (in this case, the potential between the connection point of the filter choke coil L _N and the fast recovery diode D ₂ and the ground) V ₁ is an alternating current as shown in FIG. The high-frequency component of the switching cycle is superimposed on the rectified and smoothed waveform of the level corresponding to the input voltage V _AC = 100V, and in this case, the waveform level is the actual rectified and smoothed voltage (voltage across the smoothing capacitor Ci) Ei. Equivalent to. The current I ₁ flowing through the high speed recovery type diode D _2, the high speed recovery type diode D ₂ is switched by the high frequency components of the rectified output line voltage V _1, the waveform shown in FIG. 2 (c), AC The high frequency current of the switching cycle flows during the period τ where the absolute value of the input voltage V _AC is higher than the voltage across the smoothing capacitor Ci. In addition, in the period other than the τ period, the second
The high frequency current (I _O1 ) supplied from the series resonance circuit is passed through the resonance capacitor C _{2A and} flows into the smoothing capacitor (not shown here). The average waveform of the _AC input current I _AC flowing through the commercial AC power supply AC is the waveform shown in FIG. 4 (e). Actually, for example, PF (power factor) = 0 according to the selection of the element described above. ．
The conduction angle is enlarged so that the power factor can be improved up to about 8.

Then, as shown in FIG. 4A, an AC input voltage V _AC = 230V, that is, a 200V commercial AC power supply A
When C is supplied, the second series resonance current I
_As described with reference to FIG. 2A, _O1 decreases so as to increase in proportion to the increase of the AC input voltage.
Since it is controlled by (A), the second series resonance current I _O1 (FIG. 4 (d)) at this time is the second series resonance current when the AC input voltage V _AC = 100V in FIG. 3 (d). I _O1
As can be seen by comparison with the above, the waveform has a level that is proportionally reduced of the AC input voltage. In addition, the rectified output line voltage V ₁ is the AC input voltage V _AC as shown in FIG.
= 230V, the waveform has an increased level compared to that of FIG. 3B, while the current I ₁ flowing through the fast recovery diode D _{2 corresponds} to that of the second series resonance current I _O1 as shown in FIG. The waveform has a reduced level compared to the case of c). The average waveform of the _AC input current I _AC is shown in FIG.
As shown in (e), in this case, the AC input current I _AC
The AC input voltage V _AC = 10 during the τ period corresponding to the conduction angle of
It is about the same as the AC input current I _AC (FIG. 3 (e)) in the case of 0V. That is, the power factor of about 0.8 is obtained as in the case of the AC input voltage V _AC = 100V.

Next, the circuit diagram of FIG. 5 shows the configuration of a switching power supply circuit which is another embodiment of the present invention. The same parts as those in FIG. The magnetic coupling transformer MCT (B) provided in this embodiment forms a so-called eye-shaped core by combining cores C _R11 and C _R12 made of a ferrite material having four magnetic legs as shown in FIG. Then, in this case, one of the four magnetic legs and one of the magnetic legs on the inner side adjacent to the outer magnetic leg are two primary magnetic controlled windings. The wire N _R is wound, and the secondary side control winding N _C1 is wound around the two magnetic legs on the center side. Further, although the MCT control circuit 1A in this figure is shown as a block, it may have a structure based on the class A amplifier circuit as shown in FIG. 1, or may be formed by another amplifier circuit. I do not care. Also in this embodiment, the magnetic coupling transformer MCT is controlled by the MCT control circuit 1A in response to the rise of the rectified smoothed voltage Ei.
(B) MCT control current I supplied to the secondary control winding N _C1
By changing _O1 , the inductance of the primary controlled winding N _R L _R
Are controlled to increase. As a result, in the same manner as in the previous embodiment, the full-wave rectification line flows through the second series resonance circuit (primary controlled winding N _R , second series resonance capacitor C _1A ) in proportion to the rise of the _AC input voltage V _AC. As a result of reducing the level of the second series resonance current fed back to the device, for example, as described with reference to FIG.
It is possible to keep the power factor constant with respect to the AC input voltage of the V system. Specifically, if the power factor is maintained at about 0.8, the power factor of the 100 V system is the same as in the previous embodiment. An increase in the ripple voltage component appearing in the DC output voltage E _O when the AC input voltage is input can be suppressed, and the regulation range at this time is also guaranteed.

Further, in this case, the switching elements Q ₁ , Q
₂ self-excited oscillation is to drive transformer CDT (Converte
r Drive Transformer) is not provided with the control winding N _C , and therefore the switching frequency is fixed.
And insulation transformer PRT (Power Regulating Trans)
former) is an orthogonal type saturable reactor in which the control winding N _C is provided orthogonally to the primary and secondary windings N ₁ and N ₂ , and the control circuit 1 is based on the DC output voltage E _O. Insulation transformer P by varying the control current I _C flowing through the control winding N _C
A so-called series resonance frequency control method is adopted in which the leakage magnetic flux of RT is controlled and the resonance current flowing in the series resonance circuit is changed to perform constant voltage control.

It is needless to say that the power factor control method including the magnetic coupling transformer MCT (B) shown in FIG. 1 can be applied to the series resonance frequency control method as in this embodiment. In addition, the switching frequency control system shown in FIG.
The power factor control method based on T (B) can naturally be applied.

Next, the circuit diagram of FIG. 6 shows the structure of a separately excited switching power supply circuit using a MOS-FET. The same parts as those in FIGS. The magnetic coupling transformer MCT (C) used in the embodiment shown in this figure has, for example, as shown in FIG. 9, a magnetic core of an EE type core formed by combining E type cores C _R21 and C _R22 made of a ferrite material. The next control winding N _C1 is wound. And
In this case, provided the primary controlled winding N _R so as to divide the primary controlled winding N _R1 and N _R2 (Note that the inductance L _R by combining the primary controlled winding N _R1 and N _R2 The primary controlled windings N _R1 and N _R2 are respectively wound around the magnetic legs outside the EE type core.

Also in this embodiment, the MCT control circuit 1A performs the secondary control of the magnetic coupling transformer MCT (C) so that the level of the second series resonance current I _O1 decreases in proportion to the increase of the AC input voltage. By varying the level of the MCT control current I _C1 supplied to the winding to control the inductance L _R of the primary controlled windings N _R1 and N _R2 , the 100 V system to 20
The power factor is kept constant over the AC input voltage of the 0V system, and the regulation range is also guaranteed.

Further, in the current resonance type converter in the embodiment of this figure, the switching elements Q ₁₁ and Q ₁₂ are, for example, M.
It is a separately excited type with half-bridge connection using an OS-FET. In this case, the control circuit 1 controls the oscillation drive circuit 2 based on the DC output voltage E _O , and changes the switching drive voltage supplied from the oscillation drive circuit 2 to the gates of the switching elements Q ₁₁ and Q ₁₂ ( For example, the constant voltage control is performed by performing the pulse width variable control of the drive voltage. The drain of the switching elements Q _11, Q ₁₂ - D _CL connected in the direction shown with respect to between the source, D _CL is the switching element Q _11, Q
It is a damper diode that forms a current path that is fed back when ₁₂ is turned off. The starting circuit 3 is provided to detect the voltage or current obtained in the rectifying and smoothing line at the time of starting the power supply and start the oscillation drive circuit 2. In this case, the tertiary winding N ₃ wound around the insulating transformer PIT, the rectifying diode D _4, and the smoothing capacitor C ₄
The low voltage DC voltage supplied by the MCT control circuit 1
It is connected so that it is supplied not only as A but also as the operating power supply of the starting circuit 3. Since the field effect type switching element used in this embodiment is driven by voltage and self-excited oscillation becomes difficult, it is preferable to provide the oscillation drive circuit 2 and the starting circuit 3 as shown in this figure.

Also for the separately excited current resonance type converter as in this embodiment, the magnetic coupling transformer MCT (A) or the magnetic coupling transformer MCT shown in the respective embodiments of FIGS. 1 and 5 is used. The power factor control method including (B) can be applied, and conversely, the magnetic coupling transformer M of the present embodiment.
It is also possible to apply the power factor control method by CT (C) to the configuration of the self-excited current resonance type converter as in the embodiments of FIGS. 1 and 5.

Further, the power factor improving method of the present invention which has been described so far in each of the above-described embodiments is the self-excited oscillation type / other-excited oscillation type as the current resonance type switching power supply circuit, the switching frequency control method (the drive transformer is Orthogonal P
Various methods such as RT (Power Regulating Transformer) / series resonance frequency control method (isolation transformer is orthogonal PRT), switching element half bridge coupling type / full bridge coupling type, and double voltage rectification method It is needless to say that the present invention can be applied to a power supply circuit configured by a combination pattern of types and is not limited to the combination pattern shown as an embodiment in each of the above figures.

[0052]

As described above, according to the present invention, in various types of current resonance type switching power supply circuits, the second series resonance circuit for returning the switching output to the full-wave rectification line is provided. Since the series resonance current of the second series resonance circuit is reduced in proportion to the increase of the AC input voltage by including
It is possible to keep the power factor constant over an AC input voltage of 100 V to 200 V, and the AC input voltage 100
The regulation range for the V system is also guaranteed. Further, in the present invention, for example, a circuit configuration for boosting the rectified and smoothed voltage is not required to guarantee the regulation range, and the ripple voltage component of the DC output voltage E _O that increases at a low AC input voltage is suppressed. In addition, since it is not necessary to increase the capacity of the smoothing capacitor on the secondary side, as a result, it is possible to reduce the size and cost of the power supply circuit.

[Brief description of drawings]

FIG. 1 is a circuit diagram of a switching power supply circuit as an embodiment of the present invention.

FIG. 2 is a diagram showing an inductance / second series resonance current characteristic of the primary controlled winding of the magnetic coupling transformer MCT with respect to an AC input voltage and a power factor characteristic in the example.

FIG. 3 is a waveform chart showing the operation of the switching power supply circuit in the example.

FIG. 4 is a waveform diagram showing the operation of the switching power supply circuit in the example.

FIG. 5 is a circuit diagram showing a switching power supply circuit as another embodiment.

FIG. 6 is a circuit diagram showing a switching power supply circuit as still another embodiment.

FIG. 7 is a perspective view showing a structure of a magnetic coupling transformer according to an embodiment.

FIG. 8 is a perspective view showing a structure of a magnetic coupling transformer according to an embodiment.

FIG. 9 is a perspective view showing a structure of a magnetic coupling transformer according to an embodiment.

FIG. 10 is a circuit diagram showing a switching power supply circuit as a prior art.

FIG. 11 is a perspective view showing the structure of a magnetic coupling transformer in the prior art.

FIG. 12 is a diagram showing a power factor characteristic with respect to an AC input voltage in the prior art.

FIG. 13 is a circuit diagram showing a switching power supply circuit as a prior art.

[Explanation of symbols]

1 Control circuit 1A MCT control circuit 2 Oscillation drive circuit 3 Starting circuit L _N Filter choke coil C _N filter capacitor D ₁ Bridge rectifier circuit D ₂ Fast recovery type diode C _2A Resonant capacitor MCT (A), MCT (B), MCT ( C) Magnetic coupling transformer N _R Primary controlled winding N _C1 Secondary control winding PIT (PRT) Isolation transformer CDT (PRT) Drive transformer Q ₁ , Q ₂ , Q ₁₁ , Q ₁₂ Switching element Q ₃ transistor Ci Smoothing capacitor C ₁ series resonance capacitor C _1A 2nd series resonance capacitor N ₁ primary winding

Claims (9)

[Claims] 1. A rectifying means for rectifying a commercial power source, a smoothing means for smoothing an output of the rectifying means, a switching means for connecting and disconnecting a voltage output from the smoothing means, a primary winding of an insulation transformer and a first transformer. First series resonance circuit formed by the series resonance capacitor of the above, to which the switching output of the switching means is supplied, a filter choke coil inserted in series in the line between the rectifying means and the smoothing means, and a high speed recovery type A rectifying element, a filter capacitor provided so as to form a low-pass filter together with the filter choke coil, a resonance capacitor connected in parallel to the high-speed recovery type rectifying element, a second series resonance capacitor, and a resonance capacitor. The switching output of the switching means is formed by a winding and is connected to the rectifying means and the flat output. A second series resonance circuit provided so as to be supplied to a line between the means; and a resonance current control means for varying the level of the resonance current flowing through the second series resonance circuit according to the AC input voltage level. A current resonance type switching power supply circuit comprising: 2. The resonance current control means supplies to the control winding a magnetic coupling transformer in which the resonance winding and the control winding are magnetically coupled, and a control current having a level corresponding to an AC input voltage level. Accordingly, the current resonance type switching power supply circuit according to claim 1, further comprising: a control circuit that variably controls the inductance of the resonance winding. 3. The magnetic coupling transformer is configured as an orthogonal saturable reactor in which the winding directions of the resonance winding and the control winding are orthogonal to each other. Item 2. A current resonance type switching power supply circuit according to item 2. 4. The magnetic coupling transformer according to claim 1, wherein either one of the resonance winding and the control winding is wound over two central magnetic legs of the eye-shaped core, and the other is wound with one outer magnet. The current resonance type switching power supply circuit according to claim 2, wherein the saturable reactor is wound around a leg and one central magnetic leg adjacent to the outer magnetic leg. 5. In the magnetic coupling transformer, the control winding is wound around a central magnetic leg of an EE type core, and the resonance winding is wound around two outer magnetic legs. The current resonance type switching power supply circuit according to claim 2, wherein the current resonance type switching power supply circuit is configured as a saturable reactor. 6. The current control circuit according to claim 2, wherein the control circuit is an amplification circuit provided so as to perform an amplification operation based on a rectified and smoothed voltage level. Resonant switching power supply circuit. 7. The constant voltage control is performed by changing the switching frequency of the switching means based on the DC output voltage obtained on the secondary side of the insulation transformer. The current resonance type switching power supply circuit according to any one of claims 1 to 6. 8. A constant voltage control is performed by varying a magnetic flux of the insulation transformer based on a DC output voltage obtained on the secondary side of the insulation transformer. 7. The current resonance type switching power supply circuit according to claim 6. 9. The switching means is a separately-excited current resonance type converter, and constant voltage control is performed by changing a switching drive signal based on a DC output voltage obtained on the secondary side of the insulation transformer. 7. The current resonance type switching power supply circuit according to claim 1, wherein the current resonance type switching power supply circuit is configured as follows.