JPH08308219A - Chopper type dc-dc converter - Google Patents

Abstract

PURPOSE: To decrease the duplicate part of the voltage waveform and the current waveform of a switching element and to reduce the loss at the time of the on and off operation of a switching element by zero-voltage or zero- current switching the element. CONSTITUTION: A main switching element 2 is switched to an off state in the state that the element 2 is turned on. The current flowing to the element 2 is immediately switched to the current flowing to a first resonance capacitor 8 and a second resonance capacitor 14, and the capacitor 8 is gradually discharged. simultaneously, the capacitor 14 is gradually charged, and the voltage across the element 2 is smoothly raised from OV. Thus, since the zero-voltage switching at the time of turning off the element 2 is performed, the switching loss at the time of turning off the element 2 can be reduced.

Description

Detailed Description of the Invention

[0001]

TECHNICAL FIELD The present invention relates to a chopper type DC-D.
The present invention relates to a C converter, and more particularly to a chopper type DC-DC converter having a small switching loss and high efficiency.

[0002]

2. Description of the Related Art A main switching element, a reactor and a main return rectification element are connected in a T-shape between a DC power supply and a load, an output capacitor is connected in parallel with the load, and a main return rectification terminal is connected to one end of the output capacitor. One end of the element is connected,
By controlling the main switching element on / off,
BACKGROUND ART A chopper type DC-DC converter configured to supply a DC output of a constant voltage different from the voltage of a DC power supply to a load has been widely used in power supply circuits of electronic devices and the like. For example, the conventional step-down chopper type DC-DC converter shown in FIG. 26 includes a DC power source 1, a main transistor 2 as a main switching element whose collector terminal (one main terminal) is connected to one end of the DC power source 1, A main freewheeling diode 3 as a main freewheeling rectifying element connected between the emitter terminal (the other main terminal) of the main transistor 2 and the other end of the DC power supply 1, and the main transistor 2 and the main freewheeling diode 3
Of the reactor 4 connected to the connection point of and the output capacitor 5 connected between the reactor 4 and the other end of the DC power supply 1.
A load 6 connected in parallel with the output capacitor 5; and a control circuit 7 for applying a control pulse signal to the base terminal of the main transistor 2 to turn on / off the main transistor 2. In this step-down chopper type DC-DC converter, the DC output of a voltage lower than the voltage of the DC power supply 1 is supplied to the load 6 by controlling the ON / OFF of the main transistor 2.

The conventional step-up chopper type DC-DC converter shown in FIG. 27 has a DC power supply 1 and a reactor 4 connected to the positive side line (one line) of the DC power supply 1.
And a collector terminal connected via a reactor 4 and an emitter terminal connected to the negative side line (the other line) of the DC power supply 1 as a main switching element, and to the collector terminal of the main transistor 2. Main freewheeling diode 3 as a main freewheeling rectifying element
, An output capacitor 5 connected between the main freewheeling diode 3 and the negative line of the DC power supply 1, a load 6 connected in parallel with the output capacitor 5, and a control pulse signal to the base terminal of the main transistor 2. Main transistor 2
And a control circuit 7 for controlling ON / OFF. In this step-up chopper type DC-DC converter, the DC output higher than the voltage of the DC power supply 1 is supplied to the load 6 by controlling the ON / OFF of the main transistor 2.

The control circuit 7 shown in FIGS. 26 and 27 changes the time width of the control pulse signal applied to the base terminal of the main transistor 2 in proportion to the fluctuation of the terminal voltage of the load 6 to change the main transistor 2 Control the on period of
The DC power supplied to the load 6 is stabilized.

[0005]

By the way, in the chopper type DC-DC converter shown in FIGS. 26 and 27, when the main transistor 2 is turned on or off, the collector-emitter voltage of the main transistor 2 as shown in FIG. There is a drawback that a large switching loss occurs due to the overlapping portion W of the waveform V _CE and the collector current waveform I _C of the main transistor 2. Further, since the collector-emitter voltage waveform V _CE and the collector current waveform I _C of the main transistor 2 rise steeply, there is a drawback that spike-like surge voltage V _sr , surge current I _sr and noise are generated.

Therefore, an object of the present invention is to provide a chopper type DC-DC converter capable of reducing switching loss, surge voltage and current.

[0007]

In the chopper type DC-DC converter of the invention according to claim 1, the main switching element, the reactor and the main return rectifying element are connected in a T shape between the DC power source and the load. The output capacitor is connected in parallel with the load, one end of the main return rectifying element is connected to one end of the output capacitor, and the voltage of the DC power supply is controlled by turning on / off the main switching element. DC outputs of different voltages are supplied to the load. In this chopper type DC-DC converter, a series circuit of an auxiliary switching element and a first resonance reactor is connected in parallel with the main switching element, and a first auxiliary return rectifying element is connected to a connection point of the series circuit. And the first
A first resonance capacitor is connected to a connection point between the resonance reactor and the main switching element, and a second resonance reactor is provided between the first resonance capacitor and the first auxiliary return rectifying element. And a rectifying element for resonance current are connected in series, and the first resonance capacitor and the second resonance capacitor are connected.
A second auxiliary return rectifier element is connected between a connection point of the resonance reactor and a connection point of the main return rectifier element and the output capacitor, and a rectifier element formed integrally with the main switching element or A circulating current rectifying element composed of an independent rectifying element is connected in parallel with the main switching element, a second resonance capacitor is connected in parallel with the circulating current rectifying element, and the first auxiliary return rectifying element is connected. And a third resonance capacitor is connected between a connection point of the rectifying element for resonance current and a connection point of the main switching element and the auxiliary switching element, before turning the main switching element from an off state to an on state. Then, the auxiliary switching element is turned from the off state to the on state, and the charging voltage of the first resonance capacitor during the on period of the auxiliary switching element is To turn off the auxiliary switching element since when the voltage of and the third resonance capacitor reaches a large value was discharged to 0V from the ON state.

Chopper type DC of the invention according to claim 2
In the DC converter, an energy feedback rectifying element is connected between a connection point of the first resonance reactor and the first auxiliary return rectifying element and a connection point of the main return rectifying element and the output capacitor. are doing. In the chopper type DC-DC converter of the invention according to claim 3, the connection point of the third resonance capacitor and the first auxiliary return rectifying element, and the connection of the main return rectifying element and the output capacitor. An energy feedback rectifying element is connected between the point and the point. Chopper type DC of the invention according to claim 4
In the DC converter, an energy feedback rectifying element is connected between a connection point of the second resonance reactor and the resonance current rectifying element and a connection point of the main return rectifying element and the output capacitor. Further, in the chopper type DC-DC converter of the invention according to "claim 5", a reverse charge preventing rectifying element is connected in parallel with the third resonance capacitor. Chopper type D of the invention according to claim 6
In the C-DC converter, a reverse charge preventing rectifying element is connected in parallel with the auxiliary switching element. Also,
In the chopper type DC-DC converter of the invention according to "claim 7", a discharge preventing rectifying element is connected in series with the auxiliary switching element and the first resonance reactor. Furthermore, the chopper type DC of the invention according to claim 8
In the DC converter, a third portion is provided between the connection point of the first resonance reactor and the first resonance capacitor and the connection point of the first auxiliary return rectifying element and the first resonance reactor. Of the auxiliary return rectifier element and the fourth resonance capacitor are connected in series, and the connection point of the third auxiliary return rectifier element and the fourth resonance capacitor and the main return rectifier element and the output capacitor The fourth auxiliary return rectifying element is connected to the connection point.

Chopper type DC of the invention according to claim 9
-In the DC converter, one main terminal of the main switching element is connected to one end of the DC power supply, and the main return rectifying element is between the other main terminal of the main switching element and the other end of the DC power supply. Connected between the load and the connection point of the main switching element and the main return rectifying element, the reactor is connected to the main switching element on / off control voltage than the DC power supply voltage. A low voltage DC output is provided to the load. Chopper type DC-DC of the invention according to claim 10
In the converter, the DC power supply includes an AC power supply and a rectifying circuit that converts an AC voltage of the AC power supply into a DC voltage. Further, in the chopper type DC-DC converter of the invention according to "claim 11", the reactor is connected to one line of the DC power supply, and one main terminal of the main switching element is connected through at least the reactor. The other main terminal of the main switching element is connected to the other line of the DC power supply, the main return rectifying element is connected between one main terminal of the main switching element and the load, By controlling ON / OFF of the main switching element, a DC output having a voltage higher than the voltage of the DC power supply is supplied to the load. In the chopper type DC-DC converter of the invention according to claim 12, the DC power supply includes an AC power supply and a rectifier circuit that converts an AC voltage of the AC power supply into a DC voltage. The reactor is connected to the input side or the DC output side.

Chopper type D of the invention according to claim 13
The C-DC converter has a main switching element, a reactor, and a main return rectifying element between the DC power supply and the load.
Connected in parallel with the load, an output capacitor is connected in parallel with the load, one end of the main reflux rectifying element is connected to one end of the output capacitor, and the main switching element is turned on / off to control the DC power supply. A DC output of a voltage different from the voltage of the above is supplied to the load. In this chopper type DC-DC converter, a first resonance capacitor is connected to a connection point between the main switching element and the main return rectifying element, and a connection point between the first resonance capacitor and the main switching element. A first auxiliary return rectifying element is connected, and a resonance reactor and a resonance current rectifying element are connected in series between the first resonance capacitor and the first auxiliary return rectifying element. A second auxiliary return rectifier element is connected between a connection point of the first resonance capacitor and the resonance reactor and a connection point of the main return rectifier element and the output capacitor, and the first auxiliary return rectifier element is connected. A second resonance capacitor is connected between a connection point of the rectifying element and the resonance current rectifying element and a connection point of the main switching element and the DC power supply, and the main switch is connected. When the switching element is turned off, the first resonance capacitor is discharged and the second resonance capacitor is gradually charged, and when the main switching element is turned on, the first resonance capacitor is gradually charged. While the second resonance capacitor is discharged, the first and second resonance capacitors resonate with the resonance reactor, and a resonance current flows through the main switching element.

Chopper type D of the invention according to claim 14
In the C-DC converter, a current limiting reactor is connected in series with the main return rectifying element or the main switching element. Further, in the chopper type DC-DC converter of the invention according to "claim 15", the connection point of the main return rectifying element and the first resonance capacitor, the first auxiliary return rectifying element, and the main switching. A third auxiliary return rectifying element and a third resonance capacitor are connected in series between the connection point of the element and the connection point of the third auxiliary return rectification element and the third resonance capacitor, and the connection point of the third resonance capacitor and the third resonance capacitor. A fourth auxiliary return rectifying element is connected between the main return rectifying element and the connection point of the output capacitor.

Chopper type D of the invention according to claim 16
In the C-DC converter, one main terminal of the main switching element is connected to one end of the DC power supply, and the main return rectifying element connects the other main terminal of the main switching element and the other end of the DC power supply. Connected between the reactor and the main switching element and the connection point of the main reflux rectifying element and the load, by controlling the main switching element on and off from the voltage of the DC power supply. A low voltage DC output is provided to the load. Chopper type DC-DC of the invention according to claim 17
In the converter, the DC power supply includes an AC power supply and a rectifier circuit that converts an AC voltage of the AC power supply into a DC voltage. Further, in the chopper type DC-DC converter of the invention according to "claim 18", the reactor is connected to one line of the DC power supply, and one main terminal of the main switching element is connected through at least the reactor. The other main terminal of the main switching element is connected to the other line of the DC power supply, the main return rectifying element is connected between one main terminal of the main switching element and the load, By controlling ON / OFF of the main switching element, a DC output having a voltage higher than the voltage of the DC power supply is supplied to the load. In the chopper type DC-DC converter of the invention according to "claim 19," the DC power supply is composed of an AC power supply and a rectifier circuit that converts an AC voltage of the AC power supply into a DC voltage.
The reactor is connected to the AC input side or the DC output side of the rectifier circuit. In the chopper type DC-DC converter of the invention according to claim 20, the DC power supply is an AC power supply, an AC-DC conversion switching element, and a rectifying element integrally formed with the AC-DC conversion switching element. Alternatively, an alternating current having a circulating current rectifying element composed of rectifying elements connected in parallel independently and converting the alternating voltage of the alternating current power source into a direct current voltage by controlling on / off of the alternating current to direct current switching element. A DC converter circuit, the reactor is connected to the AC input side of the AC-DC converter circuit, a power source resonance capacitor is connected between a pair of lines on the DC output side of the AC-DC converter circuit. There is.

When the main switching element is switched to the off state while the main switching element is on, the current flowing in the main switching element is immediately changed to the current flowing in the first and second resonance capacitors and the first resonance element is used. The capacitor gradually discharges. At the same time, the second resonance capacitor is gradually charged, and the voltage across the main switching element gradually rises from 0V. As a result, zero voltage switching (ZVS) is achieved when the main switching element is turned off, so that switching loss when the main switching element is turned off can be reduced. If the auxiliary switching element is switched from the off state to the on state before the main switching element is switched from the off state to the on state, the current of the first resonance reactor linearly increases from zero. This allows
Since zero current switching (ZCS) is achieved when the auxiliary switching element is turned on, switching loss when the auxiliary switching element is turned on can be reduced. The current of the main return rectifying element linearly decreases with the increase of the current of the first resonance reactor, and when the current of the main return rectifying element becomes 0, the main return rectifying element is cut off. At this time, the second resonance capacitor starts discharging, and a sinusoidal resonance current flows through the path of the second resonance capacitor, the auxiliary switching element, and the first resonance reactor. At the same time, the charged third resonance capacitor also starts discharging, and the third resonance capacitor, the auxiliary switching element, the first resonance reactor, the first resonance capacitor, and the second resonance capacitor. A sinusoidal resonance current flows in the path of the reactor and the resonance current rectifying element, and the first resonance capacitor is charged in a sinusoidal wave. Due to the discharge of the second resonance capacitor, the voltage of the second resonance capacitor, that is, the voltage of the main switching element drops to 0 V in a sine wave shape. When the current flowing through the first resonance reactor reaches a substantially maximum value and the voltage of the main switching element becomes 0V, the first
The voltage of the resonance reactor is also 0V, and the circulating current rectifying element becomes conductive. At this time, by turning the main switching element from the off state to the on state, zero voltage switching is achieved when the main switching element is turned on, so that the switching loss when the main switching element is turned on can be reduced. When the circulating current rectifying element is in the conductive state, the current flowing through the first resonance reactor continues to flow through the circulating current rectifying element and the auxiliary switching element. Since the third resonance capacitor continues to be discharged during this time, the first resonance capacitor is also continuously charged. Then, when the auxiliary switching element is switched from the ON state to the OFF state after the charging voltage of the first resonance capacitor reaches the maximum value and the voltage of the third resonance capacitor reaches 0 V, the current flows to the auxiliary switching element. The current that has been used is switched to the current flowing through the third resonance capacitor, and the third resonance capacitor is charged again from 0 V in a sine wave shape. Therefore, the voltage of the auxiliary switching element rises in a sinusoidal manner from 0 V, and when this voltage reaches the maximum value, the resonance current component of the current flowing through the first resonance reactor becomes zero. As a result, zero voltage switching is achieved when the auxiliary switching element is turned off, so that switching loss when the auxiliary switching element is turned off can be reduced.
As described above, it is possible to reduce the switching loss during the on / off operation of the main switching element and the auxiliary switching element. Further, spike-like surge voltage and current generated at turn-on and turn-off of the main switching element and the auxiliary switching element are absorbed by the resonance action of the resonance capacitor and the resonance reactor, and the voltage and current of the main switching element and the auxiliary switching element are absorbed. Since the rising and falling edges of the waveform are gentle, surge voltage and current can be reduced during the on / off operation of the main switching element and the auxiliary switching element. Further, since the auxiliary switching element is switched from the ON state to the OFF state after the charging voltage of the first resonance capacitor reaches the maximum value and is discharged until the voltage of the third resonance capacitor reaches 0 V, the auxiliary switching The time when the element is turned off can be clearly set.

Further, when an energy feedback rectifying element is additionally connected, the surplus energy of the first resonance reactor is fed back to the power source side (in the case of a step-down converter) or the load side through the energy feedback rectifying element ( In the case of the boost converter), the maximum value of the charging voltage of the third resonance capacitor can be prevented from becoming higher than the power supply voltage (in the case of the step-down converter) or the output voltage (in the case of the boost converter). Further, when the reverse charge prevention rectifying element is connected in parallel with the third resonance capacitor or the auxiliary switching element, the voltage of the third resonance capacitor is clamped at 0 V, so that the third resonance capacitor It is possible to prevent the third resonance capacitor from being charged with the reverse polarity due to the current flowing through the second resonance reactor when the voltage becomes 0V. Further, when the discharge prevention rectifying element is connected in series with the auxiliary switching element and the first resonance reactor, the parasitic capacitor of the auxiliary switching element is charged when the current of the first resonance reactor becomes zero. Energy can be prevented from being discharged through the main switching element and the first resonance reactor. In addition, a third auxiliary return rectifying element and a third auxiliary return rectifying element are provided between the connection point of the first resonance reactor and the first resonance capacitor and the connection point of the first auxiliary return rectifier element and the first resonance reactor. A fourth resonance capacitor is connected in series, and a fourth resonance capacitor is connected between a connection point of the third auxiliary return rectifying element and the fourth resonance capacitor and a connection point of the main return rectifying element and the output capacitor.
When the auxiliary reflux rectifying element of is connected, the charging time of the third resonance capacitor at the time of turning off the auxiliary switching element becomes long, so that zero voltage switching at the time of turning off the auxiliary switching element can be made more reliable and switching loss can be improved. Can be further reduced.

Further, in the system which does not use the auxiliary switching element, the first resonance capacitor is discharged and the second resonance capacitor is gradually charged when the main switching element is changed from the ON state to the OFF state. Go away. As a result, the voltage across the main switching element is 0
Since it gradually rises from V, zero voltage switching is achieved when the main switching element is turned off,
It is possible to reduce the switching loss when the main switching element is turned off. Further, when the main switching element changes from the off state to the on state, the second resonance capacitor is discharged, and the first and second resonance capacitors and the resonance reactor resonate and resonate with the main switching element. An electric current flows. As a result, the current of the main switching element increases from 0 in a sinusoidal manner, so that zero current switching is achieved when the main switching element is turned on, and the switching loss when the main switching element is turned on can be reduced. Therefore, it is possible to reduce the switching loss during the on / off operation of the main switching element with a simple circuit configuration, and to reduce the spike-shaped surge voltage and current due to the resonance action of the resonance capacitor and the resonance reactor. it can. Furthermore, when the current limiting reactor is connected in series with the main return rectifying element or the main switching element, the current of the main return rectifying element decreases linearly after the main switching element is turned on by the self-induction action of the current limiting reactor. Therefore, it is possible to avoid a short-circuit state of the power supply voltage (in the case of a step-down converter) or the output voltage (in the case of a step-up converter) due to the recovery recovery characteristic of the main free-wheeling rectifying element when the main switching element is turned on. The third auxiliary return rectifying element and the third auxiliary return rectifying element and the third switching capacitor are provided between the connection point of the main return rectifying element and the first resonance capacitor and the connection point of the first auxiliary return rectifying element and the main switching element.
Connected in series, and a fourth auxiliary return rectifier between the connection point of the third auxiliary return rectifier element and the third resonance capacitor and the connection point of the main return rectifier element and the output capacitor. When the element is connected, the charging time and the discharging time of the second resonance capacitor at the time of turning off and on of the main switching element become long, so that the zero voltage and the zero current switching at the time of turning off and turn on of the main switching element are more effective. It is possible to reliably reduce the switching loss.

[0016]

BEST MODE FOR CARRYING OUT THE INVENTION A chopper type D according to the present invention will now be described.
One embodiment of the C-DC converter will be described with reference to FIGS. However, in FIG. 1 and FIG.
The same parts as those shown in are denoted by the same reference numerals, and the description thereof will be omitted. FIG. 1 shows an embodiment in which the chopper type DC-DC converter according to the present invention is applied to a step-down converter. That is, the step-down chopper type DC-DC shown in FIG.
The converter includes a series circuit of an auxiliary transistor 9 as an auxiliary switching element connected in parallel with the main transistor 2 and a first resonance reactor 10 and a first auxiliary freewheeling diode connected to a connection point of the series circuit. (First auxiliary return rectifying element) 11, first resonance capacitor 10 connected to connection point of first resonance reactor 10 and main transistor 2, and first resonance capacitor 8
A second resonance reactor 15 and a resonance current diode (resonance current rectifying element) 16, which are connected in series between the first resonance diode 8 and the first auxiliary circulation diode 11, a first resonance capacitor 8 and a second resonance capacitor 15. Second auxiliary freewheeling diode (second auxiliary freewheeling rectifying element) 12 connected between the connection point of the resonance reactor 15 and the connection point of the main freewheeling diode 3 and the output capacitor 5, and the main transistor 2, a circulating current diode (circulating current rectifying element) 13 connected in parallel, a second resonance capacitor 14 connected in parallel with the circulating current diode 13, a first auxiliary return diode 11 and The third resonance capacitor 17 connected between the connection point of the resonance current diode 16 and the connection point of the main transistor 2 and the auxiliary transistor 9 is shown in FIG.
6 is connected to the circuit. Further, the control circuit 7 applies the auxiliary control pulse signal to the base terminal of the auxiliary transistor 9 before applying the main control pulse signal to the base terminal (control terminal) of the main transistor 2. In this embodiment, junction bipolar transistors are used as the main transistor 2 and the auxiliary transistor 9.

Although not shown in the drawing, in the control circuit 7,
An oscillation circuit unit that generates a triangular wave voltage of a constant cycle, an error amplification circuit unit that performs operational amplification of an error voltage of the terminal voltage of the load 6 with respect to a reference voltage, an error output voltage of the error amplification circuit unit, and a triangular wave voltage of the oscillation circuit unit A comparison circuit unit for comparison, a main control pulse generation circuit unit for generating a main control pulse signal having a time width proportional to the output voltage of the comparison circuit unit and applying it to the base terminal of the main transistor 2, and a main control pulse generation circuit unit. And an auxiliary control pulse generation circuit section for generating an auxiliary control pulse signal having a constant time width to be applied to the base terminal of the auxiliary transistor 9 before the main control pulse signal rises. The time width of the auxiliary control pulse signal generated from the auxiliary control pulse generation circuit section is extremely smaller than the off time of the main transistor 2.

In the above structure, when the main transistor 2 is in the ON state before t ₀ as shown in FIG. 2A, the main transistor 2 and the reactor 4 are connected to the load 6 as shown in FIG. 2H. A current I is flowing. At this time, as shown in FIGS. 2D and 2E, the first resonance capacitor 8 and the third resonance capacitor 17 are respectively arranged in FIG.
It is charged to the voltage E of the DC power supply 1 with the polarity shown in FIG.
As shown in FIG. 2A, at t ₀ , the main control pulse signal voltage V _B1 applied from the control circuit 7 to the base terminal of the main transistor 2 changes from high level to low level, and the main transistor 2 is turned on. When turned off, the current I _TR1 flowing through the main transistor 2 as shown in FIG.
That is, the current I of the load 6 immediately changes to the first resonance capacitor 8
And the current flowing through the second resonance capacitor 14 is switched. At this time, the first resonance capacitor 8 is gradually discharged, and the voltage V _C1 across the first resonance capacitor 8 is linear from the voltage E of the DC power supply 1 as shown in FIG. Go down to. Along with this, the second resonance capacitor 14 is gradually charged from 0V, and as shown in FIG. 2 (C), the voltage V across the second resonance capacitor 14 is increased.
_C2 , that is, the voltage V _TR1 across the main transistor 2 increases linearly from 0V. At the same time, as shown in FIG. 2 (F), the voltage V _TR2 across the auxiliary transistor 9 increases linearly from 0V. Therefore, when the main transistor 2 is turned off, zero voltage switching is performed in which the voltage waveform and the current waveform do not overlap each other.

As shown in FIGS. 2D and 2C, at t ₁ , the voltages V _C1 and V _C2 across the first and second resonance capacitors 8 and 14 are 0 V and the voltage of the DC power supply 1, respectively. E
Then, the main freewheeling diode 3 becomes conductive, and the current flowing through the first and second resonance capacitors 8 and 14 is the current I flowing through the main freewheeling diode 3 as shown in FIG. 2 (I). _Switch to _D. Main transistor 2 at this time
The voltages V _TR1 and V _TR2 across the auxiliary transistor 9 and the auxiliary transistor 9 are equal to the voltage E of the DC power source 1 as shown in FIGS. 2C and _2F , respectively. When the main transistor 2 is off, the current I of the load 6 flows from the main free wheeling diode 3 to the reactor 4.

As shown in FIG. 2B, at t ₂ , the auxiliary control pulse signal voltage V _B2 applied from the control circuit 7 to the base terminal of the auxiliary transistor 9 changes from low level to high level, and the auxiliary transistor 9 is turned on. When the state changes from the off state to the on state, the voltage V _TR2 across the auxiliary transistor 9 rapidly drops to 0 V as shown in FIG. While the main free-wheeling diode 3 is conducting, the voltage E of the DC power supply 1 is applied to the first resonance reactor 10, and FIG.
As shown in, the current I _L1 starts to flow in the first resonance reactor 10. This current I _L1 increases linearly until it equals the current I of the load 6. On the other hand, the main freewheeling diode 3
The current I _D flowing in the line decreases linearly as shown in FIG. Therefore, zero current switching is performed when the auxiliary transistor 9 is turned on.

As shown in FIG. 2G, at t ₃ , the first
When the current I _L1 of the resonance reactor 10 becomes equal to the current I of the load 6, the main freewheeling diode 3 is cut off, and no current flows through the main freewheeling diode 3 as shown in FIG. 2 (I). At this time, the energy of the second resonance capacitor 14 charged with the polarity shown in FIG.
Resonance capacitor 14 and first resonance reactor 1
0 resonates, and a resonance current flows through the path of the second resonance capacitor 14, the auxiliary transistor 9, and the first resonance reactor 10. At the same time, the energy of the third resonance capacitor 17 charged with the polarity shown in FIG. 1 is also released, and the third resonance capacitor 17, the first resonance reactor 10, the first resonance capacitor 8 and The second resonance reactor 15 resonates, and the third resonance capacitor 17,
Auxiliary switching element 9, first resonance reactor 1
A resonance current flows through the path of 0, the first resonance capacitor 8, the second resonance reactor 15, and the resonance current diode 16, and the first resonance capacitor 8 is charged in a cosine wave shape. Therefore, since a sinusoidal resonance current flows in the first resonance reactor 10 in a state of being superimposed on the current I of the load 6, the current I _L1 of the resonance reactor 10 continues to be as shown in FIG. It increases sinusoidally.
On the other hand, the voltages V _C2 and V _C3 across the second and third resonance capacitors 14 and 17 drop in a cosine wave shape as shown in FIGS. 2C and 2E, respectively.

As shown in FIG. 2G, at t ₄ , the first
When the current I _L1 of the resonance reactor 10 reaches the maximum value, that is, the sum of the current I of the load 6 and the maximum value I _p of the resonance current, the circulating current diode 13 becomes conductive. Along with this, the voltage V _C2 across the second resonance capacitor 14
Becomes 0V as shown in FIG. At this time, the control circuit 7 changes the main control pulse signal voltage V _B1 applied to the base terminal of the main transistor 2 from the low level to the high level as shown in FIG. To do. Since the voltage V _TR1 across the main transistor 2 at this time is 0 V as shown in FIG. 2C, zero voltage switching is performed when the main transistor 2 is turned on.

When the circulating current diode 13 becomes conductive at t ₄ , the resonance current component of the current I _L1 of the first resonance reactor 10 is equivalent to the circulation current diode 13, the auxiliary transistor 9, and the first resonance reactor 10. It becomes a circulating current and continues to flow. While the circulating current is flowing (t _{4 to} t ₅ ), energy is continuously emitted from the third resonance capacitor 17, so that the first resonance capacitor 8 is also continuously charged with the polarity shown in FIG. 1. Therefore, Fig. 2 (E)
And, as shown in (D), the voltage V _C3 across the third resonance capacitor 17 continues to drop and the voltage V _C1 across the first resonance capacitor 8 continues to rise. Then, at t ₅ , the voltage V of the first resonance capacitor 8
_C1 reaches the voltage E of the DC power supply 1 and the voltage V _{C3 of} the third resonance capacitor 17 becomes 0V.

After a short delay, the control circuit 7 changes the auxiliary control pulse signal voltage V _B2 applied to the base terminal of the auxiliary transistor 9 from high level to low level at t _{6 as} shown in FIG. 2B. The transistor 9 is turned off from the on state. At this time, the circulating current flowing through the auxiliary transistor 9 is switched to the current flowing through the third resonance capacitor 17, and the energy accumulated in the first resonance reactor 10 is released to cause the third resonance capacitor 17 to operate. It is charged again from 0V in a sine wave shape. Therefore, as shown in FIGS. 2F and 2G, the voltage V _TR2 across the auxiliary switching element 9 rises sinusoidally from 0 V and the current I _{L1 of} the first resonance reactor 10 is increased.
Decreases like a cosine wave. Therefore, when the auxiliary transistor 9 is turned off, the voltage V _TR2 across the auxiliary switching element 9 is 0 V, so that zero voltage switching is performed.

As shown in FIG. 2E, at t ₇ , the third
The voltage V _C3 across the resonance capacitor 17 is approximately maximum,
That is, when the voltage E of the DC power supply 1 is reached, the voltage V _TR2 across the auxiliary switching element 9 rapidly drops to 0 V as shown in FIG. Along with this, the resonance current component of the circulating current flowing in the first resonance reactor 10 becomes 0, and as shown in FIG.
The current I _L1 flowing through the load I becomes equal to the current I of the load 6. The remaining energy of the first resonance reactor 10 at this time is the second auxiliary freewheeling diode 12, the second resonance reactor 15, the resonance current diode 16, the first auxiliary freewheeling diode 11, and the first auxiliary freewheeling diode 11. It is fed back to the DC power supply 1 through the path of the resonance reactor 10 and the circulating current diode 13. As a result, the current I _L1 of the resonance reactor 10 decreases linearly and continuously as shown in FIG. 2 (G), and the main transistor 2 as shown in FIG. 2 (H).
The current I _{TR1 of} is increasing linearly from 0. And
At t ₈ , the current I _L1 of the first resonance reactor 10 becomes 0 as shown in FIG. 2 (G), and the current I _TR1 of the main transistor 2 becomes the current I of the load 6 as shown in FIG. 2 (H). Will be equal. Thus, after t ₈ a current I flows to the load 6 through the main transistor 2 and the reactor 4 from the DC power source 1.

As described above, in this embodiment, zero voltage or zero current switching is achieved when the main transistor 2 and the auxiliary transistor 9 are turned on and off, so that the main transistor 2 and the auxiliary transistor 9 are turned on and off. It is possible to reduce the power loss, that is, the switching loss. The spike-shaped surge voltage and surge current generated when the main transistor 2 and the auxiliary transistor 9 are turned on and off are generated by the first to third resonance capacitors 8, 14, 17 and the first and second resonance reactors. The rise and fall of the voltage and current waveforms of the main transistor 2 and the auxiliary transistor 9 are moderated by being absorbed by the resonance action of the transistors 10 and 15, so that surge voltage, surge current and noise during the on / off operation of the main transistor 2 are absorbed. Can be reduced. Further, the voltage V _{C1 of} the first resonance capacitor 8 is the DC power supply 1
Since the auxiliary transistor 9 is switched from the ON state to the OFF state after the voltage V _{C3 of} the third resonance capacitor 17 reaches 0V and the voltage V _{C3 of} the third resonance capacitor 17 becomes 0V, the time when the auxiliary transistor 9 is turned off is clarified. Can be set.

Next, an embodiment in which the chopper type DC-DC converter of the present invention is applied to a boost converter will be described with reference to FIGS. 3 and 4. However, in FIG. 3, the same parts as those in FIG. 1 are designated by the same reference numerals, and the description thereof will be omitted. Details of the inside of the control circuit 7 of FIG. 3 are the same as those of the control circuit 7 shown in the embodiment of FIG. The step-up chopper type DC-DC converter shown in FIG. 3 is obtained by connecting the circuit elements shown by reference numerals 9 to 17 to the circuit shown in FIG. 27, similarly to the circuit configuration shown in the embodiment shown in FIG.

In the configuration of FIG. 3, when the main transistor 2 is in the ON state before t ₀ as shown in FIG. 4A, the path of the reactor 4 and the main transistor 2 is as shown in FIG. 4H. The current I ₀ is flowing. At this time,
As shown in (D) and (E), the first resonance capacitor 8 and the third resonance capacitor 17 are charged to the terminal voltage of the load 6, that is, the output voltage E _{0 with the} polarities shown in FIG. 3, respectively. . As shown in FIG. 4A, at t ₀ , the main control pulse signal voltage V _B1 applied from the control circuit 7 to the base terminal of the main transistor 2 changes from high level to low level,
When the main transistor 2 is changed from the on state to the off state, the current I _TR1 flowing in the main transistor 2, that is, the current I _{0 of the} reactor 4 immediately changes to the first resonance capacitor 8 and the first resonance capacitor 8 as shown in FIG. The current is switched to the current flowing through the second resonance capacitor 14. At this time, the first resonance capacitor 8 gradually discharges, and as shown in FIG.
The voltage V _C1 across the resonance capacitor 8 is the output voltage E ₀
It goes straight down from. Along with this, the second resonance capacitor 14 is gradually charged from 0 V, and as shown in FIG. 4C, the voltage V _C2 across the second resonance capacitor 14, that is, the main transistor 2 Voltage on both ends V _TR1
Rises linearly from 0V. At the same time, the voltage V _TR2 across the auxiliary transistor 9 is 0 as shown in FIG.
It rises linearly from V. Therefore, the main transistor 2
At the time of turn-off, zero voltage switching is used in which the voltage waveform and the current waveform do not overlap.

As shown in FIGS. 4D and 4C, at t ₁ , the voltages V _C1 and V _C2 across the first and second resonance capacitors 8 and 14 become 0 V and the output voltage E ₀ , respectively. Then, the main freewheeling diode 3 becomes conductive, and the current flowing through the first and second resonance capacitors 8 and 14 is the current I _D flowing through the main freewheeling diode 3 as shown in FIG. 4 (I). Switch to. At this time, the voltages V _TR1 and V _TR2 across the main transistor 2 and the auxiliary transistor 9 are equal to the output voltage E ₀ as shown in FIGS. 4C and _4F , respectively.
Further, when the main transistor 2 is off, the current I _{0 of the} reactor 4 flows to the load 6 through the main free wheeling diode 3.

As shown in FIG. 4B, at t ₂ , the auxiliary control pulse signal voltage V _B2 applied from the control circuit 7 to the base terminal of the auxiliary transistor 9 changes from low level to high level, and the auxiliary transistor 9 is turned on. When the state changes from the off state to the on state, the voltage V _TR2 across the auxiliary transistor 9 rapidly drops to 0 V as shown in FIG. 4 (F). While the main free wheeling diode 3 is conducting, the output voltage E ₀ is applied to the first resonance reactor 10 and the current I _L1 is applied to the first resonance reactor 10 as shown in FIG. 4 (G). It begins to flow. This current I _L1 increases linearly until it equals the current I of the load 6. On the other hand, the current _ID flowing through the main freewheeling diode 3 decreases linearly as shown in FIG. Therefore, zero current switching is performed when the auxiliary transistor 9 is turned on.

As shown in FIG. 4G, at t ₃ , the first
When the current I _L1 of the resonance reactor 10 becomes equal to the current I _{0 of the} reactor 4, the main freewheeling diode 3 is cut off, and no current flows through the main freewheeling diode 3 as shown in FIG. 4 (I). At this time, the energy of the second resonance capacitor 14 charged with the polarity shown in FIG. 3 is released, the second resonance capacitor 14 and the first resonance reactor 10 resonate, and the second resonance capacitor 14 resonates. 14, the resonance current flows through the path of the first resonance reactor 10 and the auxiliary transistor 9. At the same time, the energy of the third resonance capacitor 17 charged with the polarity shown in FIG. 3 is also released, and the third resonance capacitor 17, the first resonance reactor 10, the first resonance capacitor 8 and The second resonance reactor 15 resonates, and the third resonance capacitor 17, the resonance current diode 16, the second resonance reactor 15, the first resonance capacitor 8, the first resonance reactor 10, and the auxiliary. A resonance current flows through the path of the switching element 9, and the first resonance capacitor 8 is charged in a cosine wave shape. Therefore, a sinusoidal resonance current flows through the first resonance reactor 10 while being superimposed on the current I _{0 of the} reactor 4, and thus the resonance reactor 10
Current I _L1 continues to increase sinusoidally as shown in FIG. On the other hand, the voltages V _C2 and V _C3 across the second and third resonance capacitors 14 and 17 are respectively shown in FIG.
And as shown in (E), it descends like a cosine wave.

As shown in FIG. 4G, at t ₄ , the first
When the current I _L1 of the resonance reactor 10 reaches the maximum value, that is, the sum of the current I _{0 of the} reactor 4 and the maximum value I _p of the resonance current, the circulating current diode 13 becomes conductive.
At the same time, the voltage V _C2 across the second resonance capacitor 14 becomes 0 V as shown in FIG. 4 (C). At this time, the control circuit 7 changes the main control pulse signal voltage V _B1 applied to the base terminal of the main transistor 2 from the low level to the high level as shown in FIG. To do. Since the voltage V _TR1 across the main transistor 2 at this time is 0 V as shown in FIG.
Zero voltage switching occurs when the main transistor 2 is turned on.

When the circulating current diode 13 becomes conductive at t ₄ , the resonant current component of the current I _L1 of the first resonance reactor 10 is equivalent to the circulation current diode 13, the first resonance reactor 10 and the auxiliary transistor 9. It becomes a circulating current and continues to flow. While the circulating current is flowing (t _{4 to} t ₅ ), energy is continuously emitted from the third resonance capacitor 17, so that the first resonance capacitor 8 is also continuously charged with the polarity shown in FIG. 3. Therefore, Fig. 4 (E)
And, as shown in (D), the voltage V _C3 across the third resonance capacitor 17 continues to drop and the voltage V _C1 across the first resonance capacitor 8 continues to rise. Then, at t ₅ , the voltage of the first resonance capacitor 8 reaches the output voltage E ₀ and the voltage of the third resonance capacitor 17 becomes 0V.

After a short delay, the control circuit 7 changes the auxiliary control pulse signal voltage V _B2 applied to the base terminal of the auxiliary transistor 9 from high level to low level at t _{6 as} shown in FIG. 4B. The transistor 9 is turned off from the on state. At this time, the circulating current flowing through the auxiliary transistor 9 is switched to the current flowing through the third resonance capacitor 17, and the energy accumulated in the first resonance reactor 10 is released to cause the third resonance capacitor 17 to operate. It is charged again from 0V in a sine wave shape. Therefore, as shown in FIGS. 4F and 4G, the voltage V _TR2 across the auxiliary switching element 9 rises sinusoidally from 0 V, and the current I _{L1 of} the first resonance reactor 10 is increased.
Decreases like a cosine wave. Therefore, when the auxiliary transistor 9 is turned off, the voltage V _TR2 across the auxiliary switching element 9 is 0 V, so that zero voltage switching is performed.

As shown in FIG. 4 (E), at t ₇ ,
The voltage V _C3 across the resonance capacitor 17 is approximately maximum,
That is, when the output voltage E ₀ is reached, the voltage V _TR2 across the auxiliary switching element 9 quickly becomes 0 as shown in FIG.
Descend to V. At the same time, the resonance current component of the circulating current flowing in the first resonance reactor 10 becomes 0, and the current I _L1 flowing in the first resonance reactor 10 becomes the current of the reactor 4 as shown in FIG. Will be equal to I ₀ . The remaining energy of the first resonance reactor 10 at this time is the circulating current diode 13, the first resonance reactor 10, the first auxiliary return diode 11, the resonance current diode 16, and the second resonance current diode. Reactor 15 and second
Is supplied to the load 6 through the path of the auxiliary free-wheeling diode 12 of. Accordingly, the current I _L1 of the resonance reactor 10
Is linearly and continuously decreased as shown in FIG. 4 (G), and the current I _TR1 of the main transistor 2 is linearly increased from 0 as shown in FIG. 4 (H). Then, at t ₈ , the current I _L1 of the first resonance reactor 10 is as shown in FIG.
It becomes 0 as shown in (G), and the current I of the main transistor 2
_TR1 becomes equal to the current I _{0 of the} reactor 4 as shown in FIG. Therefore, after t ₈ , the current I ₀ flows from the DC power supply 1 through the reactor 4 and the main transistor 2.

As described above, also in the embodiment shown in FIG. 3, the switching loss of the main transistor 2 and the auxiliary transistor 9 can be reduced and the surge voltage, surge current and noise can be reduced as in the embodiment shown in FIG. .

The circuits of the embodiments shown in FIGS. 1 and 3 can be modified. For example, in the circuits of the embodiments shown in FIGS. 5 and 6, the connection points of the first resonance reactor 10 and the first auxiliary freewheeling diode 11 of the circuits of FIGS. 1 and 3 and the main freewheeling diode 3 and An energy feedback diode 18 as a rectifying element for energy feedback is connected to the connection point of the output capacitor 5. The energy feedback diode 18 includes a third resonance capacitor 17 and a first auxiliary return diode 1 as shown by a broken line A.
1 and the connection point of the main freewheeling diode 3 and the output capacitor 5, or as shown by the broken line B, the connection point of the second resonance reactor 15 and the resonance current diode 16 and the main freewheeling diode 3 And the connection point of the output capacitor 6 may be connected. In the circuits shown in FIGS. 5 and 6, after the auxiliary transistor 9 is turned from the ON state to the OFF state (after time t _{6 in} FIGS. 2 and 4), the energy of the first resonance reactor 10 causes the third resonance Capacitor 1
When charging 7, the excess energy of the first resonance reactor 10 is fed back to the DC power supply 1 side through the energy feedback diode 18 (in the case of the circuit of FIG. 5) or supplied to the load 6 side (of the circuit of FIG. 6). If). Therefore, in the embodiment shown in FIGS. 5 and 6, the third resonance capacitor 17 is used.
It is possible to prevent the maximum value of the charging voltage V _C3 of 1 from becoming higher than the voltage E of the DC power supply 1 (in the case of the circuit of FIG. 5) or the output voltage E ₀ (in the case of the circuit of FIG. 6).

Further, the circuits of the embodiments shown in FIGS. 7 and 8 are arranged in parallel with the third resonance capacitor 17 of the circuits of FIGS. 19 is connected. The reverse charge prevention diode 19 may be connected in parallel with the auxiliary transistor 9 as shown by a broken line C. In the circuits shown in FIGS. 7 and 8, when the resonance current component of the current I _L1 of the first resonance reactor 10 is circulated through the circulation current diode 13 and the auxiliary transistor 9 (FIGS. 2 and 4).
Period from t ₄ to t ₅ ), the third resonance capacitor 17
Continues to be discharged, and the first resonance capacitor 8 is continuously charged. Then, when the charging voltage V _C3 of the third resonance capacitor 17 becomes 0V, the voltage V _C3 across the third resonance capacitor 17 is clamped at 0V. That is, when the charging voltage V _{C3 of} the third resonance capacitor 17 becomes 0 V, the second resonance reactor 15 is
When current is flowing in the second resonance reactor 15
Since all the current flowing through the reverse charging prevention diode 19 flows, the voltage V _C3 across the third resonance capacitor 17 is maintained at 0V. Therefore, FIG. 7 and FIG.
In the embodiment shown in (3), the third resonance capacitor 17 is charged with the opposite polarity by the current flowing through the second resonance reactor 15 when the charging voltage V _C3 of the third resonance capacitor 17 becomes 0V. Can be prevented.

The circuits of the embodiments shown in FIGS. 9 and 10 are connected to the auxiliary transistor 9 and the first resonance reactor 10 of the circuits of FIGS. 1 and 3, respectively, in series to prevent discharge as a rectifying element for discharge prevention. The diode 20 for connection is connected. In the circuits shown in FIGS. 9 and 10, when the auxiliary transistor 9 is changed from the ON state to the OFF state (time t _{6 in} FIGS. 2 and 4), the energy of the first resonance reactor 10 causes the third resonance capacitor At the same time that 17 is charged again from 0V, the parasitic capacitor of the auxiliary transistor 9 is also charged with the same polarity and voltage. The current I flowing through the first resonant reactor 10 after these charges are completed
_{When L1} becomes 0, discharge of the energy charged in the parasitic capacitor is blocked by the discharge prevention diode 20. Thereby, it is possible to prevent the energy charged in the parasitic capacitor of the auxiliary transistor 9 from being discharged through the main transistor 2 and the first resonance reactor 10 when the current of the first resonance reactor 10 becomes zero. . Therefore, in the embodiment shown in FIGS. 9 and 10,
It is possible to reduce the power loss of the auxiliary transistor 9 due to the discharge of the parasitic capacitor.

Further, the circuit of the embodiment shown in FIGS. 11 and 13 includes a connection point of the first resonance reactor 10 and the first resonance capacitor 8 and a first auxiliary circuit of the circuits of FIGS. 1 and 3, respectively. A third auxiliary return diode 22 and a fourth resonance capacitor 23 are connected in series between the return rectifying diode 11 and the connection point of the first resonance reactor 10, and a third auxiliary return diode 22 and The fourth auxiliary freewheeling diode 24 is connected between the connection point of the fourth resonance capacitor 23 and the connection point of the main freewheeling diode 3 and the output capacitor 5.

In the circuits shown in FIGS. 11 and 13, when the main transistor 2 is in the ON state before t ₀ as shown in FIGS. 12A and 14A, in the case of the circuit of FIG. As shown in (D), (E) and (G), the first and third
And the fourth resonance capacitors 8, 17, and 23 are charged to the voltage E of the DC power supply 1 with the polarities shown, respectively,
In the case of the circuit of FIG. 13, as shown in FIGS. 14D, 14E, and 14G, the first, third, and fourth resonance capacitors 8 and 1 are provided.
7 and 23 are charged to the terminal voltage of the load 6, that is, the output voltage E ₀ with the polarities shown in the drawing. As shown in FIGS. 12 (A) and 14 (A), at t ₀ , the main control pulse signal voltage V _B1 applied from the control circuit 7 to the base terminal of the main transistor 2 changes from high level to low level. When the transistor 2 changes from the on state to the off state, the current I _TR1 flowing in the main transistor 2, that is, the current I of the load 6
(In the case of the circuit of FIG. 11) or the current I of the reactor 4
₀ (in the case of the circuit of FIG. 13) is immediately the first, second and fourth
The current is switched to the resonance capacitors 8, 14, 23. At this time, the first and fourth resonance capacitors 8 and 23 gradually discharge, and in the case of the circuit of FIG. 11, as shown in FIGS. 12D and 12G, the first and fourth resonance capacitors The voltages V _C1 and V _C4 across the capacitors 8 and 23 linearly drop from the voltage E of the DC power supply 1, and in the case of the circuit of FIG. 13, as shown in FIGS. 14 (D) and 14 (G). The voltages V _C1 and V _C4 across the first and fourth resonance capacitors 8 and 23 linearly drop from the output voltage E ₀ . Along with this,
The second resonance capacitor 14 is gradually charged from 0V, and as shown in FIGS. 12C and 14C, the voltage V _C2 across the second resonance capacitor 14, that is, the main transistor 2 The voltage V _TR1 across the voltage rises linearly from 0V. At the same time, as shown in FIGS. _{12F and 14F,} the voltage V _TR2 across the auxiliary transistor 9 increases linearly from 0V. Therefore, when the main transistor 2 is turned off, zero voltage switching is performed in which the voltage waveform and the current waveform do not overlap each other.

12 (D), (G) and (C) or FIG.
As shown in (D), (G), and (C), at t ₁ , the voltage V _C1 across the first and fourth resonance capacitors 8 and 23,
Both V _C4 become 0V, and the voltage V _C2 across the second resonance capacitor 14 becomes the voltage E of the DC power supply 1 (in the case of the circuit of FIG. 11) or the output voltage E ₀ (in the case of the circuit of FIG. 13). Then, the main freewheeling diode 3 becomes conductive, and the current flowing through the first, fourth, and second resonance capacitors 8, 23, 14 is switched to the current flowing through the main freewheeling diode 3. The voltages V _TR1 and V _TR2 across the main transistor 2 and the auxiliary transistor 9 at this time are, as shown in FIGS. 12C and _12F , the DC power supply 1 when the circuit of FIG. 11 is used.
14 is equal to the voltage E of FIG.
It is equal to the output voltage E ₀ as shown in (C) and (F). When the main transistor 2 is off, the current I of the load 6 flows from the main freewheeling diode 3 to the reactor 4 in the case of the circuit of FIG. 11, and the current I _{0 of the} reactor 4 in the case of the circuit of FIG. Flows to the load 6 through the main free-wheeling diode 3. Note that FIG. 1 from time t ₁ to time t ₆
The operations of the circuits shown in FIGS. 1 and 13 are substantially the same as the operations of the circuits shown in FIGS. In addition, in the circuits of FIGS. 11 and 13, the first
The waveforms of the current I _L1 flowing in the resonance reactor 10, the current I _TR1 flowing in the main transistor 2, and the current I _D flowing in the main freewheeling diode 3 are shown in FIG. 2 (G) to (I) or FIG.
Since the waveforms are substantially the same as those shown in (G) to (I), the illustration in FIG. 12 or 14 is omitted.

As shown in FIGS. 12 (B) and 14 (B),
At t ₆ , the control circuit 7 changes the auxiliary control pulse signal voltage V _B2 applied to the base terminal of the auxiliary transistor 9 from the high level to the low level to turn the auxiliary transistor 9 from the ON state to the OFF state. At this time, the circulating current flowing through the auxiliary transistor 9 is changed to the third and fourth resonance capacitors 1
The electric current stored in the first and second resonance reactors 10 is discharged, and the energy stored in the first resonance reactor 10 is discharged, so that the third and fourth resonance capacitors 17 and 23 are charged in a sinusoidal waveform again from 0V. As a result, the voltage across the third and fourth resonance capacitors 17 and 23 is, as shown in the solid lines in FIGS. 12 (E) and 14 (E) and FIGS. 12 (G) and 14 (G). V _C3 and V _C4 rise in a gentle sine wave shape from 0V. Therefore, as shown by the solid lines in FIGS. 12 (F) and 14 (F), the voltage V _TR2 across the auxiliary switching element 9 rises in a sine wave shape gentler than 0 V, and at the same time for the first resonance. The current of the reactor 10 decreases like a cosine wave. Therefore, when the auxiliary transistor 9 is turned off, the voltage V across the auxiliary switching element 9 is reduced.
_{Since TR2} is 0V, zero voltage switching is performed.

As shown in FIGS. 12 (E) and (G) and FIGS. 14 (E) and (G), the voltages V _C3 and V across the third and fourth resonance capacitors 17 and 23 at t ₇ are obtained. _{When C4} reaches approximately the maximum value, that is, the voltage E of the DC power supply 1 (in the case of the circuit of FIG. 11) or the output voltage E ₀ (in the case of the circuit of FIG. 13),
As shown in (F) and FIG. 14 (F), the voltage V _TR2 across the auxiliary switching element 9 rapidly drops to 0V. Along with this, the resonance current component of the circulating current flowing through the first resonance reactor 10 becomes 0, and the current flowing through the first resonance reactor 10 becomes the current I of the load 6 (in the case of the circuit of FIG. 11) or the reactor. 4 current I ₀ (for the circuit of FIG. 13). The remaining energy of the first resonance reactor 10 in the circuit of FIG. 11 at this time is the second auxiliary freewheeling diode 12 and the second resonance reactor 1
5, the resonance current diode 16, the first auxiliary return diode 11, the first resonance reactor 10, and the circulating current diode 13 are fed back to the DC power supply 1. As a result, the current of the resonance reactor 10 continues to decrease linearly and the current I _TR1 of the main transistor 2 increases linearly from 0. And t ₈
In FIG. 2, the current of the first resonance reactor 10 becomes 0, and the current of the main transistor 2 becomes equal to the current I of the load 6. Thus, after t ₈ a current I flows to the load 6 through the main transistor 2 and the reactor 4 from the DC power source 1. At this time, the energy of the remaining first resonance reactor 10 in the circuit of FIG. 13 is the circulating current diode 13, the first resonance reactor 10, the first auxiliary return diode 11, and the resonance current diode. 16, the second resonance reactor 15 and the second auxiliary free-wheeling diode 12 supply the load 6 to the load 6. As a result, the current of the resonance reactor 10 continues to decrease linearly and the current I of the main transistor 2 is increased.
_TR1 increases linearly from 0. Then, at t ₈ (FIG. 4), the current of the first resonance reactor 10 becomes 0, and the current of the main transistor 2 becomes the current I of the reactor 4.
_Is equal to ₀ . Therefore, after t ₈ , the current I ₀ flows from the DC power supply 1 through the reactor 4 and the main transistor 2.

In the embodiment shown in FIGS. 11 and 13, t
During the period from _{6 to} t ₇ , the energy accumulated in the first resonance reactor 10 when the auxiliary transistor 9 is turned off charges the third and fourth resonance capacitors 17 and 23 in a sinusoidal waveform from 0V. Therefore, the time required for the charging voltage V _{C3 of} the third resonance capacitor 17 to reach the maximum value, that is, the charging time becomes long. Therefore, FIG.
2 (E) and the solid line portion of FIG. 14 (E), the slope of the voltage V _C3 across the third resonance capacitor 17 is compared with the case of the embodiment shown in FIGS. 1 and 3 (broken line portion). And then loosen. As a result, as shown by the solid lines in FIGS. 12 (F) and 14 (F), the voltage V
The slope of _TR2 is also gentler than in the case of the embodiment shown in FIGS. 1 and 3 (broken line portion). Therefore, the zero voltage switching when the auxiliary transistor 9 is turned off becomes more reliable as compared with the embodiment shown in FIGS. 1 and 3, and the switching loss can be further reduced.

Further, the circuits of the embodiments shown in FIGS. 1 and 3 can be modified to a system which does not use the auxiliary transistor 9 as shown in FIGS. 15 and 17, respectively. That is, the circuits of the embodiments shown in FIG. 15 and FIG.
6 and FIG. 27, the first resonance capacitor 8 is connected to the connection point between the main transistor 2 and the main freewheeling diode 3, and the first auxiliary return path is connected to the connection point between the first resonance capacitor 8 and the main transistor 2. The diode 11 is connected, and the first resonance reactor 10 and the resonance current diode 16 are connected in series between the first resonance capacitor 8 and the first auxiliary free-wheeling diode 11, and the first resonance capacitor 10 and the resonance current diode 16 are connected in series. The connection point between the capacitor 8 and the first resonance reactor 10 and FIG.
6 and the connection point of the main freewheeling diode 3 and the output capacitor 5 in FIG. 27, the second auxiliary freewheeling diode 12 is connected, and the connection point of the first auxiliary freewheeling diode 11 and the resonant current diode 16 is connected. 26 and FIG.
The second resonance capacitor 14 is connected between the main transistor 2 and the connection point of the DC power supply 1 in FIG. Further, the voltage E of the DC power supply 1 (in the case of the circuit of FIG. 15) or the output voltage E depending on the recovery and recovery characteristics of the main freewheeling diode 3 when the main transistor 2 is turned on.
_In order to avoid the short circuit condition of ₀ (in the case of the circuit of FIG. 17),
A current limiting reactor 21 is connected in series with the main return diode 3.

The operation of the circuit of the embodiment shown in FIG. 15 is as follows. As shown in FIG. 16 (A), when the main transistor 2 is in the ON state before t ₀ , the current I flows to the load 6 through the main transistor 2 and the reactor 4 as shown in FIG. 16 (B). At this time, as shown in FIG. 16 (E), the first resonance capacitor 8 is charged to the voltage E of the DC power supply 1 with the polarity shown in FIG. Figure 16 (A)
As shown in, at t ₀ , the main control pulse signal voltage V _B applied from the control circuit 7 to the base terminal of the main transistor 2 changes from high level to low level, and the main transistor 2
16B from the ON state to the OFF state, the current I _TR flowing through the main transistor 2, that is, the current I of the load 6 immediately changes to the first resonance capacitor 8 and the second resonance capacitor 8 as shown in FIG. 16B.
Is switched to the current flowing through the resonance capacitor 14. At this time, the first resonance capacitor 8 gradually discharges, and as shown in FIG.
The voltage V _C1 across both ends of the voltage drops linearly from the voltage E of the DC power supply 1. Along with this, the second resonance capacitor 1
4 is gradually charged from 0 V, and the voltage V _C2 across the second resonance capacitor 14 increases linearly from 0 V as shown in FIG. 16 (D). As a result, the voltage V _TR across the main transistor 2 increases linearly from 0 V as shown in FIG. Therefore, when the main transistor 2 is turned off, zero voltage switching is performed in which the voltage waveform and the current waveform do not overlap each other.

As shown in FIGS. 16E and 16D, at t ₁ , the voltages V _C1 and V _C2 across the first and second resonance capacitors 8 and 14 are 0 V and the voltage of the DC power supply 1, respectively. When it becomes E, the main freewheeling diode 3 becomes conductive,
The current flowing in the first and second resonance capacitors 8 and 14 is switched to the current I _D flowing in the main freewheeling diode 3 as shown in FIG. 16 (F). The voltage V _TR across the main transistor 2 at this time is equal to the voltage E of the DC power supply 1 as shown in FIG. When the main transistor 2 is off, the current I of the load 6 flows from the main free wheeling diode 3 to the reactor 4.

As shown in FIG. 16A, at t ₂ , the main control pulse signal voltage V _B applied from the control circuit 7 to the base terminal of the main transistor 2 changes from low level to high level, and the main transistor 2 is turned on. When the state changes from the off state to the on state, the voltage V _TR across the main transistor 2 rapidly drops to 0 V as shown in FIG. 16 (C). At the same time,
The second resonance capacitor 14 starts discharging, and the first and second resonance capacitors 8 and 14 and the first resonance reactor 10 resonate to cause the second resonance capacitor 14, the main transistor 2, and A resonance current flows through the path of the first resonance capacitor 8, the first resonance reactor 10 and the resonance current diode 16. Therefore, the current I _L flowing through the first resonance reactor 10 changes in a sine wave shape as shown in FIG. At this time, the first resonance capacitor 8 is charged in a cosine wave shape, and the voltage V _C1 across the first resonance capacitor 8 rises in a cosine wave shape from 0V as shown in FIG. 16 (E). go. At the same time, the voltage V _C2 across the second resonance capacitor 14 drops from the voltage E in a cosine wave shape as shown in FIG. This allows
The current I _TR of the main transistor 2 increases sinusoidally from 0 as shown in FIG. 16 (B). Therefore, when the main transistor 2 is turned on, zero current switching is achieved in which the voltage waveform and the current waveform do not overlap each other.

On the other hand, the current _ID flowing in the main free-wheeling diode 3 decreases linearly as shown in FIG. 16 (F) by the self-induction action of the current-limiting reactor 21, and becomes 0 at t ₃ . The current I of the load 6 switches to the current I _TR flowing through the main transistor 2. Then, at t ₄ , the current I _L of the first resonance reactor 10 becomes as shown in FIG.
When it becomes 0 as shown in FIG. 16, the voltages V _C1 and V _C2 across the first and second resonance capacitors 8 and 14 become as shown in FIG.
As shown in (D), the voltage E of the DC power source 1 and 0V, respectively
Becomes At this time, the current I _TR of the main transistor 2 is as shown in FIG.
6 (B), it becomes equal to the current I of the load 6. Therefore, after t ₄ , the current I flows from the DC power supply 1 to the load 6 through the main transistor 2 and the reactor 4.

The operation of the circuit of the embodiment shown in FIG. 17 is as follows. As shown in FIG. 18A, when the main transistor 2 is in the ON state before t ₀ ,
As shown in (B), the current I ₀ flows through the path of the reactor 4 and the main transistor 2. At this time, as shown in FIG. 18 (E), the first resonance capacitor 8 is charged to the terminal voltage of the load 6, that is, the output voltage E _{0 with the} polarity shown in FIG. As shown in FIG. 18A, at t ₀ , the main control pulse signal voltage V _B applied from the control circuit 7 to the base terminal of the main transistor 2 changes from high level to low level, and the main transistor 2 is turned on. When in the off state, the current I of the main transistor 2 as shown in FIG.
_TR immediately switches to the current flowing in the second resonance capacitor 14, and the current I ₀ of the reactor 4 switches to the current flowing in the first resonance capacitor 8. At this time, the first
The resonance capacitor 8 of FIG.
As shown in (E), the voltage V _C1 across the first resonance capacitor 8 drops linearly from the voltage E of the DC power supply 1. Along with this, the second resonance capacitor 14 becomes 0V.
18D, the voltage V _C2 across the second resonance capacitor 14 rises linearly from 0V as shown in FIG. 18D. As a result, the voltage V _TR across the main transistor 2 linearly increases from 0 V as shown in FIG. Therefore, when the main transistor 2 is turned off, zero voltage switching is performed in which the voltage waveform and the current waveform do not overlap each other.

As shown in FIGS. 18E and 18D, at t ₁ , the voltages V _C1 and V _C2 across the first and second resonance capacitors 8 and 14 become 0 V and the output voltage E ₀ , respectively. Then, the main freewheeling diode 3 becomes conductive, and the current I _D flowing in the first and second resonance capacitors 8 and 14 flows in the main freewheeling diode 3 as shown in FIG. 18 (F). Switch to. The voltage V _TR across the main transistor 2 at this time is the output voltage E ₀ as shown in FIG.
be equivalent to. Also, when the main transistor 2 is off,
The current I _{0 of the} reactor 4 is flowing to the load 6 through the main return diode 3.

As shown in FIG. 18A, at t ₂ , the main control pulse signal voltage V _B applied from the control circuit 7 to the base terminal of the main transistor 2 changes from low level to high level, and the main transistor 2 is turned on. When the state changes from the off state to the on state, the voltage V _TR across the main transistor 2 rapidly drops to 0 V as shown in FIG. At the same time,
The second resonance capacitor 14 starts discharging, and the first and second resonance capacitors 8 and 14 and the first resonance reactor 10 resonate to cause the second resonance capacitor 14 and the resonance current diode. 16, 1st resonance reactor 1
A resonance current flows through the path of 0, the first resonance capacitor 8 and the main transistor 2. Therefore, the current I _L flowing through the first resonance reactor 10 changes in a sine wave shape as shown in FIG. At this time, the first resonance capacitor 8 is charged in a cosine wave shape, and the voltage V _C1 across the first resonance capacitor 8 is 0 V as shown in FIG. 18 (E).
It goes up like a cosine wave. At the same time, the voltage V _C2 across the second resonance capacitor 14 drops from the voltage E in a cosine wave shape as shown in FIG. As a result, the current I _TR of the main transistor 2 increases in a sinusoidal manner from 0 as shown in FIG. Therefore, when the main transistor 2 is turned on, zero current switching is achieved in which the voltage waveform and the current waveform do not overlap each other.

On the other hand, the main wheeling diode current I _D flowing in the 3 went decreases linearly as shown in FIG. 18 (F) by self-induction action of the current limiting reactor 21, becomes zero at t ₃ The current I _{0 of the} reactor 4 switches to the current I _TR flowing through the main transistor 2. And
At t ₄ , when the current I _L of the first resonance reactor 10 becomes 0 as shown in FIG. 18 (G), the voltages V _C1 and V _C2 across the first and second resonance capacitors 8 and 14 become FIG.
As shown in (E) and (D), output voltages E ₀ and 0, respectively
It becomes V. At this time, the current I _TR of the main transistor 2 becomes equal to the current I _{0 of the} reactor 4 as shown in FIG. 18 (B). Therefore, after t ₄ , the current I ₀ flows from the DC power supply 1 through the reactor 4 and the main transistor 2.

As described above, in the embodiment shown in FIGS. 15 and 17, zero voltage and zero current switching is achieved when the main transistor 2 is turned off and turned on. Therefore, as in the embodiment shown in FIGS. 1 and 3, The switching loss of the main transistor 2 can be reduced. Further, spike-like surge voltage and surge current generated at the time of turning on and off of the main transistor 2 are also absorbed by the resonance action of the first and second resonance capacitors 8 and 14 and the first resonance reactor 10. As in the embodiment shown in FIGS. 1 and 3, it is possible to reduce the surge voltage, surge current and noise during the on / off operation of the main transistor 2. Further, since the auxiliary transistor 9 is not used in the embodiments shown in FIGS. 15 and 17, there is an advantage that the circuit configuration can be simplified as compared with the embodiments shown in FIGS. Note that in the circuits of the embodiments shown in FIGS. 15 and 17, when the recovery recovery characteristic of the main freewheeling diode 3 when the main transistor 2 is turned on can be ignored, the current limiting reactor 21 as shown in FIGS. Can be omitted. Also,
The current limiting reactor 21 in the circuit of the embodiment shown in FIG. 15 and FIG. 17 works even if connected to the main transistor 2 in series or the DC power supply 1 (in the case of FIG. 15) or the load 6 (in the case of FIG. 17). And the effect is the same.

The circuits of the embodiments shown in FIGS. 15 and 17 can be modified in the same manner as the embodiments of FIGS. 11 and 13 described above, as shown in FIGS. 21 and 22, respectively.
That is, in the circuits of the embodiments shown in FIGS. 21 and 22, the connection point of the main freewheeling diode 3 and the first resonance capacitor 8 and the first auxiliary freewheeling diode 11 and the main transistor in FIGS. 15 and 17, respectively. The third auxiliary freewheeling diode 22 and the third resonance capacitor 17 are connected in series between the second auxiliary connection diode 22 and the third resonance capacitor 17 and the main connection point between the third auxiliary freewheeling diode 22 and the third resonance capacitor 17 is connected. A fourth auxiliary return diode 24 is connected between the return diode 3 and the connection point of the output capacitor 5. 21 and 22, the circuit of the embodiment shown in FIG.
5 and the current limiting reactor 21 shown in FIG. 17 are connected in series with the main transistor 2. In the circuits shown in FIGS. 21 and 22, the charging time and the discharging time of the second resonance capacitor 14 when the main transistor 2 is turned off and on are longer than those of the circuits shown in FIGS. The zero voltage and zero current switching at turn-off and turn-on of the transistor 2 is shown in FIG.
Further, compared to the case of the embodiment shown in FIG. 17, it becomes more reliable and the switching loss can be further reduced.

Furthermore, the embodiment of the present invention is not limited to the above embodiment, and various modifications can be made. For example, in each of the above-described embodiments, an example in which the junction type bipolar transistor is used as the main switching element and the auxiliary switching element has been shown. , Other switching elements such as SCR (reverse blocking three-terminal thyristor) may be used. Especially, MOS-FET
When using a built-in diode that is formed integrally with the MOS-FET, it is possible to use
The circulating current diode 13 in the embodiments shown in FIGS. 11 and 13 can be omitted. The main switching element and the auxiliary switching element are not limited to the same kind of combination.

By the way, the DC power supply 1 in each of the above-described embodiments is actually a single-phase or three-phase commercial AC power supply 25, 26 and a single-phase or three-phase commercial AC as shown in FIGS. Often composed of single-phase or three-phase rectifier bridge circuits 27, 28 as a rectifier circuit for converting single-phase or three-phase AC voltage of the power supplies 25, 26 into DC voltage (of course, the DC power supply 1 is a dry battery). It goes without saying that batteries and batteries can also be used). For example, the circuit of the embodiment shown in FIG. 23 is the step-up chopper type DC-DC converter A shown in FIG. 3, FIG. 6, FIG. 8, FIG. 10 or FIG. 13 or the step-up chopper type DC shown in FIG. -D
The DC power supply 1 in the C converter B is composed of a single-phase commercial AC power supply 25 and a single-phase rectification bridge circuit 27, and the reactor 4 in the step-up chopper type DC-DC converter A or B is connected to the AC input side of the single-phase rectification bridge circuit 27. Connected to. Of course, it is also possible to configure the DC power supply 1 with the single-phase commercial AC power supply 25 and the single-phase rectification bridge circuit 27 without changing the connection position of the reactor 4 in the step-up chopper type DC-DC converter A or B. Also, FIG.
4, the circuit of the embodiment shown in FIG.
Alternatively, the DC power supply 1 in the step-up chopper type DC-DC converter A shown in FIG. 13 or the step-up chopper type DC-DC converter B shown in FIG. 17, FIG. 20 or FIG. And a step-up chopper type DC-DC converter A or B
Instead of the reactor 4 in the three-phase rectification bridge circuit 28
The reactors 4a, 4b and 4c are connected to the respective phases on the AC input side of the. Of course, also in this case, the DC power supply 1 can be configured by the three-phase commercial AC power supply 26 and the three-phase rectification bridge circuit 28 without changing the connection position of the reactor 4 in the step-up chopper type DC-DC converter A or B. is there. The step-down chopper type DC-D shown in FIG. 1, FIG. 5, FIG. 7, FIG. 9, FIG. 11, FIG. 15, FIG. 19 or FIG.
Also in the case of the C converter, the DC power supply 1 can be configured by the single-phase or three-phase commercial AC power supplies 25, 26 and the single-phase or three-phase rectifying bridge circuits 27, 28.
The rectifier circuit is not limited to the single-phase or three-phase rectifier bridge circuits 27 and 28 shown in FIGS. 23 and 24, and may be a single-phase or three-phase half-wave rectifier circuit, a full-wave rectifier circuit or a double-wave rectifier circuit as needed. Other rectifier circuits such as voltage rectifier circuits can also be used.

The circuit of the embodiment shown in FIG. 25 is the step-up chopper type DC-D shown in FIG. 17, 20 or 22.
The DC power supply 1 in the C converter B is connected in parallel to the three-phase commercial AC power supply 26, and each of the six AC-DC conversion transistors 29 to 34 and each of the transistors 29 to 34 as AC-DC conversion switching elements. Three-phase AC-DC for converting the three-phase AC voltage of the three-phase commercial AC power supply 26 into a DC voltage by having six circulating current diodes 35-40 and controlling each of the transistors 29-34 on / off. A converter circuit 41 is used, and instead of the reactor 4 in the step-up chopper type DC-DC converter B, the reactors 4a, 4b, 4c are connected to the respective phases on the AC input side of the three-phase AC-DC converter circuit 41, and three-phase A power source resonance capacitor 42 is connected between a pair of lines on the DC output side of the AC-DC converter circuit 41. When MOS-FETs are used as the six AC-DC converting transistors 29 to 34, a built-in diode integrally formed therewith can be used. Therefore, the six circulating current diodes 35 to 40 should be connected. It can be omitted. Further, in the case of a single-phase AC input, instead of the three-phase AC-DC converter circuit 41, four AC-DC conversion transistors and four circulating current diodes connected in parallel with the same configuration are used. It will be easily understood that a single-phase AC-DC converter circuit may be used.

[0060]

According to the present invention, since zero voltage or zero current switching of the switching element can be easily achieved, the overlapping portion of the voltage waveform and the current waveform of the switching element can be reduced to reduce the chopper type DC-DC converter. It is possible to reduce power loss at the time of on / off operation of the switching element, that is, switching loss. Further, due to the resonance action of the resonance reactor and the resonance capacitor, surge voltage, surge current and noise during the switching operation of the switching element of the chopper type DC-DC converter can be reduced. Further, since the auxiliary switching element is changed from the ON state to the OFF state after the charging voltage of the first resonance capacitor reaches the maximum value and the voltage of the third resonance capacitor is discharged to 0V,
The time when the auxiliary switching element is turned off can be clearly set. Further, when the energy feedback rectifying element is additionally connected, the maximum value of the charging voltage of the third resonance capacitor is prevented from becoming higher than the power supply voltage (in case of step-down converter) or the output voltage (in case of step-up converter). Then, the power loss of the chopper type DC-DC converter can be further reduced. Further, when the reverse charge prevention rectifying element is connected in parallel with the third resonance capacitor or the auxiliary switching element, the third resonance capacitor is prevented from being charged with the reverse polarity, and the chopper type DC- It is possible to further reduce the power loss of the DC converter. Further, when the discharge prevention rectifying element is connected in series with the auxiliary switching element and the first resonance reactor, the power loss due to the discharge of the parasitic capacitor of the auxiliary switching element is reduced, and the power of the chopper type DC-DC converter is reduced. It is possible to further reduce the loss. Also, the third and fourth
If the auxiliary reflux rectifying element and the fourth resonance capacitor are additionally connected, the charging time of the third resonance capacitor when the auxiliary switching element is turned off becomes longer, and zero voltage switching is performed when the auxiliary switching element is turned off. Is more reliable, it is possible to further reduce the switching loss and further reduce the power loss of the chopper type DC-DC converter. In addition, in the method that does not use the auxiliary switching element, it is possible to reduce switching loss, surge voltage, surge current, and noise during on / off operation of the switching element with a simple circuit configuration, thus reducing the number of parts. It is possible to reduce the manufacturing cost and further reduce the power loss of the chopper type DC-DC converter.

[Brief description of drawings]

FIG. 1 is an electric circuit diagram showing an embodiment of a step-down chopper type DC-DC converter according to the present invention.

FIG. 2 is a waveform diagram showing the voltage and current of each part of the circuit of FIG.

FIG. 3 is an electric circuit diagram showing an embodiment of a step-up chopper type DC-DC converter according to the present invention.

FIG. 4 is a waveform diagram showing the voltage and current of each part of the circuit of FIG.

5 is an electrical circuit diagram showing a first modified embodiment of the circuit of FIG.

FIG. 6 is an electrical circuit diagram showing a first modified embodiment of the circuit of FIG.

7 is an electrical circuit diagram showing a second modified embodiment of the circuit of FIG.

8 is an electrical circuit diagram showing a second modified embodiment of the circuit of FIG.

FIG. 9 is an electrical circuit diagram showing a third modified embodiment of the circuit of FIG.

10 is an electrical circuit diagram showing a third modified embodiment of the circuit of FIG.

FIG. 11 is an electrical circuit diagram showing a fourth modified embodiment of the circuit of FIG.

FIG. 12 is a waveform diagram showing the voltage of each part of the circuit of FIG.

13 is an electric circuit diagram showing a fourth modified embodiment of the circuit of FIG.

FIG. 14 is a waveform diagram showing the voltage of each part of the circuit of FIG.

FIG. 15 is an electric circuit diagram showing a fifth modified embodiment of the circuit of FIG.

16 is a waveform diagram showing the voltage and current of each part of the circuit of FIG.

17 is an electrical circuit diagram showing a fifth modified embodiment of the circuit of FIG.

FIG. 18 is a waveform diagram showing the voltage and current of each part of the circuit of FIG.

FIG. 19 is an electric circuit diagram showing a first modified embodiment of FIG.

FIG. 20 is an electric circuit diagram showing a first modified embodiment of FIG.

FIG. 21 is an electric circuit diagram showing a second modified embodiment of FIG.

FIG. 22 is an electric circuit diagram showing a second modified embodiment of FIG.

FIG. 23 is an electric circuit diagram showing an embodiment in which the chopper type DC-DC converter of the present invention is connected to a single-phase rectification bridge circuit.

FIG. 24 is an electric circuit diagram showing an embodiment in which the chopper type DC-DC converter of the present invention is connected to a three-phase rectification bridge circuit.

FIG. 25 is an electric circuit diagram showing an embodiment in which the chopper type DC-DC converter of the present invention is connected to a three-phase AC-DC converter circuit.

FIG. 26 is an electric circuit diagram showing a conventional step-down chopper type DC-DC converter.

FIG. 27 is an electric circuit diagram showing a conventional step-up chopper type DC-DC converter.

FIG. 28 is a waveform diagram showing an overlapping portion of the switching voltage waveform and the switching current waveform of the circuits of FIGS. 26 and 27.

[Explanation of symbols]

1. ． ． DC power supply, 2. ． ． Main transistor (main switching element), 3. ． ． Main freewheeling diode (main freewheeling rectifying element), 4. ． ． Reactor, 5. ． ． Output capacitor, 6. ． ． Load, 7. ． ． Control circuit, 8, 14, 1
7, 23. ． ． First to fourth resonance capacitors,
9. ． ． Auxiliary transistor (auxiliary switching element),
10,15. ． ． First and second resonance reactors, 1
1, 12, 22, 24. ． ． First to fourth auxiliary return diodes (first to fourth auxiliary return rectifying elements), 13,
35-40. ． ． Circulating current diode (circulating current rectifying element), 16. ． ． Resonant current diode (resonant current rectifying element), 18. ． ． Energy feedback diode (energy feedback rectifier), 19. ． ． Reverse charge prevention diode (reverse charge prevention rectifier), 20. ． ． Discharge prevention diode (discharge prevention rectifying element), 21. ． ．
Current limiting reactor, 25. ． ． Single-phase commercial AC power supply (AC power supply), 26. ． ． Three-phase commercial AC power supply (AC power supply), 2
7. ． ． Single-phase rectifier bridge circuit (rectifier circuit), 2
8. ． ． Three-phase rectifier bridge circuit (rectifier circuit), 29 to 3
4. ． ． AC-DC conversion transistor (AC-DC conversion switching element), 41. ． ． Three-phase AC-DC converter circuit (AC-DC converter circuit), 4
2. ． ． Power source resonance capacitor, A, B. ． ． Step-up chopper type DC-DC converter

Claims (20)

[Claims] 1. A main switching element, a reactor and a main return rectifying element are connected in a T-shape between a DC power supply and a load, an output capacitor is connected in parallel with the load, and the main capacitor is connected to one end of the output capacitor. In a chopper type DC-DC converter, which is connected to one end of a reflux rectifying element and which controls the main switching element to be turned on / off to supply a DC output of a voltage different from the voltage of the DC power source to the load, A series circuit of an auxiliary switching element and a first resonance reactor is connected in parallel with the main switching element, a first auxiliary return rectifying element is connected to a connection point of the series circuit, and the first resonance reactor and the first resonance reactor are connected. A first resonance capacitor is connected to a connection point of the main switching element,
A second resonance reactor and a resonance current rectification element are connected in series between the first resonance capacitor and the first auxiliary return rectification element, and the first resonance capacitor and the second resonance capacitor are connected in series. A second auxiliary return rectifier element is connected between a connection point of the resonance reactor and a connection point of the main return rectifier element and the output capacitor, and a rectifier element formed integrally with the main switching element or A circulating current rectifying element composed of an independent rectifying element is connected in parallel with the main switching element, a second resonance capacitor is connected in parallel with the circulating current rectifying element, and the first auxiliary return rectifying element is connected. And a third resonance capacitor connected between a connection point of the rectifying element for resonance current and a connection point of the main switching element and the auxiliary switching element, and the main switching element. Before switching the child from the OFF state to the ON state, the auxiliary switching element is changed from the OFF state to the ON state, and during the ON period of the auxiliary switching element, the charging voltage of the first resonance capacitor reaches the maximum value and A chopper type DC-DC converter, characterized in that the auxiliary switching element is switched from an on state to an off state after the voltage of the third resonance capacitor is discharged to 0V. 2. An energy feedback rectifying element is connected between a connection point of the first resonance reactor and the first auxiliary return rectifying element and a connection point of the main return rectifying element and the output capacitor. The chopper type DC-DC converter according to claim 1. 3. An energy feedback rectifying element is connected between a connection point of the third resonance capacitor and the first auxiliary return rectifying element and a connection point of the main return rectifying element and the output capacitor. The chopper type DC-DC converter according to claim 1. 4. An energy feedback rectifying element is connected between a connection point of the second resonance reactor and the resonance current rectifying element and a connection point of the main return rectifying element and the output capacitor. Item 1 "chopper type DC
-DC converter. 5. The chopper type DC-DC converter according to claim 1, wherein a reverse charge preventing rectifying element is connected in parallel with the third resonance capacitor. 6. The chopper type DC-DC converter according to claim 1, wherein a reverse charge prevention rectifying element is connected in parallel with the auxiliary switching element. 7. The auxiliary switching element and the first
2. The chopper type DC-DC converter according to claim 1, wherein a discharge preventing rectifying element is connected in series with the resonance reactor of FIG. 8. A third point between a connection point of the first resonance reactor and the first resonance capacitor and a connection point of the first auxiliary return rectifying element and the first resonance reactor. Of the auxiliary reflux rectifier element and the fourth resonance capacitor are connected in series, and the connection point of the third auxiliary reflux rectifier element and the fourth resonance capacitor and the main reflux rectifier element and the output capacitor The chopper type DC-DC converter according to claim 1, wherein a fourth auxiliary return rectifying element is connected to the connection point. 9. One of the main terminals of the main switching element is connected to one end of the DC power supply, and the main return rectifying element is between the other main terminal of the main switching element and the other end of the DC power supply. Is connected between the load and the connection point of the main switching element and the main return rectifying element, by controlling the main switching element on and off, from the voltage of the DC power supply. The chopper type DC-DC converter according to any one of "claim 1" to "claim 8", in which a direct current output having a low voltage is supplied to the load. 10. The chopper type DC-DC converter according to claim 9, wherein the DC power supply includes an AC power supply and a rectifier circuit that converts an AC voltage of the AC power supply into a DC voltage. 11. The reactor is connected to one line of the DC power supply, and one main terminal of the main switching element is connected through at least the reactor,
The other main terminal of the main switching element is connected to the other line of the DC power supply, the main return rectifying element is connected between one main terminal of the main switching element and the load, the main switching The chopper type DC- according to any one of claims 1 to 8 in which a DC output having a voltage higher than a voltage of the DC power supply is supplied to the load by controlling ON / OFF of the element. DC converter. 12. The DC power supply includes an AC power supply and a rectifier circuit that converts an AC voltage of the AC power supply into a DC voltage, and the reactor is connected to an AC input side or a DC output side of the rectifier circuit. The chopper type DC-DC converter according to claim 11. 13. A main switching element, a reactor and a main return rectifying element are connected in a T shape between a DC power source and a load, and an output capacitor is connected in parallel with the load,
One end of the main return rectifying element is connected to one end of the output capacitor, and a chopper for supplying a DC output of a voltage different from the voltage of the DC power supply to the load by controlling ON / OFF of the main switching element. In a DC-DC converter of the type, a first resonance capacitor is connected to a connection point of the main switching element and the main return rectifying element, and a first resonance capacitor is connected to a connection point of the first resonance capacitor and the main switching element. The auxiliary reflux rectifying element is connected, and the resonance reactor and the resonance current rectifying element are connected in series between the first resonance capacitor and the first auxiliary reflux rectifying element. A second auxiliary return rectifier element between a connection point of the resonance capacitor and the resonance reactor and a connection point of the main return rectifier element and the output capacitor. A second resonance capacitor is connected between a connection point of the first auxiliary return rectification element and the resonance current rectification element and a connection point of the main switching element and the DC power supply, When the main switching element is turned off, the first resonance capacitor is discharged and the second resonance capacitor is gradually charged, and when the main switching element is turned on. A chopper type DC-characterized in that the second resonance capacitor is discharged and the first and second resonance capacitors resonate with the resonance reactor to cause a resonance current to flow in the main switching element. DC converter. 14. The chopper type DC-DC converter according to claim 13, wherein a current limiting reactor is connected in series with the main return rectifying element or the main switching element. 15. A third point is provided between a connection point of the main return rectifying element and the first resonance capacitor and a connection point of the first auxiliary return rectifying element and the main switching element.
And a third resonance capacitor are connected in series, and the third auxiliary circulation rectification element and the third resonance condenser are connected in series.
"Claim 13" or "Claim 1", wherein a fourth auxiliary return rectifier element is connected between the connection point of the resonance capacitor of and the connection point of the main return rectifier element and the output capacitor.
4) The chopper type DC-DC converter described in 4 above. 16. One main terminal of the main switching element is connected to one end of the DC power supply, and the main return rectifying element is between the other main terminal of the main switching element and the other end of the DC power supply. Is connected between the load and the connection point of the main switching element and the main return rectifying element, by controlling the main switching element on and off, from the voltage of the DC power supply. The chopper type DC-DC converter according to any one of "claim 13" to "claim 15", wherein a direct current output of a low voltage is supplied to the load. 17. The chopper type DC-D according to claim 16, wherein the DC power supply includes an AC power supply and a rectifier circuit that converts an AC voltage of the AC power supply into a DC voltage.
C converter. 18. The reactor is connected to one line of the DC power supply, and one main terminal of the main switching element is connected through at least the reactor,
The other main terminal of the main switching element is connected to the other line of the DC power supply, the main return rectifying element is connected between one main terminal of the main switching element and the load, the main switching The chopper type DC- according to any one of claims 13 to 15 in which a DC output having a voltage higher than the voltage of the DC power supply is supplied to the load by controlling ON / OFF of the element. DC converter. 19. The DC power supply includes an AC power supply and a rectifier circuit that converts an AC voltage of the AC power supply into a DC voltage, and the reactor is connected to an AC input side or a DC output side of the rectifier circuit. The chopper type DC-DC converter according to claim 18. 20. The DC power supply is an AC power supply and an AC-
A switching element for direct current conversion and a rectifying element for circulating current composed of a rectifying element integrally formed with the switching element for AC-DC conversion or independently connected in parallel, and the switching element for AC-DC conversion And an AC-DC converter circuit that converts the AC voltage of the AC power supply into a DC voltage by controlling ON / OFF, the reactor is connected to the AC input side of the AC-DC converter circuit, and the AC- 19. The chopper type D according to claim 18, wherein a power source resonance capacitor is connected between a pair of lines on the DC output side of the DC converter circuit.
C-DC converter.