JPH06269165A - Pwm dc-dc converter - Google Patents

Abstract

PURPOSE:To reduce switching loss, surge voltage and current by providing a control circuit for detecting the output voltage from a smoothing circuit and delivering an auxiliary control pulse signal to an auxiliary switching element prior to provision of a main control signal to a main switching element. CONSTITUTION:The converter commutates the current I of a load 7 to a resonance capacitor 9 when a main control signal voltage applied from a control circuit 8 to the base terminal of a transistor 3 makes a transition from High to Low level thus turning the transistor 3 OFF. Voltage across the transistor 3 increases linearly up to the sum E of voltages E/2 of first and second DC power supplies 1, 2. A thyristor 10 is turned ON before the transistor 3 is turned ON to feed resonance capacitor 9 and reactor 11 with a resonance current. Voltage across the transistor 3 is lowered according to cosine function and the transistor 3 is turned ON when the voltage goes zero.

Description

Detailed Description of the Invention

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a PWM (pulse width modulation) type DC-DC converter, particularly a step-down type PWM type D
The present invention relates to a C-DC converter.

[0002]

2. Description of the Related Art A PWM (Pulse Width Modulation) type DC-D which controls a ratio of ON / OFF periods of switching elements
C converters, so-called chopper circuits, have been widely used in power supply circuits for electronic devices and the like. The conventional PWM type DC-DC converter shown in FIG. 3 has a transistor 3 as a main switching element whose collector terminal is connected to the DC power supply 1, and a freewheeling connection connected between the emitter terminal of the transistor 3 and the DC power supply 1. A smoothing circuit composed of a diode 4, a smoothing reactor 5 and a smoothing capacitor 6 connected to both ends of the free wheeling diode 4, a load 7 connected to both ends of the smoothing capacitor 6 of the smoothing circuit, and an output voltage of the smoothing circuit are detected. And a control circuit 8 for applying a control pulse signal to the base terminal of the transistor 3. This P
In the WM type DC-DC converter, the turn-on period of the transistor 3 is controlled by changing the time width of the control pulse signal applied to the base terminal of the transistor 3 in proportion to the fluctuation of the output voltage of the smoothing circuit, and the load 7 We are trying to stabilize the power supplied to.

[0003]

By the way, the PW of FIG.
The M-type DC-DC converter has a drawback that a large switching loss occurs due to the overlap between the collector-emitter voltage waveform of the transistor 3 and the collector current waveform of the transistor 3 during the on / off operation of the transistor 3. Further, there is a drawback that spike-like surge voltage and current are generated when the voltage and current waveforms rise during the on / off operation of the transistor 3.

Therefore, an object of the present invention is to provide a PWM type DC-DC converter capable of reducing switching loss, surge voltage and current.

[0005]

A PWM type D according to the present invention
The C-DC converter includes a first and a second connected in series.
Of the DC power supply, a main switching element having one main terminal connected to the first DC power supply, a resonance capacitor connected in parallel with the main switching element, and a DC power supply of the first and second DC power supplies. Between the series circuit of the auxiliary switching element and the resonance reactor connected between the connection point and the other main terminal of the main switching element, and between the other main terminal of the main switching element and the second DC power supply. A connected return rectifying element, and a smoothing circuit including a reactor and a capacitor connected to both ends of the return rectifying element,
An auxiliary control pulse signal is applied to the control terminal of the auxiliary switching element before the load connected to the smoothing circuit and the output voltage of the smoothing circuit are detected and the main control pulse signal is applied to the control terminal of the main switching element. And a control circuit for applying the control signal.

[0006]

When the main switching element is switched to the off state while the current is flowing to the load while the main switching element is on, the load current is commutated from the main switching element to the resonance capacitor, and the resonance capacitor is The voltage is charged to the sum of the voltages of the second DC power supply, and the voltage across the resonance capacitor, that is, the voltage across the main switching element, rises gently. Thereby, the switching loss at the time of switching the main switching element from the on state to the off state can be reduced. Before applying the main control pulse signal to the control terminal of the main switching element to turn on the main switching element, apply the auxiliary control pulse signal to the control terminal of the auxiliary switching element to turn on the auxiliary switching element. Then, a resonance current flows through the resonance capacitor and the resonance reactor, and the voltage across the resonance capacitor, that is, the voltage across the main switching element, drops gently. When this voltage becomes 0 V, the main switching element is turned on, whereby the switching loss at the time of switching the main switching element from the off state to the on state can be reduced. As described above, it is possible to reduce the switching loss during the on / off operation of the main switching element. In addition, the spike-shaped surge voltage and current that occurs when the main switching element is switched on and off is absorbed by the resonance capacitor and resonance reactor, so the surge voltage and current when the main switching element is on and off is reduced. can do.

[0007]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS An embodiment of a PWM type DC-DC converter according to the present invention will be described below with reference to FIGS. However, in FIG. 1, the same parts as those shown in FIG. 3 are designated by the same reference numerals, and the description thereof will be omitted. PWM type D of this embodiment
As shown in FIG. 1, the C-DC converter has first and second DC power supplies 1 and 2 connected in series, and a collector terminal (one main terminal) connected to the first DC power supply 1. A transistor 3 as a main switching element, a resonance capacitor 9 connected in parallel with the transistor 3, a connection point between the first and second DC power supplies 1 and 2, and an emitter terminal (the other main terminal) of the transistor 3. A series circuit of a thyristor 10 and a resonance reactor 11 as an auxiliary switching element connected between the two, and a freewheeling diode (freewheeling rectifying element) connected between the emitter terminal of the transistor 3 and the second DC power supply 2. 4 and a smoothing reactor 5 and a smoothing capacitor 6 connected to both ends of the free wheeling diode 4.
Before the main control pulse signal is applied to the base terminal of the transistor 3 by detecting the output voltage of the smoothing circuit and the load 7 connected to both ends of the smoothing capacitor 6 of the smoothing circuit. And a control circuit 8 for applying an auxiliary control pulse signal to the terminal. The voltages of the first and second DC power supplies 1 and 2 are E / 2 in this embodiment. These voltages E / 2 can be obtained, for example, by dividing the DC voltage E obtained by rectifying the AC voltage of the commercial AC power supply into 1: 1 by two voltage dividing capacitors connected in series. Of course, two DC power supplies with output voltage E / 2 may be prepared, and each of them may be connected in series. Further, in this embodiment, the transistor 3 and the thyristor 10 are respectively a junction type power transistor and a reverse blocking circuit 3.
A terminal thyristor (SCR) is used. Although not particularly shown, the control circuit 8 includes an oscillation circuit section that generates a triangular wave voltage of a constant cycle, an error amplification circuit section that performs operational amplification of the error voltage of the output voltage of the smoothing circuit with respect to the reference voltage, and an error amplification circuit section. A comparator circuit for comparing the error output voltage and the triangular wave voltage of the oscillator circuit section, and a transistor 3 for generating a main control pulse signal having a time width proportional to the output voltage of the comparator circuit section.
Of the main control pulse generating circuit section to be applied to the base terminal of the thyristor, and an auxiliary control for generating an auxiliary control pulse signal of a certain time width to be applied to the gate terminal of the thyristor 10 before the main control pulse signal of the main control pulse generating circuit section rises And a pulse generation circuit section. The time width of the auxiliary control pulse signal generated from the auxiliary control pulse generation circuit section is extremely smaller than the off time of the transistor 3.

In the above structure, as shown in FIG. 2A, the main control pulse signal voltage V _B applied from the control circuit 8 to the base terminal of the transistor 3 changes from high level to low level, and the transistor 3 is turned on. When the state changes to the off state, the current I _TR flowing in the transistor 3, that is, the current I of the load 7 is commutated to the resonance capacitor 9 (I _CS ) as shown in FIGS. 2C and _2E . At this time, as shown in FIG. 2 (F), the resonance capacitor 9 is charged to the sum of the voltages E / 2 of the first and second DC power supplies 1 and 2, that is, the voltage E, and the resonance capacitor 9 The voltage V _CS across the terminals, that is, the voltage V _TR across the transistor 3, increases linearly. When the resonance capacitor 9 is charged to the voltage E, the freewheeling diode 4 is forward biased, and the current I _CS of the resonance capacitor 9, that is, the current I of the load 7 is shown in FIGS. 2 (E) and 2 (D). Are commutated to the free wheeling diode 4 ( _ID ). At this time, the voltage V _CS across the resonance capacitor 9, that is, the voltage V _TR across the transistor 3 is the voltage E as shown in FIG.
be equivalent to.

As shown in FIG. 2B, when the auxiliary control pulse signal voltage V _G applied from the control circuit 8 to the gate terminal of the thyristor 10 changes from low level to high level and the thyristor 10 is turned on, 2 (G), the second DC power source 2, the resonance reactor 11, the thyristor 10,
A current I _LS starts to flow in the resonance reactor 11 in the path of the smoothing reactor 5 and the load 7, and increases linearly until it becomes equal to the current I of the load 7. On the other hand, the current I _D flowing in the freewheeling diode 4 decreases linearly as shown in FIG. 2D, and when the current I _LS of the resonance reactor 11 becomes equal to the current I of the load 7, the freewheeling diode 4 No current flows through it. When current stops flowing in the free wheeling diode 4,
As shown in FIGS. 2E and 2G, a resonance current flows through the resonance capacitor 9 and the resonance reactor 11. At this time,
The voltage V _CS across the resonance capacitor 9, that is, the voltage V _TR across the transistor 3, drops by a cosine function (E / 2) · (1 + cosωt) due to the resonance action as shown in FIG. 2 (F).
Here, ω is the resonance angular frequency, and t is the elapsed time from the start of resonance. When this voltage V _TR drops to E / 2, the resonance current flowing through the resonance capacitor 9 and the resonance reactor 11 reaches the maximum value as shown in FIGS. 2 (E) and (G).
The current I _LS of the resonance reactor 11 at this time is substantially equal to the sum of the maximum value of the resonance current and the current I of the load 7. afterwards,
The resonance current flowing through the resonance capacitor 9 and the resonance reactor 11 decreases in a cosine function. When the voltage V _TR across the transistor 3 becomes 0 V, the control circuit 8 changes the main control pulse signal voltage V _B applied to the base terminal of the transistor 3 from low level to high level, as shown in FIG. Then, the transistor 3 is turned on from the off state. At this time, the current I _CS of the resonance capacitor 9 is as shown in FIG.
As shown in (E), it almost disappears, and the current I _LS of the resonance reactor 10 becomes substantially equal to the current I of the load 7 as shown in FIG. 2 (G). When the transistor 3 is turned on, FIG.
The current I _TR begins to flow in the transistor 3 as shown in FIG. 3 and increases linearly until it becomes equal to the current I of the load 7. On the other hand, the current I _LS of the resonance reactor 11 flowing in the path of the second DC power supply 2, the resonance reactor 11, the thyristor 10, the smoothing reactor 5 and the load 7 is linear as shown in FIG. When the current I _{TR of the} transistor 3 decreases and the current I _{TR of the} transistor 3 becomes equal to the current I of the load 7, the current I _LS of the resonance reactor 11 disappears. At this time, the control circuit 8 changes the auxiliary control pulse signal voltage V _G applied to the gate terminal of the thyristor 10 from the high level to the low level as shown in FIG. After that, electric power is supplied from the first and second DC power supplies 1 and 2 to the load 7.

As described above, in this embodiment, the current I of the load 7 is changed when the transistor 3 is switched from the on state to the off state.
Is commutated from the transistor 3 to the resonance capacitor 9 to charge the resonance capacitor 9 and linearly increase the voltage V _TR across the transistor 3. Also, the transistor 3
Before turning the off state from the on state, the thyristor 10 is turned on, a resonance current is passed through the resonance capacitor 9 and the resonance reactor 11, and the voltage V _TR across the transistor 3 is changed to a cosine function (E / 2). The transistor 3 is turned on when the voltage V _TR drops to 0V by (1 + cosωt). As described above, since the zero volt switching of the transistor 3 is achieved, the voltage waveform (FIG. 2 (F)) and the current waveform (FIG. 2 (C)) of the transistor 3 are less overlapped, and when the transistor 3 is turned on and turned off. It is possible to reduce the power loss, that is, the switching loss. Further, since the spike-shaped surge voltage and current generated when the transistor 3 is switched on and off is absorbed by the resonance capacitor 9 and the resonance reactor 11, the surge voltage and current during the on / off operation of the transistor 3 are reduced. can do.

The embodiment of the present invention is not limited to the above-mentioned embodiment, and can be modified. For example, in the above-described embodiment, an example in which the junction type power transistor and the reverse blocking three-terminal thyristor (SCR) are used as the main switching element and the auxiliary switching element, respectively, is shown, but the invention is not limited to this combination, and Type power transistor and reverse blocking 3-terminal thyristor (SCR) instead of FET
Other switching elements such as (field effect transistor) may be used.

[0012]

As described above, according to the present invention, the overlap between the voltage waveform and the current waveform of the main switching element is reduced to reduce the power loss when the main switching element is turned on and off, that is, the switching loss. be able to. Further, it is possible to reduce the surge voltage and the surge current during the switching operation of the main switching element.

[Brief description of drawings]

FIG. 1 is an electric circuit diagram of a PWM type DC-DC converter showing an embodiment of the present invention.

FIG. 2 is a waveform diagram showing the voltage and current of each part of the circuit of FIG.

FIG. 3 is an electric circuit diagram showing a conventional PWM type DC-DC converter.

[Explanation of symbols]

1. ． ． First DC power supply, 2. ． ． Second DC power supply,
3. ． ． Transistor (main switching element),
4. ． ． Reflux diode (recirculation rectifier),
5. ． ． Smooth reactor, 6. ． ． Smoothing capacitor,
7. ． ． Load, 8. ． ． Control circuit, 9. ． ． Resonant capacitor, 10. ． ． Thyristor (auxiliary switching element), 11. ． ． Resonance reactor

Claims (1)

[Claims] 1. A first and a second DC power supply connected in series, a main switching element having one main terminal connected to the first DC power supply, and a main switching element connected in parallel. A resonance capacitor, a series circuit of an auxiliary switching element and a resonance reactor connected between a connection point of the first and second DC power supplies and the other main terminal of the main switching element, and the main switching element Of the free-wheeling rectifier connected between the other main terminal of the free-flowing rectifier and the second DC power supply, a smoothing circuit including a reactor and a capacitor connected to both ends of the free-wheeling rectifier, and a smoothing circuit connected to the smoothing circuit. Auxiliary load control pulse to the control terminal of the auxiliary switching element before detecting the output voltage of the smoothing circuit and applying the main control pulse signal to the control terminal of the main switching element. A PWM type DC-DC converter, comprising: a control circuit for giving a signal. [0001]