JP3097886B2 - Step-up chopper type switching power supply - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a step-up chopper type switching power supply, and more particularly to a step-up chopper type switching power supply capable of reducing switching loss.

[0002]

2. Description of the Related Art In recent years, downsizing of electronic equipment has been strictly required, and downsizing of a switching power supply, which is a power supply device used for the electronic equipment, has also been strongly demanded. In order to reduce the size of the switching power supply, the switching frequency is generally increased by increasing the switching frequency.However, when the switching frequency is increased, the switching loss of the main switching element increases and the amount of heat generated by the main switching element increases. The size has increased, which has been an obstacle to downsizing. For this reason, the downsizing of the switching power supply is an important factor not only in increasing the frequency but also in increasing the efficiency. For example, a first reactor and a first rectifier are connected in series between one end of a DC power supply and a load, a first capacitor is connected in parallel with the load, and the first reactor and the first rectifier are connected. Connect a main switching element between the connection point of the element and the other end of the DC power supply, and turn on and off the main switching element to supply a DC output with a constant voltage higher than the DC power supply voltage to the load. Chopper type switching power supplies have been widely used as comparatively small switching power supplies.

[0003]

However, in the above-mentioned step-up chopper type switching power supply, a current waveform and a voltage waveform overlap each other during the on-conversion period and the off-conversion period of the main switching element, and a switching loss based on this occurs. was there. In addition, since this switching loss becomes Joule heat and the amount of heat generated by the main switching element increases, the size of the radiating fins and the like becomes large, and there is a disadvantage that it is difficult to reduce the size of the entire device.

Accordingly, an object of the present invention is to provide a boost chopper type switching power supply capable of reducing switching loss.

[0005]

A step-up chopper type switching power supply according to the present invention comprises a DC power supply (1) and a load (2).
A first reactor (4) and a first rectifying element (5) are connected in series between the first and second capacitors, and a capacitor (6) is connected in parallel with the load (2). A main switching element (3) between a connection point of the first reactor (4) and the first rectifying element (5) and the other end of the DC power supply (1);
And supplies a DC output of a constant voltage higher than the voltage of the DC power supply (1) to the load (2) by controlling the ON / OFF of the main switching element (3). The second reactor (16)
A second rectifying element (18) and an auxiliary switching element (15)
Is connected in parallel with the main switching element (3), and the connection point between the second reactor (16) and the second rectifying element (18) is connected to the first rectifying element (5) and the load (2). A third rectifying element (17) is inserted between the connection points (1) and (2). The main control pulse signal is applied to the control terminal of the main switching element (3) according to the terminal voltage of the load (2), and the auxiliary control is applied to the control terminal of the auxiliary switching element (15) before applying the main control pulse signal. Apply a pulse signal. In the above embodiment of the present invention, the second
A series circuit of the reactor (16), the second rectifier (18) and the auxiliary switching element (15) is connected in parallel with the main switching element (3), and the second reactor (16) and the second rectifier are connected. A third rectifier element (17) is inserted between the connection point of the element (18) and the connection point of the first rectifier element (5) and the load (2). In another embodiment of the present invention, A main switching element includes a series circuit of a second reactor (16), a second rectifying element (18), an auxiliary switching element (15), and a secondary winding (42) of the first reactor (4). Connect in parallel with (3).

[0006]

The auxiliary switching element is turned on by applying an auxiliary control pulse signal to the control terminal of the auxiliary switching element before applying the main control pulse signal to the control terminal of the main switching element. The current flowing in the series circuit of the element and the second reactor gradually increases from 0, and when the current becomes equal to the current flowing in the first reactor, the voltage applied to the main switching element gradually decreases. Then, when the voltage becomes 0 V, a main control pulse signal is applied to the control terminal of the main switching element to turn on the main switching element, whereby switching loss at the time of turning on the switching element can be reduced. When the secondary winding of the first reactor is inserted, the auxiliary switching element is turned off by the current, and the switching loss can be further reduced.

[0007]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS An embodiment of a boost chopper type switching power supply according to the present invention will be described below with reference to FIGS. 1 and 2 and FIGS. 6 and 7. FIG. In the step-up chopper type switching power supply of the present embodiment, as shown in FIG. 1, a first reactor 4 and a diode 5 as a first rectifier are connected in series between one end of a DC power supply 1 and a load 2. Have been. Reactor 4
An N-channel MOSFET as a main switching element is provided between the connection point of the diode 5 and the other end of the DC power supply 1.
T3 is connected. The MOSFET 3 is equivalently composed of a switching element body 12, an internal diode 13 connected in anti-parallel between the source and drain terminals of the switching element body 12, and an internal capacitor 14 connected in parallel to the internal diode 13. Is done. Built-in diode 13 and built-in capacitor 14 are MOSFET
3 is a parasitic diode and a parasitic capacitance between the source and drain terminals. A capacitor 6 is connected in parallel with the load 2. Between the source-drain terminals of MOSFET3,
Second reactor 16, diode 18 as a second rectifying element, and N-channel M as an auxiliary switching element
A series circuit with the OSFET 15 is connected. The diode 18 is a diode for preventing backflow. MOSF
The ET 15 is equivalently composed of the switching element main body 19 and the built-in diode 20 and has a parasitic capacitance between the source and drain terminals like the MOSFET 3.
It is used for a shorter time than the SFET 3 and uses a MOSFET having a small parasitic capacitance, so that the description is omitted here. The reactor 16 serves to moderate the increase in current during the on-turning period of the switching element body 19 constituting the MOSFET 15. A diode 17 as a third rectifying element is connected between a connection point between the reactor 16 and the diode 18 and a connection point between the diode 5 and the load 2. The diode 17 is for releasing the energy of the reactor 16 to the load 2 during the turning-off period of the switching element body 19 constituting the MOSFET 15. Further, both ends of the load 2 and the MOSFET 3
The terminal voltage of the load 2 is detected between the gate terminal of
The main control pulse signal is applied to the gate terminal of the OSFET 3 and the MOSFE is applied before the main control pulse signal is applied.
A control circuit 9 for applying an auxiliary control pulse signal is connected to the gate terminal of T15.

As shown in FIG. 6, the details of the control circuit 9 are: a voltage detection circuit 21 connected to output terminals 7 and 8 of a power supply;
PWM pulse forming circuit 25 including error amplifier 22, reference voltage source 23, PWM (pulse width modulation) control circuit 24, etc.
, A delay circuit 26, an AND gate 27, a monostable multivibrator 29, first and second drive circuits 28,
30. The voltage detecting circuit 21 is composed of a voltage dividing circuit.
2 inverting input terminals. Error amplifier 22
The reference voltage source 23 is connected to the non-inverting input terminal, and outputs a signal corresponding to the difference between the reference voltage of the reference voltage source 23 and the detection voltage of the voltage detection circuit 21. The PWM control circuit 24 connected to the output terminal of the error amplifier 22 includes a triangular wave generator and a voltage comparator, and generates a square wave having a constant cycle by the voltage comparator. In this embodiment, a PWM control IC (integrated circuit) is used as the PWM control circuit 24. For example, a commercially available MB3759, μPC494, or the like can be used. One input terminal of the AND gate 27 is directly connected to the PWM control circuit 24, and the other input terminal of the AND gate 27 is connected via the delay circuit 26 to the PWM control circuit 24.
It is connected to the. The monostable multivibrator 29
It is directly connected to the PWM control circuit 24. The AND gate 27 and the monostable multivibrator 29 are connected to the first and second driving circuits 28 and 30, respectively.
The ET control lines 10 and 11 are connected. The first and second FET control lines 10, 11 are MOSFETs, respectively.
3 and the gate terminal of the MOSFET 15.

FIG. 7 shows voltage waveforms at points A, B and C in FIG.
(A), (B) and (C) show. 7 from the PWM control circuit 24
A square wave pulse (PWM pulse) shown in FIG. 7A is repeatedly generated at a period T (point A in FIG. 6). When the output voltage of the power supply becomes higher than the reference value, the pulse width becomes narrow. This is the same as the operation of a general PWM control switching power supply. Since the pulse and delayed pulse delay time T ₂ for the pulse of Figure 7 to the AND gate 27 (A) is inputted, the main control pulse signal shown in FIG. 7 (B) from the output terminal of the AND gate 27 is output (Point B in FIG. 6). On the other hand, the pulse shown in FIG. 7A is also input to the monostable multivibrator 29, and the auxiliary control pulse signal shown in FIG. 7C is output from the monostable multivibrator 29 (point C in FIG. 6).
This auxiliary control pulse signal has a period T having a certain time T _1.
Pulse signal. The main control pulse signal and the auxiliary control pulse signal are applied to the gate terminals of the MOSFETs 3 and 15 via the first and second drive circuits 28 and 30, respectively. Therefore, the load 2 is controlled by the control circuit 9 having the above configuration.
Before applying the main control pulse signal to the gate terminal of the MOSFET 3 by detecting the terminal voltage of the MOSFET 3, the auxiliary control pulse signal can be applied to the gate terminal of the MOSFET 15.

In the above configuration, as shown in FIG. 2B, at t ₀ , an auxiliary control pulse signal is applied from the control circuit 9 to the gate terminal of the MOSFET 15, and the auxiliary control pulse signal voltage V _{When G2} goes from a low level to a high level, the switching element body 1
9 turns on. At this time, current I _Q2 flowing through the switching device body unit 19, gradually from 0 in FIG inclination related to the inductance L ₂ of the output voltage V _OUT and the reactor 16 as shown in _{(E) (V OUT / L} 2) Going to
The current I _L1 flowing through the reactor 4 is gradually shared. On the other hand, the current _ID5 flowing through the diode 5 gradually decreases to 0 as shown in FIG. That is, reactor 4
Current _IL1 flowing through the reactor 16 and the diode 18
And gradually flows along the path of the switching element body 19. Therefore, current I _Q shown in the figures and the current I _Q2 through the switching device body unit 19 shown in FIG. 2 (E) 2 (G)
(I _Q = I _Q1 + I _Q2 ) is the current I _L1 flowing through the reactor 4
Going to be equal to. The switching element body 1
Since the voltage V _Q2 applied to the switching element 9 quickly drops to 0 V as shown in FIG. 2D, the switching of the switching element body 19 during the on-turning period is performed by the zero current switching in which the overlap between the voltage waveform and the current waveform is small. (ZCS).

[0011] As shown in FIG. 2 (E), MO in t ₁
When the current I _Q2 flowing through the switching element main body section 19 of SFET15 reaches the current I _L1 flowing through the reactor 4 o'clock t _0, it is substantially constant. At this time, the charge of the built-in capacitor 14 in the MOSFET 3 starts to be discharged, and the voltage V _Q1 applied to the switching element body 12 as shown in FIG.
Gradually falls to 0V.

As shown in FIG. 2A, at t ₂ , a main control pulse signal is applied from the control circuit 9 to the gate terminal of the MOSFET 3, and the main control pulse signal voltage V _{G1 of} the switching element body 12 changes from a low level. At a high level,
The switching element body 12 is turned on. At this time, since the voltage V _Q1 applied to the switching element body 12 is 0 V as shown in FIG. 2C, the switching element body 12 is turned on at 0 V. Therefore, zero-voltage switching with almost no switching loss can be realized during the on-turning period of the switching element body 12.
At this time, since the switching element main body 19 is in the ON state, almost all of the current I _L1 flowing through the reactor 4 flows through the switching element main body 19, and as shown in FIG. I _Q1 hardly flows.

As shown in FIG. 2B, at t ₃ , the MO
When the auxiliary control pulse signal voltage V _G2 of the switching element body 19 of the SFET 15 changes from the high level to the low level,
The switching element body 19 is turned off. At this time, as shown in FIG.
Current I _Q1 begins to flow to the switching element main body 12 of MOSFET3 as shown in the same time the current I _Q2 does not flow FIG 2 (F) to. At the same time, the reactor 16 starts to release the stored energy, so that the current flowing through the reactor 16 is reduced as shown in FIG.
7 flows to the load 2 (I _D17 ). Diode 17
The current I _D17 flowing through, gradually decreases with a slope related to the inductance L ₂ of approximately the output voltage V _OUT and the reactor 16 as shown in FIG. _{2 (I) (-V OUT /} L 2), t 4 Becomes 0 at. At this time, the current flowing through the reactor 16 disappears.

As shown in FIG. 2A, M at t ₅
When the main control pulse signal voltage V _G1 of the switching element body 12 of the OSFET 3 changes from a high level to a low level, the switching element body 12 is turned off. At this time, the energy stored in the reactor 4 is
Is supplied to the load 2 and the capacitor 6 via the.

As described above, in this embodiment, the MOSFET
Since the switching element body 12 of No. 3 is turned on at 0 V, it is possible to reduce the switching loss at the time of turning on the switching element body 12 (at the time of turn-on). In addition, t ₂ and t ₃ is not may be performed at the same time.
In addition, the period of t ₀ ~t ₄ is short enough to almost negligible compared to the period of t ₀ ~t _5.

Next, another embodiment of the step-up chopper type switching power supply according to the present invention will be described with reference to FIGS. However, in FIG. 3, the same portions as those in FIG. 1 are denoted by the same reference numerals, and description thereof will be omitted. The details of the control circuit 9 in FIG. 3 are exactly the same as those in FIGS. 6 and 7 shown in the embodiment of FIG. The circuit of the embodiment of FIG. 3 includes a reactor 4 on the cathode terminal side of the diode 18.
Secondary winding 4 electromagnetically coupled to the primary winding 41 of the
2 is inserted. Therefore, when the switching element body 12 in the MOSFET 3 is off, a negative voltage is applied to the cathode terminal of the diode 18, and when the switching element body 12 in the MOSFET 3 is on, the diode 18 A positive voltage is applied to the cathode terminal.

[0017] In the above configuration, as shown in FIG. 4 (B), the auxiliary control pulse signal voltage V _G2 of the switching element main body section 19 of the MOSFET15 at t ₀ is composed of a low level to a high level, the switching element main body section 19 turns on. At this time, the switching element body 1
Current I _Q2 flowing through 9, as shown in FIG. 4 (E), 2 winding 4 of the output voltage V _OUT and a reactor 4 shown in FIG. 4 (H)
2 induced voltage V _{N 2} = −V ₁ (V _OUT + V
₁ ) and the gradient [(V _OUT + V ₁ ) / L ₂ ] related to the inductance L _{2 of the} reactor 10 gradually increases from 0V.
At the same time, the voltage V _Q2 applied to the switching element body 19 in the MOSFET 15 becomes 0 as shown in FIG.
Drop to V.

As shown in FIG. 4 (E), at t ₁ the MO
The current I _Q2 flowing through the switching element body 19 in the SFET 15 is the current I _Q2 flowing through the primary winding 41 of the reactor 4.
_When it becomes equal to _L41 , the charge of the built-in capacitor 14 in the MOSFET 3 starts to be discharged, and as shown in FIG.
The voltage V _Q1 applied to the switching element body 12 in the OSFET 3 gradually decreases to 0V.

As shown in FIG. 4A, at t ₂ , the MO
When the main control pulse signal voltage V _G1 of the switching element main body 12 in the SFET3 is made of a low level to a high level, the switching element main body section 12 is turned on. At this time, as shown in FIG. 4C, the voltage V _Q1 applied to the switching element body 12 in the MOSFET 3 is 0 V, so that switching without any switching loss can be realized in the ON transition period. At the same time, one of reactor 4
The current I _L41 of the next winding 41 starts to flow through the switching element body 12 in the MOSFET 3 and linearly increases as shown in FIG. 4 (F) (I _Q1 ). On the other hand, the secondary winding 42 of the reactor 4 has a MOSFET as shown in FIG.
Switching element main body 12 in the 3 applied voltage V _N2 = + V ₂ of the opposite polarity when off, the current flowing through the reactor 16, i.e. the current I _Q2 flowing through the switching element main body 19 in the MOSFET15 are 4 It decreases at a slope of -V ₂ / L ₂ as shown in (E).

As shown in FIG. 4E, at t ₃ , the MO
Current I _Q2 flowing through the switching element main body 19 in the SFET15 is zero. At this time, it auxiliary control pulse signal voltage V _G2 of the switching element main body section 19 of the MOSFET15 as shown in FIG. 4 (B) is from the high level to the low level, the switching element main body section 19 is turned off. Therefore, even during the off-turning period of the switching element body 19 in the MOSFET 15, zero current switching with small switching loss can be realized.

As shown in FIG. 4A, at t ₄ , the MO
When the main control pulse signal voltage V _G1 of the switching element body 12 in the SFET 3 changes from the high level to the low level, and the switching element body 12 is turned off, the energy stored in the reactor 4 is transferred via the diode 5 to the load 2. And the capacitor 6.

As described above, in the embodiment shown in FIG. 3, the same effect as in the embodiment shown in FIG. 1 can be obtained with respect to the switching loss. Further, in the embodiment shown in FIG.
Since the switching loss of the switching element body 19 in the off-turning period is small, the effect is greater than that of the embodiment shown in FIG.

The embodiment of the present invention is not limited to the above embodiment, and various modifications are possible. For example, the following (a)-
(e) is a part of the modification. (A) Built-in diode 1 in MOSFETs 3 and 15
Independent diodes may be used instead of the built-in diodes 3 and 20. (B) The internal capacitor 14 in the MOSFET 3 is
An independent capacitor can be connected without using the parasitic capacitance of the SFET. (C) A bipolar transistor, a thyristor, or the like may be used as the main switching element and the auxiliary switching element without using MOSFETs. It should be noted that an element that does not include a diode having a reverse polarity, such as a bipolar transistor or a thyristor, does not need to include the diode 18 for preventing backflow. (D) The circuit of the first embodiment may be modified to the circuit shown in FIG. (E) The reactor 16 in the embodiment of FIG. 3 may use the leakage inductance of the reactor 4.

[0024]

As described above, according to the present invention, since zero voltage switching of the main switching element can be easily achieved, the overlap between the voltage waveform and the current waveform of the main switching element is reduced, and It is possible to reduce the power loss at the on-turning period, that is, the switching loss at the time of turning on the main switching element. For this reason,
By reducing the heat generation of the main switching element, the size of the radiating fins etc. can be reduced, and a high-frequency, small boost chopper type switching power supply can be realized.

[Brief description of the drawings]

FIG. 1 is an electric circuit diagram of a boost chopper type switching power supply showing an embodiment of the present invention.

FIG. 2 is a waveform chart showing voltages and currents of respective parts of the circuit of FIG.

FIG. 3 is an electric circuit diagram of a step-up chopper type switching power supply showing another embodiment of the present invention.

FIG. 4 is a waveform chart showing the voltage and current of each part of the circuit of FIG.

FIG. 5 is an electric circuit diagram showing a modification of the circuit of FIG. 1;

FIG. 6 is a block diagram showing details of a control circuit shown in FIGS. 1, 3 and 5;

FIG. 7 is a waveform chart showing voltages of respective parts of the circuit of FIG. 6;

[Explanation of symbols]

1. . . DC power supply, 2. . . Load, 3,15. . . N-channel MOSFET, 4, 16,. . . Reactor 5,
17, 18. . . 5. diode, . . Capacitors,
9. . . Control circuit, 41. . . Primary winding, 42. . . 2
Next winding

Claims (2)

(57) [Claims] 1. A first reactor and a first rectifier element are connected in series between one end of a DC power supply and a load, a capacitor is connected in parallel with the load, and the first reactor and the first rectifier are connected in parallel. A dc output of a constant voltage higher than the voltage of the dc power supply by connecting a main switching element between a connection point of the rectifying element and the other end of the dc power supply and controlling on / off of the main switching element. And a series circuit of a second reactor, a second rectifier, and an auxiliary switching element is connected in parallel with the main switching element, and the second reactor and A third rectifier element is inserted between a connection point of a second rectifier element and a connection point of the first rectifier element and the load, and the main switching element is connected to a terminal voltage of the load. To the control terminal as well as imparting a main control pulse signal, the step-up chopper type switching power source, which comprises applying an auxiliary control pulse signal to a control terminal of the auxiliary switching element before applying the main control pulse signal. 2. A first reactor and a first rectifying element are connected in series between one end of a DC power supply and a load, a capacitor is connected in parallel with the load, and the first reactor and the first reactor are connected in parallel. A dc output of a constant voltage higher than the voltage of the dc power supply by connecting a main switching element between a connection point of the rectifying element and the other end of the dc power supply and controlling on / off of the main switching element. A step-up chopper-type switching power supply that supplies a series circuit of a second reactor, a second rectifier, an auxiliary switching element, and a secondary winding of the first reactor to the main switching element. Connected in parallel with the main control pulse signal to the control terminal of the main switching element according to the terminal voltage of the load, and before applying the main control pulse signal, Boost chopper type switching power source, which comprises applying an auxiliary control pulse signal to the control terminal of the auxiliary switching element.