JPH08136596A - Phase-difference measuring apparatus - Google Patents

Abstract

PURPOSE: To obtain a phase-difference measuring apparatus by which the phase difference (the very small time difference) between two signals at an identical frequency can be measured with high accuracy by reducing the required number of bits for an A/D converter. CONSTITUTION: In addition to respective outputs from an upstream-side pickup 1A and a downstream-side pickup 1B for a flowmeter, outputs from an addition circuit 2 which has added the outputs are quantized by a comparator 3, a PLL circuit 4, antialiasing filters 5A to 5C, sample-and-hold circuits 6A to 6C and A/D converters 7A to 7C. Then, only desired signals are extracted by band-pass filters 8A to 8C so as to be complex-Fourier-transformed by DFTs 9A to 9C. Thereby, the phase difference between two signals at an identical frequency such as an upstream-side pickup signal and a downstream-side pickup signal can be detected with high accuracy by reducing the required number of bits for the A/D converters.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a phase difference measuring device for measuring the phase difference between two signals having the same frequency. This type of phase difference measuring device is applicable to various fields,
As a typical example thereof, industrial equipment such as a Coriolis mass flowmeter for detecting the flow rate by detecting the phase difference of the vibration of the pipe due to the mass and speed of the fluid on the upstream side and the downstream side of the pipe generated by the Coriolis force There is an instrument. Hereinafter, an example of the Coriolis mass flowmeter will be described.

[0002]

2. Description of the Related Art FIG. 5 shows a principle block diagram of a Coriolis mass flowmeter. That is, 21 is a U-shaped pipe through which the measurement fluid flows, the permanent magnets 22 are fixed to the tip end thereof, and both ends of the U-shaped pipe 21 are fixed to the base 23. Reference numeral 24 denotes an electromagnetic drive coil provided so as to sandwich the U-shaped pipe 21.
A support frame 25 holds the electromagnetic drive coil 24, and the frame 25 is firmly fixed to the base 23. Like the tuning fork, the U-shaped pipe 21 has a configuration in which the base 23 serves as a node for vibration, and the vibration energy is less lost. 1A and 1B are electromagnetic pickups for detecting the displacement of both legs of the U-shaped pipe. When the U-shaped pipe 21 is vibrated (sin ωt) at its natural frequency by the electromagnetic force acting between the drive coil 24 and the permanent magnet 22 fixed to the U-shaped pipe 21 facing the drive coil 24, the U-shaped pipe 21 flows in the U-shaped pipe. Coriolis force is generated in the fluid.

FIG. 6 shows how the U-shaped pipe vibrates. The magnitude of this Coriolis force is proportional to the mass of the fluid flowing in the U-shaped pipe and its velocity, and the direction of the force is the direction of movement of the fluid.
It coincides with the direction of the vector product of the angular velocities that excite the U-shaped pipe 21. In addition, since the flow directions of the fluid flow on the input side and the output side of the U-shaped pipe 21 are opposite, a twisting torque is generated in the U-shaped pipe 21 due to the Coriolis force on both leg sides. This torque changes at the same frequency as the excitation frequency,
Its amplitude value is proportional to the mass flow rate of the fluid. FIG. 7 shows a vibration mode generated by this torsion torque.

The mass flow rate can be known by detecting the amplitude of the torque of this torsional vibration with the pickups 1A and 1B, but the actual vibration of the U-shaped pipe is Coriolis to the vibration excited by the electromagnetic drive coil 24. The torsional vibration due to the force is superimposed, and the upstream side is sin (ωt-α),
The downstream side is expressed in the form of sin (ωt + α). Therefore, the signals e1, detected by the pickups 1A, 1B
e2 is a waveform having a phase difference (Δt) as shown in FIG. This phase difference varies depending on the pipe and the excitation frequency. For example, in the case of the U-shaped pipe 21, if the resonance frequency of the U-shaped pipe is 80 Hz, a phase difference of about 120 μS occurs at the maximum flow rate, and the maximum phase difference is 0.01 % Resolution must be compensated. Therefore, a time measurement resolution of 12 nS is required.

There are various methods for this phase measurement, but the simplest method is the method of counting the time difference gate by the reference clock. An example thereof is shown in FIG. That is, the upstream pickup signal Pu and the downstream pickup signal Pd are amplified (amplification factor: B) by the amplifier 31 and then binarized by the comparator 32, and the exclusive OR circuit 33 performs an exclusive OR operation of the binarized signals. The gate pulse Pg having a pulse width corresponding to the time difference between the upstream pickup signal and the downstream pickup signal is obtained, and this is measured by the counter 34 with the reference clock 35. The frequency of the reference clock in this case needs to be about 85 MHz or higher.

By the way, when the U-shaped pipe is used in an actual plant, there is a problem that the pressure loss is large because it is bent and it is difficult to clean the pipe. Therefore, a straight pipe type Coriolis flowmeter using a straight pipe has also been proposed. Figure 10
Shows an example of a straight tube type Coriolis flowmeter. In FIG. 10, reference numeral 41 denotes a straight pipe through which the measurement fluid flows, a permanent magnet 43 is fixed to the central portion thereof, and both ends of the straight pipe 41 are fixed to the base 40. Reference numeral 42 denotes an electromagnetic drive coil provided so as to sandwich the straight pipe 41, 44 denotes a support frame for holding the electromagnetic drive coil 42, and this frame is firmly fixed to the base 40. In the straight pipe system, the pipe through which the fluid passes has a high rigidity and is more difficult to bend than the U-shaped pipe, so that there is a problem in that the time difference is small.

For example, the resonance frequency of a straight pipe is about 1 KHz, a phase difference of about 2 μS occurs at the maximum flow rate, and it is necessary to measure with a resolution of 0.01% of this maximum phase difference. Therefore, a time measurement resolution of 0.2 nS is required.
In addition, the measurement by the counter requires a reference clock of 5 GHz and cannot be manufactured in practice, and even if it is possible, if a comparator is used to obtain the time difference signal from the pickup signal, this will cause a dead zone of the input signal. Jitter occurs due to a problem (a halfway level where the output of the comparator is not "1" or "0" is called a dead zone, and how quickly the input signal crosses this dead zone greatly affects), and 0.2 nS It is doubtful whether accuracy can be obtained.

Therefore, conventionally, the structure shown in FIG. 11 is used to perform the measurement, and the upstream pickup signal Pu and the downstream pickup signal Pd are subtracted, that is, sin (ωt + α) -sin (ωt-α) = 2cosω
The calculation of t * sinα is performed by the differencer (subtractor) 51, and sinα
Is obtained (a period is 1 mS and a phase α is 0.1 nS), and the amplifier 52 highly amplifies this (amplification factor: B).
After extracting only the desired frequency component by 3 and obtaining a DC level of Bsinα by the full-wave rectifier / detector 54, the A / D
The converter 55 calculates the digital value of sin α, and the phase difference α is calculated from this. However, in this method, the offset current of the amplifier and the temperature drift of the voltage are received, and the error of the indicated value becomes large especially in a plant with bad temperature environment, and the accuracy of 0.01% with respect to the maximum flow rate cannot be obtained.

Therefore, in order to make the amplifier less susceptible to the temperature drift, a method of obtaining the phase difference by the complex Fourier transform of digital signal processing has been proposed. 12
FIG. 3 is a block diagram showing this type of system. That is, the upstream and downstream pickup signals Pu and Pd are fed to the amplifier 61.
The sample and hold circuit 62 amplifies the sampled signal and the sampled level is digitized by the A / D converter 63 and stored in the data memory 64. The digital signal processor (DSP) 11 performs digital filtering and frequency analysis for removing noise components on a data string of discretized digital values stored in the data memory 64, and a real part and an imaginary part at each frequency. Perform a complex Fourier transform to find the component of the part,
The phase due to the real part and the imaginary part at the excitation frequency in the electromagnetic coil can be obtained.

The principle of phase measurement by the complex Fourier transform will be described below. First, the time function f (t)
Let F (U) be the Fourier transform of F (U).

[Equation 1]

Next, if the Fourier transform of the time function f (t + a) obtained by shifting the time t of the above-mentioned expression 1 by a is F (U '), this is shown as the following expression 2.

[Equation 2]

FIG. 13 shows the relationship between F (U) and F (U '). From the above, it can be seen that the phase advance on the time axis is the phase advance in the frequency domain. Therefore, if F (U) = A + jB, then F (U ') = F (U) e ^jUa = (A + jB) (cosU
a + jsinUa), and the phase can be obtained from the vector components of both.

By the way, in the above phase calculation, the resolution of the vector component affects the resolution of the phase, so that a high-bit A / D converter is required for the pickup signal. The detection phase sensitivity depends on the number of bits of the A / D converter. For example, when the excitation frequency in the case of a straight pipe is 1 KHz and the minimum phase detection is 0.1 nS, A
It has been pointed out that the required number of bits of the / D converter is 24.

Recently, a 24-bit A / D converter is commercially available, but the conversion speed is 16 mS and 1 KHz.
Is too slow to sample the pickup signal of, and the level at the minimum bit is 0.6 μV when the full scale of the A / D converter is 10 V, which is extremely low in a plant with a poor noise environment. It cannot be reduced, and it is actually impossible to measure the phase difference with a resolution of 0.1 nS.

[0015]

As described above in detail, in the straight pipe type Coriolis flowmeter which has a small pressure loss and is easy to clean, in order to reduce the influence of temperature drift, the phase difference is changed by the complex Fourier transform of the digital signal processing. However, the practical detection resolution is low (about 5 nS), and
You will have the problem that the accuracy of% cannot be guaranteed. Therefore, an object of the present invention is to improve the detection resolution.

[0016]

In order to solve such a problem, in the invention of claim 1, an arithmetic means for obtaining the sum or difference of two signals of the same frequency, an output from this arithmetic means and the above-mentioned 2 Quantizing means for quantizing each of the two signals, band-pass filter means for extracting only a predetermined frequency component from the quantized signal, and Fourier transform of the output signal to perform a predetermined calculation to obtain the basic frequency of the signal. A phase difference calculating means for calculating the phase difference is provided.

The quantizing means includes a comparator for binarizing the amplitude of one of the two signals, a PLL circuit for generating a frequency signal which is an integral multiple of the output of the comparator, and the output of the PLL circuit as timing. It is characterized in that it comprises an output from the arithmetic means and an A / D conversion means for quantizing the two signals (the invention of claim 2). Further, in the invention of the above-mentioned claim 1, the calculation means for obtaining the sum or difference of the two signals may be provided with a gain switching function so that the detection range and the resolution can be switched according to the phase difference to be detected. It is possible (the invention of claim 3).

According to the invention of claim 1 or 2,
The quantizing means may include sample and hold means for sampling and holding the two signals and the outputs from the arithmetic means, respectively (invention of claim 4), or at least one of the two signals. Amplitude changing means for changing the amplitude and correction data for obtaining the correction data when the amplitudes of the two signals are changed, and the correction means for correcting the phase difference based on the data can be added (the invention of claim 5). ) Or a signal amplitude difference detecting means for detecting an amplitude difference between the two signals, and a gain control for matching the amplitude of one of the two signals with the amplitude of the other signal based on the detected amplitude difference. An amplifier can be added (the invention of claim 6).

[0019]

By using the amplitudes and phases of three types of signals, which are signals obtained by adding or subtracting these signals, in addition to two signals having the same frequency, the conventional method using only the above two signals can be used. The number of required bits of the A / D converter is reduced so that the phase difference between the two signals can be detected with high accuracy (accuracy of 0.01% = 0.2 nS resolution, A / D The required number of bits of the converter is 14 to 16 bits). Further, in particular, by using a comparator that binarizes the signal amplitudes of two signals of the same frequency and a PLL circuit that generates a frequency signal that is an integral multiple of its output, the constants in the program used in the digital filter calculation It is possible to change the cutoff frequency of the digital filter according to the change of the signal frequency without changing the, to improve the accuracy.

Further, the calculating means for obtaining the sum or difference of the two signals is provided with a gain switching function so that the detection range and the resolution can be switched according to the phase difference to be detected, and the quantizing means. In addition, it is possible to faithfully reproduce the original signal by including sample-hold means for sampling and holding the two signals and the outputs from the arithmetic means. In addition, whether to add amplitude varying means for changing the amplitudes of the two signals and correction means for obtaining correction data when the pair of signal amplitudes changes and correcting the phase difference based on this data , Or 2
By adding a signal amplitude difference detection means for detecting the amplitude difference between two signals and a gain control amplifier for matching the signal amplitudes of the pair of signals based on the detected amplitude difference, the amplitude difference between the two signals is detected. Be prepared to deal with the occurrence of.

[0021]

1 is a block diagram showing an embodiment of the present invention. In the figure, 1A and 1B are upstream and downstream electromagnetic pickups, 2 is an adder circuit, 3 is a comparator, 4
Is a phase-locked loop (Phase Lock Loop:
PLL), 5A, 5B, 5C are anti-alias filters, 6A, 6B, 6C are sample and hold circuits, 7A,
7B and 7C are A / D converters, 8A, 8B and 8C are bandpass filters, 9A, 9B and 9C are discrete Fourier transfer (DFT), and 10 is a phase calculation unit. The bandpass filters 8A, 8B, 8C,
The DSP 11 is configured by the DFTs 9A, 9B, 9C, the phase calculator 10, and the like.

From the upstream electromagnetic pickup 1A, A
A signal of sin (ωt−α) and a signal of −B sin (ωt + α) are obtained from the downstream electromagnetic pickup 1B. Therefore, when the adder circuit 2 as shown is used, the downstream pickup signal has a flow rate of 0.
When .alpha. = 0, the wirings of the upstream and downstream electromagnetic pickups are reversed so that there is a 180-degree phase difference with the upstream pickup signal. When using the subtraction circuit,
It is not necessary to reverse the upstream and downstream electromagnetic pickup wiring.

The output of the adder circuit 2 is: Asin (ωt-α) -Bsin (ωt + α) =-2Asinαcosωt + (AB) sin (ωt + α) (1) At this time, the adder circuit 2 having a gain switching function
The output of is k {-2Asin α cosωt + (A−B) sin (ω
t + α)}, but by making this constant k adjustable,
The detection resolution in the A / D converter 7B can be changed, and the phase difference detection resolution can also be changed.

Now, assuming that the upstream pickup signal is the A vector, the downstream pickup signal is the B vector, and the adder circuit output is the C vector, these vectors are as shown in FIG. When the flow rate is 0 (α = 0), the delay of the circuit system,
The A vector does not match the x-axis (sinωt) direction in FIG. 2 due to the start position of the waveform used for the complex Fourier transform.

When the shift angle is θd, the angle θd can be expressed by the following equation from the angles θa and θb from the circuit reference delay axis of the A vector and the B vector. (Θa−θd) + (θb−θd) = π θd = (θa + θb−π) / 2 (2) Therefore, the angle θe of the C vector with the x axis is the angle θ.
It is calculated from a, θb, and θc by the following equation. θe = θc−θd = θc− (θa + θb−π) / 2 (3)

On the other hand, in FIG. 2, when the equation for the y-axis component of the C vector is established, the following equation is obtained. Csinθe = 2Asinα- (AB) sinα (4) Therefore, the phase difference α between the upstream side and the downstream side is α = sin ^-1 [Csin {θc- (θa + θb-π) / 2} / (A + B)]. … (5)

The equation (5) is expressed by α, which is proportional to the mass flow rate, from the amplitude and phase of the A vector, B vector and C vector.
It shows that is required. That is, since the sine function of the equation (5) is "1" in the normal state, in that case, α can be simply obtained from the magnitude (amplitude) of each vector, and from this, only the A vector and the B vector are obtained. Therefore, the phase difference can be obtained relatively easily as compared with the conventional method, and the number of bits of the A / D converter can be reduced.

In obtaining the amplitude and phase of the above three signals, each signal is a kind of low-pass filter (LPF) 5A, 5B called an anti-alias filter in this embodiment.
5C is applied to remove noise components, and the output is sampled and held by the sample and hold circuits 6A, 6B and 6C. Comparator 3 makes the timing at this time
Further, the PLL 4 is adapted to perform sampling at a timing of, for example, 8 times the fundamental frequency here.

This is based on the principle that in order to reproduce the original signal by sampling, it is sufficient to sample at a frequency twice or more that of the original signal. Therefore, the PLL 4 in this case functions as a frequency multiplication circuit. The sampling in this case is performed with high bits (16 bits) in order to improve the accuracy of the phase difference to be detected. Although the comparator 3 and the PLL 4 are provided on the electromagnetic pickup 1B side in this example, it goes without saying that they may be provided on the electromagnetic pickup 1A side.

The outputs of the sample and hold circuits 6A, 6B and 6C are digitized by the A / D converters 7A, 7B and 7C and then input to the DSP 11, where the band pass filter 8 is provided.
Only the fundamental frequency component is extracted at A, 8B, and 8C, and DF
The amplitude and phase of the fundamental frequency component are obtained at T9A, 9B and 9C. The DFT is provided to obtain the amplitude and phase of a desired frequency component at high speed, and an FFT (Fast Fourier Transform) may be used to obtain all frequency components. The phase calculation unit 10 includes DFTs 9A, 9B, 9
From the output of C, the above equation (5) is calculated to obtain α.

The bandpass filter 7 used here is used.
A, 7B, and 7C act on data sampled at eight times the fundamental frequency, but even if the fundamental frequency changes due to the density of the fluid, the sampling frequency changes accordingly, Since it follows the pass band of the band pass filter, it is not necessary to change the filter constant in the program.

By the above, α should be obtained with high accuracy, but in reality it is affected by the difference in the delay time of the circuit that digitizes the three signals and the delay of the adder circuit. Will end up. The effect is, for example, a phenomenon in which α calculated by the above equation (5) changes when the amplitude difference between the upstream side and the downstream side changes (when the amplitude difference is constant, Since it is processed as a constant offset, it can be dealt with by a simple correction). In order to avoid such an influence, specifically, the following two methods can be considered.

1) A method in which the amplitude difference between the upstream side and the downstream side is intentionally changed and the correction coefficient is obtained from the data at that time (see FIG. 3).
reference). When the amplitude difference between the upstream side and the downstream side changes, α in equation (5) changes because the sin angle in the equation includes a certain error, and the cause is mainly the addition. This is due to the delay of the circuit 2. Therefore, when an amplitude difference is generated between the upstream side and the downstream side by the variable gain amplifiers (gain control amplifiers) 12A and 12B, the DFT9
By correcting (advancing) the phase difference in the output θc of the adder circuit 2 obtained via B in the phase corrector 13, the delay of the adder circuit 2 that occurs on the circuit is eliminated. .

In order to obtain the coefficient (how much to advance in the program) used in such correction, when the phase difference between the upstream side and the downstream side is constant, the phase correction data search algorithm 14 causes the variable gain amplifier 12A, By changing the gain of 12B, the amplitude on the upstream side or the downstream side is intentionally changed to generate an amplitude difference, and the change in α at that time is recorded. The constant for correcting the phase θc is changed, and the constant for correcting is calculated by repeating the operation for calculating the constant for reducing the change. At this time, in order to measure the amplitude change on the upstream side and the downstream side within 1% and with an accuracy of 0.01% with respect to the maximum flow rate, the correction constant given to the phase θc is 0.005 degrees to 0.008 degrees. It is necessary to match with the accuracy of. Further, it is important that the variable gain amplifier does not generate a phase difference even if the amplitude difference is changed.

2) A method using a gain control amplifier for keeping the amplitude difference between the upstream side and the downstream side constant (see FIG. 4). This is because the variable gain amplifiers 12A and 12B are provided at the subsequent stage of the electromagnetic pickups 1A and 1B, and the peak hold units 15A and 15B that hold the peak value of the output thereof.
, A subtraction circuit 16 for taking the difference between the two peak voltage values, and a low-pass filter (LP for removing a noise component from the difference).
F) and 17 are provided.

That is, by adjusting the gains of the variable gain amplifiers 12A and 12B according to the difference between the peak voltage values of the upstream side and the downstream side, which are the outputs of the subtraction circuit 16, the amplitude difference between the upstream side and the downstream side is adjusted. To eliminate the effect of. The variable gain amplifiers 12A and 12B need to prevent a phase difference between their input and output even if their gains are changed. Further, here, the gain of the variable gain amplifier 12A is made variable, and the variable gain amplifier 12A
Although the gain of B is fixed, it goes without saying that this relationship may be reversed. In the above, an example of a Coriolis mass flowmeter has been mainly described, but the present invention is not limited to this, and can be applied to general devices that measure a phase difference between two signals having the same frequency.

[0037]

According to the present invention, the upstream pickup signal, the downstream pickup signal, and the output signals of the adder or subtractor circuit (arithmetic circuit) from the respective amplitudes and phases on the upstream side,
Since the phase difference of the downstream signal is calculated,
An advantage that the phase difference between the upstream and downstream signals can be obtained with high accuracy while reducing the number of bits of the A / D converter as compared with the method that uses only the phase of the downstream pickup signal is obtained. By the way, according to the present invention, in order to obtain the resolution of 0.01% = 0.2 nS with respect to the maximum flow rate,
The required number of bits of the A / D converter is 14-16.

Further, by using a comparator for binarizing the signal amplitude from a pair of detectors and a PLL circuit for generating a frequency signal which is an integral multiple of the output, a program used in digital filter calculation It is possible to change the cutoff frequency of the digital filter according to the change of the pipe vibration frequency due to the change of the fluid density without changing the constant of, and it is possible to improve the accuracy. Moreover, if the arithmetic means has a gain switching function, the detection range and resolution can be switched according to the phase difference to be detected, and in addition, the quantizing means,
The original signal can be faithfully reproduced by including sample-hold means for sample-holding the two signals and the outputs from the arithmetic means.

Further, amplitude varying means for varying the amplitudes of the pickup signals on the upstream side and the downstream side, and correction data when the amplitudes of the output signals of the pair of detectors change are obtained, and the above-mentioned values are calculated based on the data. A correction means for correcting the phase difference is added, or a signal amplitude difference detection means for detecting the amplitude difference between the pickup signals on the upstream side and the downstream side, and the pair of detectors based on the detected amplitude difference. Even if there is an amplitude difference between the upstream and downstream pickup signals by adding a gain control amplifier that matches the signal amplitude of the above, stable and highly accurate phase difference between the upstream and downstream signals can be obtained. You can

[Brief description of drawings]

FIG. 1 is a configuration diagram showing an embodiment of the present invention.

FIG. 2 is a vector diagram for explaining the operation of FIG.

FIG. 3 is a block diagram showing another embodiment of the present invention.

FIG. 4 is a block diagram showing still another embodiment of the present invention.

FIG. 5 is a principle configuration diagram of a U-shaped pipe type Coriolis mass flowmeter.

FIG. 6 is an explanatory diagram for explaining a vibration state of a U-shaped pipe.

FIG. 7 is an explanatory diagram of a vibration mode generated by the torsion torque of the Coriolis force of the U-shaped pipe.

FIG. 8 is a waveform diagram showing an example of a pickup signal when Coriolis force is generated in the U-shaped pipe.

FIG. 9 is a block diagram showing a phase difference detection circuit using a counter method.

FIG. 10 is a configuration diagram showing an example of a straight pipe type Coriolis mass flowmeter.

FIG. 11 is a block diagram showing a differential amplification type phase difference detection circuit.

FIG. 12 is a block diagram showing a fast Fourier transform type phase difference detection circuit.

FIG. 13 is an explanatory diagram for explaining the principle of phase difference detection by complex Fourier transform.

[Explanation of symbols]

1A, 1B ... Electromagnetic pickup, 2 ... Adder circuit, 3, 3
2 ... Comparator, 4 ... PLL (phase locked loop), 5
A, 5B, 5C ... Ancherias filter, 6A, 6
B, 6C, 62 ... Sample and hold circuit, 7A, 7B,
7C, 55, 63 ... A / D converter, 8A, 8B, 8C ...
Band pass filter, 9A, 9B, 9C ... DFT (discrete Fourier transfer), 10 ... Phase calculation section, 11 ... DSP (digital signal processor), 12A, 12B ... Variable gain amplifier, 13 ...
Phase correction unit, 14 ... Phase correction data search algorithm,
15A, 15B ... Peak hold circuit, 16 ... Subtraction circuit, 17 ... Low-pass filter, 21 ... U-shaped pipe, 22, 43
... Permanent magnets, 23, 40 ... Base, 24, 42 ... Electromagnetic drive coils, 25, 44 ... Support frame, 31, 52, 61 ... Amplifier, 33 ... Exclusive OR circuit, 34 ... Counter, 35 ...
Reference clock, 41 ... Straight pipe, 51 ... Difference device, 53 ... Narrow band filter, 54 ... Full wave rectifier / detector, 64 ... Data memory.

Claims (6)

[Claims] 1. An arithmetic means for obtaining a sum or a difference of two signals having the same frequency, a quantizing means for quantizing the output from the arithmetic means and the two signals, and a predetermined value from the quantized signal. A phase difference characterized by including band-pass filter means for extracting only frequency components and phase-difference calculation means for Fourier-transforming the output signal and performing a predetermined calculation to calculate the phase difference at the fundamental frequency of the signal. measuring device. 2. The quantizing means, a comparator for binarizing the amplitude of one of the two signals, and a PLL circuit for generating a frequency signal that is an integral multiple of the output of the comparator.
2. The phase difference measuring device according to claim 1, wherein the phase difference measuring device comprises an output from the calculation means and an A / D conversion means for quantizing the two signals with the output of the PLL circuit as timing. 3. The calculation means for obtaining the sum or difference of the two signals has a gain switching function, and the detection range and resolution can be switched according to the phase difference to be detected. 1. The phase difference measuring device according to 1. 4. The phase difference measuring device according to claim 1, wherein the quantizing means includes sample and hold means for sampling and holding the two signals and the outputs from the arithmetic means, respectively. 5. An amplitude varying means for changing the amplitude of at least one of the two signals, correction data when the amplitudes of the two signals are changed, and the phase difference is corrected based on the data. The phase difference measuring device according to claim 1, further comprising a correcting unit. 6. A signal amplitude difference detection means for detecting an amplitude difference between the two signals, and a gain for matching the amplitude of one of the two signals with the amplitude of the other signal based on the detected amplitude difference. The phase difference measuring device according to claim 1 or 2, further comprising a control amplifier.