JPH0785680B2 - Synchronous motor speed controller - Google Patents

Description

Description: BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a speed controller for a synchronous motor, and more particularly to a speed controller for a synchronous motor that controls the speed of the motor without using a rotation detector.

[Background of the Invention]

As a speed control device for this type of synchronous motor, there is known a thyristor motor device that controls the speed of the synchronous motor using a frequency converter. Such a speed control device includes a converter that supplies a variable frequency alternating current to a synchronous motor, and a converter control circuit that forms a value (amount) of the alternating current output from the converter and a signal that controls the frequency. It is configured. Such a speed control device controls the electric motor current to a predetermined phase with respect to the induced electromotive force of the electric motor, and in order to control the rotational speed, a rotation sensor such as a position detector or a speed detector is inevitably installed in the electric motor. Need to be attached to the shaft end. However, according to such a speed control device, it is necessary to attach these rotation sensors to the electric motor, which increases man-hours and the number of parts, and a signal cable from the sensor to the conversion device is required. In addition, there were inconveniences such as an increase in the number of parts, the number of parts, and lack of reliability because the rotation sensor was placed in a bad environment.

[Object of the Invention]

The present invention has been made in view of the above disadvantages, and an object of the present invention is to provide a speed control device for a synchronous motor capable of controlling the speed of the electric motor without using a rotation sensor.

[Outline of Invention]

The present invention relates to a converter (1) for supplying a variable frequency alternating current to a synchronous motor, and a converter control circuit (6) for controlling the value of the alternating current output from the converter and its frequency based on a rotation speed command. In the speed control device for a synchronous motor, the converter control circuit (6) receives the frequency command (kω ₁ ) of the output AC of the converter, and outputs a two-phase sine wave having a frequency corresponding to the frequency command. Oscillating means for oscillating a wave signal (12)
Voltage detecting means (4) for detecting the motor voltage of the synchronous motor, and the same phase as the effective component current related to the torque of the synchronous motor based on the detected motor voltage and the two-phase sine wave signal. Of the first voltage component (e _m ′) of the second and the second component of the same phase as the reactive current related to the amount of magnetic flux of the synchronous motor.
Voltage component (e _t ′) of the voltage component detection means (9,
10) and the deviation between the rotation speed command and the second voltage component (e _t ′), and the effective component current command ▲ (i ^* _t ) ▼ corresponding to the deviation to reduce the deviation. And a speed deviation increasing means (8) for outputting
Voltage component (e _m ′) of which the polarity is inverted and added, and is input to the oscillating means as the frequency command (kω ₁ ), and the frequency command output from the adding means and the first The deviation of the voltage component (e _t ′) of 2 is obtained, and in order to reduce the deviation, the reactive current command ▲ (i ^* _m ) is calculated according to the deviation.
A current increasing means (13) for generating ▼, a command ▲ (i ^* _m ) ▼ of the reactive current and a command ▲ (i ^* _t ) ▼ of the active current and
A phase number conversion means (14, 15) for outputting a current command of three phases based on the phase sine wave signal, and a converter drive circuit for driving the converter according to the current command output from the conversion means ( 16, 17) and is configured.

Example of Invention

 Embodiments of the present invention will be described below with reference to the drawings.

FIG. 1 is a block diagram showing an embodiment of a speed control device for a synchronous motor according to the present invention. In the figure, 1 is a converter that outputs an alternating current of a variable frequency, and this converter 1 is a gate turn-off thyristor.
(Hereinafter, referred to as GTO)} Or it is configured as a PWM inverter with transistors and diodes. Reference numeral 2 denotes a synchronous motor, which includes a stator 2A and a rotor (in this case, a field winding) 2B. 3 is a rotation speed command circuit, 4 is a motor voltage detecting transformer, and 5 is a current detector for detecting an instantaneous value of the output current of the inverter 1. Further, the converter control circuit 6 that has received the rotation speed command 100 from the rotation speed command circuit 3 is
The control signal 200 for controlling the magnitude and frequency of the alternating current output from the converter 1 can be formed and output based on the above. This converter control circuit 6 includes a rate-of-change limiter 7 for controlling the rate of change of the speed command signal, a speed command signal 100, and a phase reference signal of a reactive current, which is a fundamental wave component of a motor voltage described later. 90 degree The speed deviation amplifier 8 which takes the deviation from the phase difference component and increases the deviation signal ▲ i ^* _t ▼, and the phase reference signal of the reactive current and the phase reference signal of the reactive current of 90 degrees. A voltage component detector 9 for detecting a component having a different phase, and a voltage component detector 10 for detecting a component of the fundamental wave component of the motor voltage that is in phase with the phase reference signal of the reactive current. , An adder 11 for adding the output signals of the voltage component detector 10 and the change rate limiter 7 and outputting a frequency command signal (Kω ₁ ), and a frequency command signal Kω ₁
An oscillator 12 which outputs a two-phase sine wave signal having a frequency proportional to, and a reactive current command signal ▲ i ^* _m ▼ depending on the deviation between the output signal of the adder 11 and the output signal of the voltage component detector 9. Outputting amplifier 13 and reactive current command signal ▲ i ^* _m ▼ from amplifier 13 and active current command signal ▲ i ^* _t ▼ from amplifier 8.
And a signal output from the oscillator 12 are multiplied to output a two-phase current command pattern signal ▲ i ^* _α ▼ and ▲ i ^* _β ▼
14, a phase number converter 15 that outputs three-phase current command pattern signals ▲ i ^* _u , i ^* _v , i ^* _w ▼ based on the signals ▲ i ^* _α ▼ and ▲ i ^* _β ▼, and the current command. Compare the pattern signal with the current detection signal from the current detector 5, and turn on the GTO of the inverter 1.
Comparator 16 that outputs a signal that forms PWM (control) signal 200 for off control, and gate signal (control signal) 2 to GTO
And a gate circuit 17 for giving 00. The current detector 5, the gate circuit 17, and the comparator 16 are circuits corresponding to the U-phase output, and there are similar circuits corresponding to the V-phase and the W-phase, respectively. Omitted.

Next, the operation of the embodiment configured as described above will be described, but before that, the principle of the present invention will be described first.

One axis of the orthogonal rotating magnetic field coordinate system of the synchronous motor is m ′ axis,
'Represented by the horizontal and vertical axes assuming this m' an axis orthogonal to it t is controlled so as to coincide with the magnetic flux axis of the motor shaft, current components i _m 'and i _t' each of the axes It corresponds to the reactive component current component i _m related to the amount of magnetic flux of the motor and the active component current component i _t related to the torque of the motor.

Therefore, if i _t ′ is controlled by the signal of the speed control circuit and i _m ′ is controlled by the signal of the voltage control circuit, the electric motor can be controlled to a predetermined speed and voltage.

By the way, the output frequency (phase of current) of the inverter is
In the prior art, it was determined based on the signal from the position detector. However, in the present invention, instead of this, the frequency is determined by using the signal from the voltage detector 4 which detects the voltage applied to the electric motor.

Now, the control for matching the m'axis used in the present invention with the magnetic flux axis of the electric motor will be described below.

FIGS. 2 (I) and (II) show the m ′ axis and the magnetic pole when the active component current command ▲ i ^* _t ▼ from the amplifier 8 is constant and the reactive component current command ▲ i ^* _m ▼ is set to 0. It is a characteristic view which shows the change characteristic of each magnetic flux of an electric motor with respect to the phase difference with an axis (straight axis). Where φ _m ′ and φ _t ′ are the magnetic flux components in the m ′ and t ′ axis directions,
φ is the vector composite magnetic flux of φ _m ′ and φ _t ′. Further, FIGS. 2 (I) and (II) show that the current command ▲ i ^* _t ▼ is a positive rated value and the current command ▲ i ^* _m ▼ is zero, and the rated torque is generated in electric operation. FIG. 3 is a diagram showing a case where the vehicle is operated with no load, and FIG.

Hereinafter, description will be sequentially given from FIG.

In FIG. (I), the operating point indicated by X is the magnetic flux component φ _t ′ =
0, that is, the normal operating point where the m ′ axis coincides with the magnetic flux axis of the electric motor.

By the way, the sign of φ _t ′ is inverted between positive and negative at the boundary of this normal operating point. Therefore, when the magnetic flux component φ _t ′ is positive, the motor frequency f ₁ is increased to increase the phase angle θ ′ from the magnetic pole axis (straight axis) to the m ′ axis in the increasing direction, and conversely, the magnetic flux component φ _t ′.
Is negative, if the frequency f ₁ is lowered and the phase angle θ ′ is corrected and controlled to decrease, the operating point is always shown in the figure.
Moving to the mark, the magnetic flux component φ _t ′ = 0 is maintained.

Further, the control of the magnetic flux component φ _m ′ (which corresponds to the total magnetic flux φ _r when φ _t ′ = 0) is performed as follows. That is, since the magnetic flux component φ _m ′ changes according to the current component i _m ′, the magnetic flux component φ _m ′ is detected, the current component i _m ′ is controlled according to the variation from the predetermined value, and the magnetic flux component φ _{m ′.} It is sufficient to keep ′ at a predetermined value.

The characteristics shown in FIG. 2 (II) are the same as those in FIG. 2 (I), and the same control as described above may be performed.

As described above, the operation with a constant magnetic flux is performed. At this time, the induced electromotive force of the electric motor 2 (this is a fundamental wave component of the electric motor voltage and a voltage component having a phase difference of 90 degrees from the phase reference signal of the reactive current) is proportional to the electric motor frequency, that is, the rotation speed. To do. The present invention utilizes this development to detect the induced electromotive force of the electric motor 2 and feed the detection signal to the speed control circuit to perform speed control.

The operating principle of the present invention has been described above.

Next, the operation of the circuit of FIG. 1 which is an embodiment of the present invention will be described.

In the voltage component detectors 9 and 10, biaxial components of the motor voltage are calculated according to the following equations, that is, a voltage component e _t ′ and a component e _m ′ of the same phase which are 90 degrees different in phase from the phase reference signal of the reactive current
Are detected respectively.

Where v _α = v _u e _m ′: Output signal of detector 10 _et ′: Output signal of detector 9 v _u , v _v , v _w : Motor phase voltage ^cosω 1 ^t : phase reference signal of reactive current of U phase The above-described calculation can be realized by using, for example, a multiplier and an adder.

The signals e _m ′ and e _t ′ have the following relationship with φ _m ′ and φ _t ′ described above, ignoring the influence of the leakage impedance drop of the motor.

That is, the signal e _m 'from a magnetic flux component phi _t' worth of signal is detected. Signal e _m 'is applied to the adder 11 together with the output signal of the change rate limiter 7. At this time, the signal e _m 'is negative (phi _t' in the case of corresponding to> 0), the output signal of the adder 11 is large, i.e., so that the inverter output frequency is added by the polarity increases. In this way, the motor frequency f ₁ is controlled so that e _m ′ = 0 (φ _t ′ = 0) is always maintained according to the above-mentioned principle, and the phase angle θ ′ from the magnetic pole axis to the m ′ axis is the normal operation. Controlled by the value of the point.

In addition, since the induced electromotive force e _t ′ (φ _m ′) is proportional to the current command ▲ i ^* _m ▼, if the induced electromotive force e _t ′ is smaller than the predetermined value, the current command ▲ i ^* _m ▼ If the induced electromotive force e _t ′ is larger than the predetermined value, the induced electromotive force e _t ′ can be controlled to the predetermined value by changing the current command ▲ i ^* _m ▼ in the decreasing direction. . Therefore, in the thickener 13, the thickener 11
Depending on the output signal from the deviation between the signal e _t ', the current command ▲ i ^*
By controlling _m ▼, the following relationship is obtained.

k ₁ ω ₁ −e _t ′ = 0 (3) or _et ′ / ω ₁ = k ₁ …… (4) where k ₁ : proportional constant, that is, the magnetic flux component φ _m ′ (= e _t ′ / Ω ₁ ) can be automatically kept at a predetermined value.

On the other hand, the induced electromotive force e _t ′ is proportional to the magnetic flux component φ _m ′ and the angular frequency ω ₁ , as shown in the equation (2). Therefore, since the magnetic flux component φ _m ′ is maintained at a constant value by the above-described control, the induced electromotive force e _t ′ is proportional to ω ₁ . Therefore, by matching the rotation speed command signal from the rate-of-change limiter 7 and the signal e _t ′, and changing the active current command ▲ i ^* _t ▼ according to the deviation, the rotation speed can be adjusted according to the command value. Can be controlled.

Therefore, according to this embodiment, there is an advantage that stable frequency control can be performed without using a rotation sensor and highly accurate speed control can be performed.

FIG. 3 is a circuit diagram showing another embodiment of the present invention. This embodiment is an application example to a drive system using a converter in which a power supply side converter and a motor side inverter are connected via a DC circuit. In FIG. 3, the converter is
Converter 21 that controls the magnitude of the current supplied to the motor
A DC reactor 22 and an inverter 23 for controlling the motor frequency and current phase. Further, the difference of this embodiment from the embodiment shown in FIG. 1 is that the converter control circuit 6'includes a voltage component detector 9, a voltage component detector 10, an adder 11, an oscillator 12, and an amplifier. Circuit 24 for calculating a current phase command based on the output signals from the amplifier 8 and the amplifier 13, and an oscillator.
Based on the output signal from 12 as a reference, the phase control circuit 25 which determines the commutation timing of the inverter 23 based on the command signal from the arithmetic circuit 24, and the current based on the output signals from the amplifiers 8 and 13 An arithmetic circuit 26 that calculates the command signal I ^* and an arithmetic circuit
The amplifier 27 is configured to widen the deviation between the output signal of 26 and the output signal of the current detector 5, and the circuit 28 for determining the firing phase of the converter 21 according to the output signal from the amplifier 27. Has been done.

Next, the operation of the above embodiment will be described.

In the voltage component detectors 9 and 10, the biaxial components of the voltage of the electric motor 2 according to the equation (1), that is, the voltage component e _t ′ and the same phase which are different in phase by 90 degrees with respect to the phase reference signal of the reactive current Each of the components e _m ′ of is detected. Both components e _m ′ and e _t ′ are
Ignoring the effect of motor leakage impedance drop,
Since there is the second (2) of the relationship, 'by phi _t' this signal e _m corresponding signal can be detected. Signal e _m 'is applied to the adder 11 together with the output command signal 100 from the rotation speed command circuit 3. In this case, if e _m 'is negative (phi _t' equivalent to> 0), the output signal of the adder 11 is large, that is, added by the polarity inverter output frequency is increased. In this way, in accordance with the above-mentioned principle, e _m ′ = 0 (φ _t ′ = 0)
The frequency f ₁ is controlled so that the phase angle θ ′ from the magnetic pole axis (straight axis) to the m ′ axis is controlled to the value of the normal operating point.

In addition, since the induced electromotive force e _t ′ (φ _m ′) and the current command ▲ i ^* _m ▼ are in a proportional relationship, if the induced electromotive force e _t ′ is smaller than the predetermined value, the current command ▲ i ^* _m ▼ The induced electromotive force e _t ′ can be controlled to a predetermined value by changing the induced electromotive force e _t ′ in the increasing direction and conversely, when the induced electromotive force e _t ′ is larger than the predetermined value, in the decreasing direction. Therefore, the magnetic flux component φ _m ′ can be maintained at a predetermined value by controlling the current command ▲ i ^* _m ▼ in the amplifier 13 according to the deviation between the output signal of the amplifier 11 and the signal e _t ′. .

On the other hand, the induced electromotive force e _t ′ is proportional to the magnetic flux component φ _m ′ and the angular frequency ω ₁ as shown in the equation (2), but by controlling as described above, the magnetic flux component φ _m ′ is Since it is maintained at a constant value, the induced electromotive force e _t ′ is proportional to the angular frequency ω ₁ . Therefore, by controlling the output signal from the rotation speed command circuit 3 and the signal e _t ′ and changing the active component current command ▲ i ^* _t ▼ according to the deviation, the rotation speed is controlled according to the command value. can do.

The arithmetic circuit 24 outputs a signal for instructing the phase angle φ of the current i with respect to the t'axis shown in FIG. 4 based on the reactive current command signal ▲ i ^* _m ▼ and the active current command signal ▲ i ^* _t ▼. To do. The phase angle φ is shown by the following equation.

The commutation timing of the inverter 23 is determined in the phase control circuit 25 based on this phase command signal and the output signal from the oscillator 12, and at the same time, the control advance angle β is set. Thus, the inverter 23 is ignition-controlled by the signal from the phase control circuit 25.

In addition, the command value I ^* of the motor current is set to the reactive current command signal ▲ i ^*
_The calculation circuit 26 calculates from _m ▼ and the effective component current command signal ▲ i ^* _t ▼, and the deviation between this motor current command signal and the output signal from the current detector 5 is increased by the amplifier 27, and in response to this, The ignition phase of the converter 21 is controlled to adjust the current.

Therefore, according to this embodiment, similarly to the above-mentioned embodiment, without using the rotation sensor, there is no step out at the time of variable speed, stable frequency control can be performed, and highly accurate speed control can be performed. It Further, according to the present embodiment, since there is no rotation sensor, it can be used even in a bad environment.

[Effects of the Invention] As described above, according to the present embodiment, there is an effect that a rotation sensor is not required and highly accurate speed control is possible even when used in a bad environment.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing an embodiment of a speed control device for a synchronous motor according to the present invention, FIGS. 2 (I) and (II) are characteristic diagrams shown for explaining the principle of the present invention, and FIG. Is a circuit diagram showing another embodiment of the present invention, and FIG. 4 is a vector diagram shown for explaining another embodiment of the present invention. 1 ... Converter (inverter), 2 ... Synchronous motor, 3 ...
... Speed command circuit, 4 ... Voltage detector, 5 ... Current detector, 6 ... Converter control circuit, 8 ... Speed deviation amplifier, 9,
10 …… Voltage component detector, 11 …… Adder that outputs frequency command signal, 12 …… Oscillator, 13 …… Amplifier that outputs reactive current command signal.

Claims (1)

[Claims] 1. A converter (1) for supplying a variable frequency alternating current to a synchronous motor, and a converter control circuit (6) for controlling the value of the alternating current output from the converter and its frequency based on a rotational speed command. In the speed control device for a synchronous motor, the converter control circuit (6) inputs the frequency command (kω ₁ ) of the output AC of the converter, and outputs the frequency of 2 according to the frequency command. An oscillating means (12) for oscillating a phase sine wave signal, a voltage detecting means (4) for detecting a motor voltage of the synchronous motor, and the synchronization based on the detected motor voltage and the two-phase sine wave signal. A first voltage component (e _m ′) in phase with the active component current related to the torque of the motor and a second voltage component (e _t ′) in phase with the reactive component current related to the amount of magnetic flux of the synchronous motor. ) Voltage component detection means (9, 10) for detecting Speed serial a deviation between the and the rotational speed command second voltage component (e _t '), and outputs the command for the active current ▲ (i ^* _t) ▼ corresponding to the deviation in order to reduce the deviation Deviation increasing means (8) and adding means for adding the first voltage component (e _m ′) to the rotation speed command by inverting the polarity and inputting it as the frequency command (kω ₁ ) to the oscillating means. (11), the deviation between the frequency command output from the adding means and the second voltage component (e _t ′) is obtained, and the command of the reactive current ▲ ( i ^* _m ) ▼ current increasing means (13), the reactive component current command ▲ (i ^* _m ) ▼, the active component current command ▲ (i ^* _t ) ▼, and the two-phase sine wave Phase number conversion means (14, 15) for outputting a three-phase current command based on the signal, and the converter is driven according to the current command output from the conversion means. And a converter drive circuit (16, 17) for controlling the speed of the synchronous motor.