JPH0750987B2 - Resonance type DC-DC converter control method - Google Patents

Description

The present invention relates to a control method of a resonance type DC-DC converter that can improve the efficiency thereof.

[Conventional technology]

In order to reduce the switching loss of the switching semiconductor element for the purpose of improving the efficiency and downsizing of the DC-DC converter, the vibration phenomenon of the inductance (L) and the capacitance (C) is used to switch the switching semiconductor element. A resonant converter has been proposed in which current and voltage are not applied at the same time to reduce switching loss.

These resonant converters are roughly classified into three types according to the state of the current flowing through the switching semiconductor element and the voltage applied thereto in the vicinity of the switching operation point. The first
Is a zero current switching type in which the current is almost zero immediately after turn-on and immediately before turn-off of the switching semiconductor device, the second is a zero voltage switching type in which the voltage is almost zero immediately before turn-on and immediately after turn-off, and the third is current / voltage switching. It is a zero-current / zero-voltage switching type that has almost zero before and after the operating point.

[Problems to be solved by the invention]

The main causes of power loss in the switching semiconductor element of a high-frequency converter are the relationship between the main terminal capacitance C _{os of} the switching semiconductor element and the switching characteristics, the on-voltage, and the converter operating power factor.

The conventional zero-current switching type can realize a circuit with a good operating power factor by clamping the voltage of the resonance capacitor with a diode, but since it is not zero-voltage switching, there is power loss due to the capacitance _Cos and the frequency is The higher this is, the greater the problem of this power loss.

In addition, the zero-voltage switching type has no power loss due to the capacitance C _os , but when operating in a wide power control range, the current at turn-off of the switching semiconductor element and the current regenerated in the input power supply increase and the operating power factor increases. It is difficult to operate with high efficiency. In addition, the conventional zero-current / zero-voltage switching type deviates from the zero-current / zero-voltage switching mode when power control is performed, and the zero-current switching type or zero-voltage switching type operation is performed. Loss increases. Therefore,
This method is not suitable for DC-DC converters that control large power over a relatively wide range.

Therefore, in the past, for large power capacity, a zero-current switching converter that uses an IGBT or the like to operate them at a relatively low frequency has been used.
A zero-voltage switching converter that operates at high frequency using SFET was used. Therefore, the large power capacity
It was extremely difficult to reduce the size of the DC-DC converter at the same time as making it higher in frequency and operating it with high efficiency.

Here, the power loss of the resonant converter that is not the zero-voltage switching type or operates outside the zero-voltage switching type operation mode will be described.

Table 1 shows the amount of charge and discharge of the capacitance between the main terminals of MOSFETs and IGBTs used as switching semiconductor devices and the voltage applied during switching. In addition, MOSFE
T is the case where two pieces are connected in parallel, and both the MOSFET and the IGBT have the charge amount for one arm.

Next, the power loss due to the charging / discharging of the capacitance between the main terminals of the switching semiconductor element becomes the product of the input voltage, the charge amount, and the frequency in the worst operation mode.
(B) shows the relationship between the operating frequency and the power loss. From this figure, it is clear that as the operating frequency of the switching semiconductor device and the voltage between the main terminals increase, the power loss due to charging and discharging of the capacity between the main terminals increases.

[Means and Actions for Solving Problems]

In the present invention, in order to eliminate the above-mentioned conventional drawbacks, in a DC-DC converter that generates an output by utilizing series resonance of a resonance capacitor and a resonance inductor caused by alternating opening and closing of a switching semiconductor element, After the turning off of one of the switching semiconductor elements, the magnetic energy stored in the inductance during the conduction period stores energy in one parasitic capacitance of the switching semiconductor element and the energy stored in the other parasitic capacitance of the switching semiconductor element. Is discharged, and when the voltage across the other side of the switching semiconductor element drops below a sufficiently low set voltage, the other side of the switching semiconductor element is turned on. Therefore, in the converter according to the present invention, the power loss due to the parasitic capacitance between the main terminals of the switching semiconductor element is practically negligible.

〔Example〕

The present invention provides a control method capable of operating a resonant converter in a substantially zero current / zero voltage switching operation over a wide power control range.

Embodiments of the present invention will be described below with reference to the drawings.

FIG. 1 shows a resonance type DC for realizing the control method of the present invention.
-A schematic circuit configuration of a DC converter is shown. In this figure, 1 and 2 are DC input power supplies, 3 and 4 are MOSFETs, switches indicating switching semiconductor elements such as IGBTs, 5 and 6 are parasitic capacitances between main terminals of the main switch, 7 and 8 are diodes, and 9 and Reference numeral 10 is a resonance capacitor, 11 and 12 are clamping diodes, 13 is a resonance inductor, 14 is a transformer having a primary winding N ₁ and a secondary winding N _2, and 15 and 16 are main switches 3 and 4. Voltage monitoring circuit that monitors the voltage across both ends of the
4 is a drive circuit, 19 is a controller, 20 is a rectifier circuit connected to the secondary winding N ₂ of the transformer 14, 21 is a load, 22 is a voltage detection circuit, and 23 is a smoothing capacitor. The diode 7
And 8 are parasitic diodes of the main switches 3 and 4 when they are MOSFETs, and diodes connected in parallel when they are IGBTs. If the main switch is an element with a slow switching speed such as an IGBT, it is desirable to connect a capacitor of appropriate capacity in parallel with the element as necessary. Furthermore, the resonance inductor 13 may be removed and the leakage inductance of the transformer 14 may be used instead of the inductance.

Next, the operation of the circuit having such a configuration will be described.

First current of the third figure main switches 3 and 4, respectively voltage I _3,
I ₄ , V ₃ , V ₄ and the primary winding voltage of the transformer 14 is V _N1 . When the main switch 4 is on, the current supply to the transformer 14 is terminated, and the exciting current of the transformer 14 circulates through the primary winding N ₁ , the resonance inductor 13, the main switch 4, and the diode 12. I shall. This exciting current corresponds to a flat portion where the level of the current I ₄ of the main switch 4 is low. When the main switch 4 is turned off next in this state, the exciting current of the primary winding N ₁ of the transformer 14 starts discharging the charge stored in the parasitic capacitance 5 and charging the parasitic capacitance 6. Along with this charge and discharge
The voltage of N _{2 in} the secondary winding of 14 rises. Secondary winding N ₂
Voltage is the output voltage of, i.e. reaches the voltage of the smoothing capacitor 23, the rectifying circuit 20 becomes conductive, the exciting current of the transformer 14 starts to flow in the secondary winding of N _2, the primary winding N ₁ Although the exciting current decreases, the parasitic capacitors 5 and 6 continue to be discharged and charged because there is energy stored in the resonance inductor 13, and the voltage across the main switch 3 becomes zero. FIG. 4 shows a waveform diagram of each part in which the time axis of this switching operation section is enlarged.
The voltage monitoring circuit 15 monitors the voltage across the main switch 3, and when the voltage across the main switch 3 becomes zero, it sends a detection signal to the drive circuit 17 to send it to the drive circuit 17.
To turn on the main switch 3 at zero voltage. Since the charging and discharging of the parasitic capacitance during this period is performed by the exciting current of the transformer 14 and the current of the resonance inductor 13, the circuit loss due to the parasitic capacitance of the main switch is extremely small, and the main switch 3 is virtually switched at zero voltage. Turn on without loss.

When the main switch 3 is turned on, charging of the resonance capacitor 10 and discharge of the resonance capacitor 9 by the DC input power supply form series resonance by the action of the resonance inductor 13 and pass through the transformer 14, the rectifier circuit 20, etc. Send the series resonant current to the DC output. Initially, the series resonance current becomes a sine wave due to the actions of the resonance capacitors 9 and 10 and the resonance inductor 13. Then, when the voltage of the resonance capacitor 9 becomes zero, the crimping diode 11 becomes conductive, the resonance capacitor 9 is clamped to zero voltage, and the resonance capacitor 10 is clamped to the DC input power supply voltage, thereby ending the LC resonance mode. The mode in which the current of the resonance inductor 13 linearly decreases is entered. In this output current supply period, the exciting current of the transformer 14 has a small value, but increases due to the relationship determined by the number of turns of the secondary winding N ₂ and the output voltage. When the current I ₁₃ shown in FIG. 4 of the resonance inductor 13 becomes equal to the primary conversion value of the exciting current of the transformer 14, the rectifier circuit 20 becomes non-conducting and the half cycle of current supply to the output ends and the current Enter the supply suspension period. Up to this point, the excitation energy accumulated in the transformer 14 flows through the resonance inductor 13, the clamping diode 11, and the main switch 3, and the voltage of the primary winding N ₁ of the transformer 14 is almost zero. Therefore, the exciting current of the transformer 14 is kept substantially constant while the main switch 3 is in the conducting state.

Next, as in the case of turning on the main switch 3, the main switch 4 is turned on at the time when the voltage across the main switch 4 becomes zero (detected by the voltage monitoring circuit 16). The cycle begins. By repeating such an operation, zero voltage switching, that is, the main switches 3 and 4 can be turned on in the state where no voltage is applied.

Next, the turn-off control will be described. The turn-off of the main switches 3 and 4 is controlled based on the error signal obtained by comparing the reference signal with the detection voltage signal from the output voltage detection circuit 22 by the error amplifier of the controller 19. This error signal controls the turn-off times of the main switches 3 and 4 without controlling the turn-on time of the main switches as in the conventional case.
The output voltage is controlled by controlling the turn-off of the main switches 3 and 4 in accordance with the error signal. Therefore,
The zero voltage turn-on operation of the main switches 3 and 4 can always be maintained while controlling the output power.

Here, when the main switch is turned off, the exciting current necessary to make the voltage across the one main switch to be turned on next zero is flowing through the other main switch, as described above. Although it is not a zero current turn-off, the exciting current is much smaller than the main resonance current because the parasitic capacitances of the main switches 3 and 4 are sufficiently smaller than the resonance capacitors 9 and 10. Also, when the main switch is turned off, the charging charge of those parasitic capacitances is zero. In this circuit, the circuit constants are set so that the main switch is completely turned off before charging the parasitic capacitance. The voltage immediately after the turn-off is zero, that is, the zero-voltage turn-off is performed. Therefore, the main switch can be turned off with virtually zero current and zero voltage, and extremely low loss turn-off can be realized.

Next, referring to FIG. 2, a case will be described in which the resonance voltage control inductor 25 is provided between the connection points of the DC input power supplies 1 and 2 and the connection points of the resonance capacitors 9 and 10. A configuration in which a pair of capacitors connected in series for dividing the voltage of a single DC input power supply is provided and a resonance voltage control inductor 25 is provided between the connection point and the connection point between the resonance capacitors. But it's okay. Further, the resonance voltage control inductor may be connected to the negative terminal of the DC input power source through the capacitor from the connection point between the resonance capacitors.

This resonance voltage control inductor 25 is for reducing the resonance current at the time of low current output, and is inserted for the purpose of improving the output control at light load. The resonance voltage control inductor 25 includes a resonance inductor 13, a resonance capacitor 9,
After the main resonance of each half cycle of 10 is completed, the second resonance is performed with the resonance capacitors 9 and 10, and the resonance capacitor 9 and 10
Vary the voltage of 10. Then, the main resonance current can be reduced by turning on the next main switch in a phase in which the voltage of the resonance capacitor is low. This makes it possible to cope with light loads. The frequency of the second resonance is selected to be lower than the frequency of the main resonance, and the ratio of these is the approximate range of the control frequency required to control the output power from no load to the rated load. When the main resonance current is reduced in response to a light load, the output current supply period is shortened as shown by I ₀ in Fig. 5, and the excitation energy accumulated in the transformer 14 is naturally reduced. Also decreases. Therefore, the parasitic capacitances 5 and 6 of the main switches 3 and 4 are charged and discharged by the action of the exciting current,
Although the exciting current is insufficient to turn on the main switches 3 and 4 at zero voltage, the resonance voltage controlling inductor 25 acts to increase the exciting current by reapplying the voltage to the transformer 14. Energizing current sufficient to turn on the switches 3 and 4 at zero voltage can be secured. Therefore, the main switch can be turned on at zero voltage even under light load.

In addition, when the main switch is turned off, as in the steady state, the main switch is turned off at zero voltage in the state where only the exciting current is flowing, so an extremely low loss turn-off can be realized.

FIG. 4 shows the results of power loss production by the conventional method and the method of the present invention. It can be seen from the figure that the power loss increases sharply in the conventional method as the frequency increases, but the power loss hardly changes in the method of the present invention.

〔The invention's effect〕

As described above, according to the present invention, (1) the parasitic capacitance of the switching semiconductor element used as the main switch is charged and discharged by the stored magnetic energy of the inductance. There is virtually no power loss.

(2) Since the switching semiconductor element is turned on when the voltage across the switching semiconductor element becomes zero as the parasitic capacitance of the switching semiconductor element is discharged, zero current / zero voltage turn-on can be realized. ， No turn-on loss occurs.

(3) The main switch is controlled to turn off in the state that the exciting current of the transformer, which is much smaller than the main resonance current, is flowing, and the parasitic capacitance of the main switch is used to realize the zero voltage turn-off. The loss can be made so small that it can be ignored.

(4) In response to the error signal depending on the output voltage, the turn-off time of the main switch can be controlled to keep a small exciting current constant during the output current quiescent period, so that the quiescent period can be greatly controlled to control the output. It is possible to maintain zero current and zero voltage switching even if the operation is performed over a wide range.

It is possible to provide a control method for a resonant converter having the effects described above.

[Brief description of drawings]

1 and 2 are diagrams showing schematic circuit configurations of different resonance type D-DC converters for carrying out the control method according to the present invention, and FIG. 3 is each part used for explaining the present invention. FIG. 4 is a diagram showing details of the operation waveforms of the respective parts of the present invention, especially at the time of switching, and FIG. 5 is an operation waveform diagram of the respective parts used for explaining the present invention at light load, FIG. 6 is a diagram for comparing the power loss between the method of the present invention and the conventional method, and FIG. 7 is a diagram showing the power loss associated with charging and discharging of the capacitance between main terminals of the switching semiconductor element according to the conventional method. 1,2 ...... DC input power supply, 3,4 ...... Main switch, 5,6 ...... Main switch capacitance between main terminals, 9,10 ...... Resonance capacitor, 11,12 ...... Voltage clamp diode, 13 …… Resonance inductor, 14 …… Transformer, 15,16 …… Voltage monitoring circuit, 17,18 …… Drive circuit, 19 …… Controller, 25 …… Resonance voltage control inductor,

Claims (3)

[Claims] 1. A pair of resonance capacitors and a pair of switching semiconductor elements connected in series are connected in parallel to a DC input power source, and a diode is connected in parallel to the resonance capacitor to connect the switching semiconductor elements to each other. A resonance inductor is connected in series with the primary winding of the transformer between the point and the connection point between the resonance capacitors, and a direct current is supplied via a rectifying circuit and a smoothing circuit connected to the secondary winding of the transformer. In a control method of a resonance type DC-DC converter configured to obtain an output, the output is generated by utilizing series resonance of the resonance capacitor and the resonance inductor caused by alternating opening and closing of the switching semiconductor element. , A magnetic energy stored in the transformer and the resonant inductor during a conduction period after one of the switching semiconductor elements is turned off. Energy is stored in one of the parasitic capacitances of the switching semiconductor element, the energy stored in the other parasitic capacitance of the switching semiconductor element is discharged, and the voltage across the other end of the switching semiconductor element is set sufficiently low. A method for controlling a resonance type DC-DC converter, characterized in that the other of the switching semiconductor elements is turned on when the voltage drops below a voltage value. 2. A pair of resonance capacitors and a pair of switching semiconductor elements connected in series are respectively connected in parallel to a DC input power source, and a diode is connected in parallel to the resonance capacitor, and the DC input power source side and A resonance voltage control inductor is connected between the resonance capacitors, and the primary winding of the transformer is connected in series between the switching semiconductor elements and the resonance capacitors. A resonance inductor is connected, and a DC output is obtained through a rectifying circuit and a smoothing circuit connected to the secondary winding of the transformer, and the resonance occurs when the switching semiconductor element is alternately opened and closed. Output using series resonance between a capacitor and the resonance inductor and series resonance between the resonance conductor and the resonance voltage control inductor A method of controlling a resonant DC-DC converter for generating and controlling, wherein one of the switching semiconductor elements is turned on by one of the switching semiconductor elements by magnetic energy stored in the transformer and the resonant inductor after the one of the switching semiconductor elements is turned off. Energy is stored in the parasitic capacitance of the switching semiconductor element, the energy stored in the other parasitic capacitance of the switching semiconductor element is discharged, and when the voltage across the other end of the switching semiconductor element drops below a sufficiently low set voltage value. A method of controlling a resonance type DC-DC converter, characterized in that the other one of the switching semiconductor elements is driven to be turned on. 3. The method of controlling a resonance type DC-DC converter according to claim 1, wherein the leakage inductance of the transformer is used instead of the inductance of the resonance inductor. After turning off one of the two, the magnetic energy stored in the leakage inductance of the transformer during its conduction period stores energy in one parasitic capacitance of the switching semiconductor element and in the other parasitic capacitance of the switching semiconductor element. The resonance type DC-DC, characterized in that the stored energy is discharged and the other end of the switching semiconductor device is turned on when the voltage across the other end of the switching semiconductor device drops below a sufficiently low set voltage value. Converter control method.